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Input-Powered Interface Circuits for Electrodynamic Vibrational Energy Harvesting Systems

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Title:
Input-Powered Interface Circuits for Electrodynamic Vibrational Energy Harvesting Systems
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1 online resource (160 p.)
Language:
english
Creator:
Rao, Yuan
Publisher:
University of Florida
Place of Publication:
Gainesville, Fla.
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Thesis/Dissertation Information

Degree:
Doctorate ( Ph.D.)
Degree Grantor:
University of Florida
Degree Disciplines:
Electrical and Computer Engineering
Committee Chair:
Arnold, David P
Committee Members:
Nishida, Toshikazu
Bashirullah, Rizwan
Sodano, Henry

Subjects

Subjects / Keywords:
converter -- electrodynamic -- energy -- harvesting -- human -- input -- powered -- vibration
Electrical and Computer Engineering -- Dissertations, Academic -- UF
Genre:
Electrical and Computer Engineering thesis, Ph.D.
bibliography   ( marcgt )
theses   ( marcgt )
government publication (state, provincial, terriorial, dependent)   ( marcgt )
born-digital   ( sobekcm )
Electronic Thesis or Dissertation

Notes

Abstract:
Vibrational energy harvesting systems that convert ambient mechanical energy in the environment to usable electrical energy represent a promising emerging technology to achieve autonomous, self-renewable, and maintenance-free operation of wireless electronic devices and systems. Typical energy harvesting systems are composed of three components: an energy harvester that converts the mechanical vibrations into electrical energy, an interface circuit that conditions and regulates the energy, and an electronic load that uses or stores the harvested energy. This dissertation specifically focuses on the development and experimental characterization of input-powered energy harvesting circuits, including ac/dc converters and a dc/dc converter, for electrodynamic vibrational energy harvesters. This input-powered feature allows the active interface circuitry to automatically enter a zero-power-consumption standby mode when the voltage from the harvester is below a threshold level, thus eliminating any energy drain between energy harvesting cycles. Implemented in a 0.5 µm CMOS technology, the interface circuit is bench-top characterized with a sine wave signal generator and also with real vibrational energy harvesters. The measurement result shows that the minimum input threshold voltage is 1 V at open-load. When the ac input amplitude is 2.6 V and regulated dc output is 3.7 V, the interface circuit can achieve a peak net efficiency of 61% with 16.7 mW of output power delivered. A simplified equivalent circuit model for a resonant-type electrodynamic energy harvesting system is developed including a lumped element model (LEM) for the resonant harvester, a simplified interface circuit model, and a load model. The overall system model is validated via comparison of circuit simulations with experimental measurements. Lastly, a complete and fully self-sufficient energy harvesting system is demonstrated using the input-powered interface circuit and a non-resonant electrodynamic harvester, designed specifically for harvesting energy from human movements. Tested under normal human activities (walking, jogging, cycling), the 70 cm3 system is shown to charge a 3.7 V rechargeable battery with an average power of 234 µW during jogging.
General Note:
In the series University of Florida Digital Collections.
General Note:
Includes vita.
Bibliography:
Includes bibliographical references.
Source of Description:
Description based on online resource; title from PDF title page.
Source of Description:
This bibliographic record is available under the Creative Commons CC0 public domain dedication. The University of Florida Libraries, as creator of this bibliographic record, has waived all rights to it worldwide under copyright law, including all related and neighboring rights, to the extent allowed by law.
Statement of Responsibility:
by Yuan Rao.
Thesis:
Thesis (Ph.D.)--University of Florida, 2013.
Local:
Adviser: Arnold, David P.

Record Information

Source Institution:
UFRGP
Rights Management:
Applicable rights reserved.
Classification:
lcc - LD1780 2013
System ID:
UFE0045301:00001

MISSING IMAGE

Material Information

Title:
Input-Powered Interface Circuits for Electrodynamic Vibrational Energy Harvesting Systems
Physical Description:
1 online resource (160 p.)
Language:
english
Creator:
Rao, Yuan
Publisher:
University of Florida
Place of Publication:
Gainesville, Fla.
Publication Date:

Thesis/Dissertation Information

Degree:
Doctorate ( Ph.D.)
Degree Grantor:
University of Florida
Degree Disciplines:
Electrical and Computer Engineering
Committee Chair:
Arnold, David P
Committee Members:
Nishida, Toshikazu
Bashirullah, Rizwan
Sodano, Henry

Subjects

Subjects / Keywords:
converter -- electrodynamic -- energy -- harvesting -- human -- input -- powered -- vibration
Electrical and Computer Engineering -- Dissertations, Academic -- UF
Genre:
Electrical and Computer Engineering thesis, Ph.D.
bibliography   ( marcgt )
theses   ( marcgt )
government publication (state, provincial, terriorial, dependent)   ( marcgt )
born-digital   ( sobekcm )
Electronic Thesis or Dissertation

Notes

Abstract:
Vibrational energy harvesting systems that convert ambient mechanical energy in the environment to usable electrical energy represent a promising emerging technology to achieve autonomous, self-renewable, and maintenance-free operation of wireless electronic devices and systems. Typical energy harvesting systems are composed of three components: an energy harvester that converts the mechanical vibrations into electrical energy, an interface circuit that conditions and regulates the energy, and an electronic load that uses or stores the harvested energy. This dissertation specifically focuses on the development and experimental characterization of input-powered energy harvesting circuits, including ac/dc converters and a dc/dc converter, for electrodynamic vibrational energy harvesters. This input-powered feature allows the active interface circuitry to automatically enter a zero-power-consumption standby mode when the voltage from the harvester is below a threshold level, thus eliminating any energy drain between energy harvesting cycles. Implemented in a 0.5 µm CMOS technology, the interface circuit is bench-top characterized with a sine wave signal generator and also with real vibrational energy harvesters. The measurement result shows that the minimum input threshold voltage is 1 V at open-load. When the ac input amplitude is 2.6 V and regulated dc output is 3.7 V, the interface circuit can achieve a peak net efficiency of 61% with 16.7 mW of output power delivered. A simplified equivalent circuit model for a resonant-type electrodynamic energy harvesting system is developed including a lumped element model (LEM) for the resonant harvester, a simplified interface circuit model, and a load model. The overall system model is validated via comparison of circuit simulations with experimental measurements. Lastly, a complete and fully self-sufficient energy harvesting system is demonstrated using the input-powered interface circuit and a non-resonant electrodynamic harvester, designed specifically for harvesting energy from human movements. Tested under normal human activities (walking, jogging, cycling), the 70 cm3 system is shown to charge a 3.7 V rechargeable battery with an average power of 234 µW during jogging.
General Note:
In the series University of Florida Digital Collections.
General Note:
Includes vita.
Bibliography:
Includes bibliographical references.
Source of Description:
Description based on online resource; title from PDF title page.
Source of Description:
This bibliographic record is available under the Creative Commons CC0 public domain dedication. The University of Florida Libraries, as creator of this bibliographic record, has waived all rights to it worldwide under copyright law, including all related and neighboring rights, to the extent allowed by law.
Statement of Responsibility:
by Yuan Rao.
Thesis:
Thesis (Ph.D.)--University of Florida, 2013.
Local:
Adviser: Arnold, David P.

Record Information

Source Institution:
UFRGP
Rights Management:
Applicable rights reserved.
Classification:
lcc - LD1780 2013
System ID:
UFE0045301:00001


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1 INPUT POWERED INTERFACE CIRCUITS FOR ELECTRODYNAM IC VIBRATIONAL ENERGY HARVEST I NG SYSTE M S By YUAN RAO A DISSERTATION PRESENTED TO THE GRADUATE SCHOOL OF THE UNIVERSITY OF FLORIDA IN PARTIAL FULFILLMENT OF THE REQUIREMENTS FOR THE DEGREE OF DOCTOR OF PHILOSOPHY UNIVERSITY OF FLORIDA 2013

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2 2013 Yuan Rao

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3 To the memory of my father

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4 ACKNOWLEDGMENTS This work was funded in part by Texas Instruments through a fellowship. My most heartfelt thanks go to my advisor, Dr. David P. Arnold, for his tremendous guidance from day one of the project to the complet ion of this dissertation. I would not have been where I am today without his encouragement and kind ness I would also like to acknowledge my advisory committee members for their guidance and evaluation of this work: Dr. Toshikazu Nishida, Dr. Rizwan Bashirullah and Dr. Henry Sodano I would like to sincerely thank all my current and previous colleague s in the Interdisciplinary Microsystems Group (IMG) who helped me in the past few years. Specifically, I want like to thank Dr. Shuo Cheng Dr. Vinod Challa and Kelly McEachern for their great help on the harvester, Ying Jing and Yaxing Zhang for their val uable discussions. I am thankful to Jessica Meloy Dylan Alexander, Xiaoyu Cheng and all other IMG committee members for their help on maintaining the operation of IMG labs. I am grateful to my mentors at Texas Instruments during my internship who helped me understand what power electronics industry is all about. Thank s to Chris Sanzo, Siyuan Zhou and Fred Trafton I have been fortunate to have my beloved husband Jikai Chen. He is always the strongest shoulder I can lean in my most difficult tim es. I owe everything to him for his love patience and help throughout this PhD process. Finally, I want to thank my dearest parents : my father Jianmin Rao and my mother Jianhua Li for their unconditional love and support to me. I miss my father every day and to dedicate this dissertation to him is the least I can do.

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5 TABLE OF CONTENTS ACKNOWLEDGMENTS ................................ ................................ ................................ .. 4 LIST OF TABLES ................................ ................................ ................................ ............ 8 LIST OF FIGURES ................................ ................................ ................................ .......... 9 LIST OF ABBREVIATIONS ................................ ................................ ........................... 14 ABSTRACT ................................ ................................ ................................ ................... 15 CHAPTER 1 INTRODUCTION ................................ ................................ ................................ .... 17 1.1 Vibrational Energy Harvesters ................................ ................................ .......... 19 1.1.1 Electrostatic Vibrational Energy Harvester ................................ ............ 20 1.1.2 Piezoelectric Vibrational Energy Harvest er ................................ ............ 22 1.1.3 Electrodynamic Vibrational Energy Harvester ................................ ....... 24 1.2 Vibrational Energy Harvesting Interface Circuits ................................ .............. 26 1.2.1 Circuit Design Challenges ................................ ................................ ..... 27 1.2.2 Literature Review ................................ ................................ ................... 28 1.2.3 Input powered Energy Harvesting Circuit ................................ .............. 40 1.3 Vibrational Energy Harvesting Systems ................................ ........................... 42 1.3.1 System Design Challenges ................................ ................................ .... 42 1.3.2 Literature Review ................................ ................................ ................... 43 1.4 Research Objectives ................................ ................................ ........................ 45 1.5 Dissertation Organization ................................ ................................ ................. 46 2 INPUT POWERED AC/DC CONVERTERS ................................ ........................... 48 2.1 Half wave Ac/dc Converter ................................ ................................ ............... 48 2.1.1 Circuit Design ................................ ................................ ........................ 48 2.1.2 Ci rcuit Implementation ................................ ................................ ........... 52 2.1.3 Measurement Result ................................ ................................ .............. 52 2.2 Full wave Ac/dc Converter ................................ ................................ ............... 55 2.2.1 Circuit Design ................................ ................................ ........................ 55 2.2.2 Circuit Implementation ................................ ................................ ........... 57 2.2.3 Measurement Result ................................ ................................ .............. 57 2.3 Voltage Doubling Ac/dc Converter ................................ ................................ ... 61 2.3.1 Circuit Design ................................ ................................ ........................ 61 2.3.2 Circuit Implementation ................................ ................................ ........... 64 2.3.3 Measurement Result ................................ ................................ .............. 65 2.4 Summary ................................ ................................ ................................ .......... 70 3 INPUT POWERED DC/DC CONVERTER ................................ .............................. 75

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6 3.1 Circuit Design ................................ ................................ ................................ ... 76 3.1.1 Pulse Skip Modulation ................................ ................................ ........... 76 3.1.2 Circuit Diagram ................................ ................................ ...................... 79 3.1.3 Error Comparator ................................ ................................ ................... 81 3.1.4 Level Shifter ................................ ................................ ........................... 82 3.1.5 Voltage Controlled Oscillator ................................ ................................ 84 3.1.6 Switching MOSFET and Buffer ................................ .............................. 85 3.2 Circuit Implementation ................................ ................................ ...................... 86 3.3 Experimental Result ................................ ................................ ......................... 89 3.3.1 Function Test ................................ ................................ ......................... 89 3.3.2 Power Efficiency ................................ ................................ .................... 91 3.4 Su mmary ................................ ................................ ................................ .......... 92 4 COMPLETE INPUT POWERED INTERFACE CIRCUIT ................................ ........ 94 4.1 Circuit Design ................................ ................................ ................................ ... 94 4.2 Experi mental Result ................................ ................................ ......................... 96 4.2.1 Measurement Setup ................................ ................................ .............. 96 4.2.2 Minimum Operating Voltage ................................ ................................ .. 97 4.2.3 Output Power and Efficiency ................................ ................................ 99 4.3 Summary ................................ ................................ ................................ ........ 101 5 RESONANT ELECTRODYNAMIC ENERGY HARVESTING SYSTEM MODELING ................................ ................................ ................................ ........... 102 5.1 Reduced order Models ................................ ................................ ................... 103 5.1.1 Electrodynamic Energy Harvester Model ................................ ............. 103 5.1.2 Interface Circuit Model ................................ ................................ ......... 109 5.1.3 Load Model ................................ ................................ .......................... 114 5.2 Model Parameter Extraction ................................ ................................ ........... 114 5.2.1 Electrodynamic Energy Harvester Parameters ................................ .... 114 5.2.2 Interface Circuit Parameters ................................ ................................ 125 5.3 Energy Harvesting System Modeling ................................ ............................. 128 5.4 Summary ................................ ................................ ................................ ........ 131 6 NON RESONANT ENERGY HARVESTING SYSTEM FOR HUMAN MOVEMENTS ................................ ................................ ................................ ....... 133 6.1 System Design ................................ ................................ ............................... 134 6.1.1 Energy Harvester ................................ ................................ ................. 134 6.1. 2 Energy Storage ................................ ................................ .................... 137 6.2 System Prototype ................................ ................................ ........................... 137 6.3 System Demonstration ................................ ................................ ................... 139 6. 3.1 Measurement Method ................................ ................................ .......... 139 6.3.2 Delivered Energy ................................ ................................ ................. 140 6.3. 3 Average Power ................................ ................................ .................... 140 6.4 Summary ................................ ................................ ................................ ........ 142

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7 7 CONCLUSIONS AND FUTURE DIRECTIONS ................................ .................... 144 7.1 Research Contributions ................................ ................................ .................. 144 7.2 Summary of Research ................................ ................................ .................... 144 7.3 Future Directions ................................ ................................ ............................ 146 APPENDIX A : PUBLICATIONS ................................ ................................ ................................ ...... 149 B : CHIP BONDING DIAGRAM ................................ ................................ .................... 150 LIST OF REFERENCES ................................ ................................ ............................. 153 BIOGRAPHICAL SKETCH ................................ ................................ .......................... 160

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8 LIST OF TABLES Table 1 1 Summary of three conversion mechanisms. ................................ ....................... 26 1 2. State of the art low voltage energy harvesting circuits. ................................ ...... 39 3 1 Transistor size of the error comparator ................................ ............................... 82 3 2 Transistor size of the level shifter ................................ ................................ ....... 84 3 3 Transistor size of the current starved logic gates ................................ ............... 86 3 4 List of discrete components in the system prototype ................................ .......... 88 5 1 Variables and elements in energy domain conversion. ................................ .... 105 5 2 List of extracted parameters of the electrodynamic harvester .......................... 122 5 3 List of parameters of the boost converter ................................ ......................... 127 6 1 Measured average power delivered to the battery during human movements 142

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9 LIST OF FIGURES Figure 1 1 Parallel plate capacitor used in electrostatic transducers. ................................ .. 21 1 2 Electrostatic energy harvester topologies. ................................ .......................... 22 1 3 Electrostatic energy harvester topology: Out of plane gap. ................................ 22 1 4 Direct Piezoelectric effect ................................ ................................ ................... 23 1 5 A typical piezoelectric transducer with cantilevered beam structure ................... 23 1 6 Illustration of motional induction ................................ ................................ ......... 24 1 7 Prototyp e of an electrodynamic transducer. ................................ ....................... 25 1 8 A typical energy harvesting interface circuit. ................................ ....................... 27 1 9 Conventional full wave bridge rectifier ................................ ................................ 30 1 10 Active diode ................................ ................................ ................................ ........ 31 1 11 Two stage rectifier reported in [36] ................................ ................................ ..... 32 1 12 Full wave active rectifier topology reported in [37] ................................ .............. 33 1 13 Voltage doubler presented in [38] ................................ ................................ ....... 33 1 14 Ring oscillator with floating gate PMOS reported in [ 40]. ................................ .... 35 1 15 Tree topology charge pump reported in [41]. ................................ ...................... 35 1 16 Feedback and feedforward PWM dc/dc converter presented in [23]. ................. 36 1 17 Boost converter diagram reported in [45]. ................................ .......................... 37 1 18 Direct ac/dc converter reported in [47]. ................................ ............................... 38 1 19 Block diagram of conventional energy harvesting circuits ................................ .. 41 1 20 Block diagram of the input powered energy harvesting circuits .......................... 41 1 21 Block diagram of an energy harvesting system. ................................ ................. 42 1 22 Energy harvesting system reported in [56]. ................................ ........................ 44

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10 1 23 Energy harvesting system reported in [23] ................................ ......................... 44 1 24 Energy harvesting system reported in [55]. ................................ ........................ 45 2 1 Schematic of the input powered half wave ac/dc converter ............................... 49 2 2 Schematic of the active diode ................................ ................................ ............. 50 2 3 Schematic of the input powered comparator ................................ ...................... 51 2 4 Layout of the half wave ac/dc converter ................................ ............................. 52 2 5 Measurement result of the half wave ac/dc converter ................................ ........ 53 2 6 Test setup of input powered ac/dc converter ................................ ...................... 54 2 7 Power efficiency of the half wave ac/dc converter with 1.5 V, 20 Hz sine wave input. ................................ ................................ ................................ ......... 54 2 8 Circuit diagram of the input powered active ac/dc converter .............................. 55 2 9 Schematic of full wave rectifier ................................ ................................ ........... 56 2 10 Chip photo and micrograph of the ac/dc converter chip ................................ ..... 58 2 11 Open load experimental result when input is a sine wave with 1V amplitude: .... 58 2 12 Output power of the full wave ac/dc converter at various input voltage amplitudes (20 Hz) and load resistances ................................ ............................ 59 2 13 Experimental measurements and simulation predictions of power efficiency of ac/dc converter for 1V amplitude, 20 Hz input sine wave ............................... 60 2 14 Functional test of the circuit using a vibration electrodynamic harvester ................................ ................................ ....... 60 2 15 Voltage doubler based on active diodes ................................ ............................. 62 2 16 Schematic of negative side comparator ................................ ............................. 62 2 17 Voltage doubler with input powered scheme ................................ ...................... 63 2 18 Microphotograph of voltage doubler chipset ................................ ....................... 64 2 19 Photo of the voltage doubler chipset in SOIC16 package. ................................ 65 2 20 Measurement result of minimum input voltage ................................ ................... 66 2 21 Measured output voltage versus load resistances ................................ .............. 67

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11 2 22 Measured output p ower versus load resistances ................................ ................ 67 2 23 Experimental measurements and simulation predictions of power efficiency with a sine wave of 20 Hz frequency and 1.5 V voltage amplitude ..................... 69 2 24 Test result with a vibrational energy harvester ................................ ................... 69 2 25 Simulation result of voltage drop comparison between passive and active half wave ac/dc converters at different ac input amplitudes ............................... 71 2 25 Output power comparison of full wave, half wave and voltage doubling ac/dc converters. ................................ ................................ ................................ .......... 72 2 26 Power efficiency comparison of full wave, half wave and voltage doubling ac/dc converters with the input amplitude of 1.5 V ................................ ............. 73 3 1 Basic schematic of a boost converter ................................ ................................ 76 3 2 Boost converter circuit during two operating intervals ................................ ........ 77 3 3 Basic boost converter with a dc/dc controller ................................ ...................... 77 3 4 An example of PSM control scheme. ................................ ................................ .. 78 3 5 Block diagram of the input powered boost converter ................................ .......... 80 3 6 Schematic of the error comparator ................................ ................................ ..... 81 3 7 ................................ .............................. 82 3 8 Schematic of the level shifter ................................ ................................ .............. 83 3 9 Schematic of the voltage controlled oscillator ................................ ..................... 85 3 10 Current starved logic gates in th e ring oscillator ................................ ................. 85 3 11 Schematic of the voltage buffer ................................ ................................ .......... 86 3 12 Micrograph of the dc/dc controller die ................................ ................................ 87 3 13 Photo of the packaged dc/dc controller chip ................................ ........................ 87 3 14 Screenshot of input voltage, inductor current, switching signal and output voltage of the boost converter ................................ ................................ ............ 90 3 15 Simulation and measurement result of VCO output frequency ........................... 91 3 16 Measured power efficiency of the boost converter at different loads for regulated 3 V dc output ................................ ................................ ...................... 92

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12 4 1 Block diagram of the complete input powered interface circuit ........................... 95 4 2 Measurement setup of the complete input powered interface circuit .................. 97 4 3 sine wave ................................ ................................ ................................ ........... 98 4 4 Circuit start up process at open load when input is a 20 Hz, 1.2 V amplitude sine wave ................................ ................................ ................................ ........... 98 4 5 Power delivered to 3.7 V CV load at different input voltage amplitude ............... 99 4 6 Power efficiency of inte rface circuit vs. input voltage amplitude for regulated 3.7 V dc output ................................ ................................ ................................ 100 5 1 Schematic of a resonant electrodynamic energy harvester .............................. 103 5 2 LEM of the electrodynamic energy harvester. ................................ .................. 106 5 3 LEM reflected into electrical domain. ................................ ................................ 108 5 4 Thvenin equivalent circuit of electrodynamic harvesters. ............................... 108 5 5 A half wave ac/dc converter ................................ ................................ ............. 110 5 6 Equivalent circuit model of the half wave ac/dc converter ................................ 110 5 7 Equivalent circuit model of the voltage doubling ac/dc c onverter ..................... 111 5 8 PSM boost converter ................................ ................................ ........................ 112 5 9 Boost converter circuit ................................ ................................ ...................... 113 5 10 Equivalent circuit model of the boost converter. ................................ ............... 113 5 11 Simplified Equivalent circuit model of the boost converter. ............................... 114 5 12 Photos of the resonant electrodynamic energy harvester prototype ................. 115 5 13 LEM model of electrodynamic energy harvester ................................ ............... 115 5 14 Test setup for spring constant ................................ ................................ .......... 116 5 15 Measured mechanical force at different displacement ................................ ...... 117 5 16 Test setup of the transduction coefficient. ................................ ........................ 118 5 17 Measured mechanical force at different dc current ................................ ........... 118

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13 5 18 Resulting transient waveform from the flicker test ................................ ............ 120 5 19 Measurement setup of output voltage v ersus input acceleration amplitude ..... 123 5 20 Open circuit harvester output voltage versus acceleration amplitude ............... 124 5 21 Output power versus load resistance ................................ ............................... 124 5 22 Open load harvester output voltage versus acceleration frequency at 1.5 g .... 125 5 23 Parameter extraction of PMOS turn on resistance R D ................................ ...... 125 5 24 Interface circuit output voltage for model validation ................................ .......... 126 5 25 Parameter extraction of NMOS turn on resistance R on ................................ ..... 127 5 26 Model simulation and measurement result of the dc/dc converter .................... 1 27 5 27 Complete energy harvesting system model ................................ ...................... 129 5 28 Comparison of measurement and model simulation result of entire system at various load ................................ ................................ ................................ ...... 130 5 2 9 Comparison of measurement and model simulation result of entire system at various input acceleration amplitude ................................ ................................ 132 6 1 Block diagram of the self sufficient energy harvesting system ......................... 134 6 2 Non resonant electrodynamic energy harvester ................................ ............... 136 6 3 Example open circuit output voltage waveform when hand shaking the harvester ................................ ................................ ................................ .......... 136 6 4 Photograph of the system prototype ................................ ................................ 138 6 5 Photograph of the double sid ed circuit PCB boards ................................ ......... 138 6 6 ......................... 140 6 7 Energy delivered to the battery ................................ ................................ ......... 141

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14 LIST OF ABBREVIATION S Term Definition CC Constant current CCM Continuous conduction mode CR Constant resistor CV Constant voltage DCM Discontinuous conduction mode EH Energy harvesting E MF Electromotive Force LEM Lumped element model MPPT Maximum power point tracking PSM Pulse skip modulation PWM Pulse width modulation PFM Pulse frequency modulation VCO Voltage controlled oscillator

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15 ABSTRACT Abstract of Dissertation Presented to the Graduate School of the University of Florida in Partial Fulfillment of the Requirements for the Degree of Doctor of Philosophy INPUT POWERED INTERFACE CIRCUITS FOR ELECTRODYNAM IC VIBRATIONAL ENERGY HARVEST I NG SYSTE M S By Y uan Rao August 2013 Chair: David P. Arnold Major: Electrical and Computer Engineering Vibrational e nergy harvesting systems that convert ambient mechanical energy in the environment to usable electrical energy represent a promising emerging technology to achieve autonomous, self renewable, and maintenance free operation of wireless electronic devices and systems. Typica l energy harvesting systems are composed of three components : an energy harvester that converts the mechanical vibrations into electrical energy, an interface circuit that condition s and regulates the energy, and an electronic load that uses or stores the harvested energy. This dissertation specifically focuses on the development and experimental characterization of input powered energy harvesting circuits, including ac/dc converters and a dc/dc converter, for electrodynamic vibrational energy harvesters. T his input powered feature allows the active interface circuit ry to automatically enter a zero power consumption standby mode when the voltage from the harvester is below a threshold level, thus eliminating any energy drain between energy harvesting cycles. Implemented in a 0.5 m CMOS technology, the interface circuit is bench top characterized with a

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16 sine wave signal generator and also with real vibrational energy harvester s The measurement result shows that the minimum input threshold voltage is 1 V at o pen load. When the ac input amplitude is 2. 6 V and regulated dc output is 3.7 V the interface circuit can achieve a peak net efficiency of 61 % with 16.7 mW of output power delivered A simplified equivalent circuit model for a resonant type electrodynamic energy harvesting system is developed including a lumped element model (LEM) for the resonant harvester, a simplified interface circuit model, and a load model The overall system model is validated via comparison of circuit simulation s wit h experimental measurement s Lastly, a complete and fully self sufficient energy harvesting system is demonstrated using the input powered interface circuit and a non resonant electrodynamic harvester, designed specifically for harvesting energy from human movements. Tested under normal human activities (walking, jogging, cycling) the 70 cm 3 system is shown to charge a 3.7 V rechargeable battery with an average power of 234 jogging

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17 CHAPTER 1 INTRODUCTION 1 Over the past decades, the reduction in size and power consumption of CMOS circuitry has led to the proliferation of small handheld electronic devices that allow consumers to entertain, communicate and co mpute wirelessly With size reduction and technology integration, many compact, wireless sensor network technologies have been proposed and investigated Electrochemical batteries typically power these wireless devices. Batteries are th e earliest and most convenient power solution for wireless devices because they provide a relatively constant and stable dc voltage However, batteries present some disadvantages. Compared with advanced integrated electronic components, their relatively la rge size and weight create a bottleneck for the whole system. Moreover, periodic replacement or recharging of depleted batteries is not convenient as the number of devices is increased. Access to the batteries may be difficult or even impossible, for examp le in biomedical devices implanted in human body or sensors located in extreme environments. Therefore how to achieve autonomous, self renewable, and maintenance free operation while avoiding battery replacement or recharge has become an increasingly impor tant topic in the development of modern wireless systems. One promising solution to this problem is to build self powered systems using energy harvesting techniques. Energy harvesting is a relatively new area, and has attracted much attention for applicat ion in biomedical devices, sensors, industrial and 1 Sections of text, figures, and tables of this dissertation may be reproduced from the publications listed in APPENDIX A

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18 from the environment, such as vibration, sol ar, wind or temperature gradients, and convert ing them into electrical charge [1] [2] [3] Although the energy harvesting solution adds some complexity to the implementation of a system, it provides the potential benefit of making the system completely self sustaining for its entire lifetime. Therefore the lifetime of the device only depends on the reliability of its own parts and the availability of the ambient energy, not necessarily the batteries. While there is a great effort underway for developing of the energy harvesting transducers ( commonly called energy harvesters ), this work focuses on the design and implementation of the interface circuits fo r vibrational energy harvesting systems, and in particular for electrodynamic transducers. Electrodynamic vibrational energy harvesters generate voltage and current based on the relative motion between permanent magnets and coils. Functional front end low voltage low power interface circuits are designed, fabricated, and characterized. Two complete energy harvesting system s one resonant and one non resona n t are also built and characterized with real electrodynamic vibrational energy harvesters under mechani cal excitation The unique aspect that differentiates this work from other investigations is that the active electronic interface circuits are completely powered by the ac voltage from the harvester, rather than a dc voltage from a load side energy reservo ir (capacitor or the motivations and benefits of this input powered approach are detailed in subsequent sections.

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19 This remainder of this chapter is organized as follows: In S ection 1.1 the operating principles of three different types of vibrational energy harvesters are briefly introduced. Previously reported vibrational energy harvesting interface circuits and design challenges are described in S ection 1.2 Section 1.3 introduces the state of the art energy harvesting systems, with a special focus on electrodynamic harvesting systems. In Section 1.4 the research objectives are presented, followed by an outline of the dissertation organization. 1.1 Vibrational Energy Harvesters There are many kinds of energy sources in the environment, but in this work the scope is limited to studying low level vibrations as a power source. Such vibrations are ubiquitous in human motion, automobiles, aircraft, ships, trains, industrial environmen ts, etc [4] [5] [6] For example, watches can be powered using the kinetic energy of a swinging arm [5] Additionally, p iezoelectric shoe in serts have been used to power wireless transceivers when a person walks [6] The reader is directed to the following references for extensive overviews and reviews on the subject of vibrational energy harvesting: [6] [7] [8] Typically three methods are used to convert mechanical vibrations to electrical signals: electrostatic ( also ( also They commonly use resonant mass spring damper architecture s, which ha ve the benefit of enabling large mechanical responses at resonance and therefore allow the system to mechanically The basic principle and merits of each transduction method will be described in the re mainder of this section.

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20 1.1.1 Electrostatic Vibrational Energy Harvester An electrostatic vibrational energy harvester converts mechani cal vibrations to electrical power through a variable capacitor. A simple example is shown in Figure 1 1 The two conductors are separated by a dielectric, and theref ore the transducer acts like a variable capacitor. There are two operating modes of electrostatic transducer: constant voltage mode and constant charge mode. In constant voltage mode, an electrical circuit is normally required to provide initial charge an d maintain constant voltage during the conversion. The applied voltage generates initial charges on the two plates, and therefore an electrostatic force exists between them. W hen the converter is driven by vibrations, the relative movement of the plates le ads to a capacitance change by either a change of the overlap area ( ) or distance between two plates ( ) thereby makes the applied voltage change in proportion to the ampli tude of the electrode's motion, converting the mechanical energy to electrical energy As the name implies, constant charge mode is realized by maintaining a fixed amount of charge on the two plates during conversion. In this mode, a control circuit and an external source are required to establish an initial charge. Then the extern al source is removed before the capacitor is changed by motion, so that the charge stored in the capacitor is fixed. charges are physically implanted into one or both plates. When the two plates are separated due to external vibrations, the harvested energy is stored in the electric field as work has been done against the electrostatic force, so that mechanical energy is turned into electricity.

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21 Figure 1 1 Parallel plate capacitor used in electrostatic transducers. The most attractive advantage of electrostatic transducer is the compatibility with tradit i o nal microelectronic materials and microfabrication processes, and hence t he potential for monolithic integration with integrated circuits [1] [9] [10] Compared with electrodynamic and piezoelectric energy harvesters, electrostatic energy harvesters can be easily micromachined using MEMS techniques. Another advantage is the comparatively high voltage output (volts to hundreds of volts) compared with ele ctrodynamic harvesters. However, the primary disadvantage of electrostatic transducers is that a separate voltage source is needed to provide the bias field. Although electret based harvesters eliminate the need for external sources, they often have limite d charge lifetimes [11] [12] [13] [14] Another disadvantage for electrostatic harvesters is that mechanical limit stops are always required to avoid contact of the capacitor electrodes and short the circuit, which may cause reliability problems and additional mechanical damping.

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22 (A) (B) Figure 1 2 Electrostatic energy harvester topologies: ( A ) In plane overlap (B) In plane gap. Figure 1 3 Electrostatic energy harvester topology: Out of plane gap. 1.1.2 Piezoelectric Vibrational Energy Harvester Another widely reported type of vibrational energy harvesters utilizes a piezoelectric material, which develop s a n electric potential across its boundaries in response to a mechanical stress, or vice versa [15] [16] [17] [18] [19] As shown in Figure 1 4 when a p iezoelectric material is mechanically strained, either in compression or tension, it will create a voltage due to its so called direct piezoelectric effect. Conversely, when a voltage is applied to a piezoelectric material, a mechanical strain will be indu ced, which is called inverse piezoelectric effect. The direct piezoelectric effect is used in vibrational energy harvesters to convert mechanical vibration energy to electrical energy.

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23 (a) (B) (C) Figure 1 4 Direct Piezoelectric effect (A) Equilibrium (B) Compression (C) Tension Figure 1 5 shows a typical cantilever beam type piezoelectric harvester structure, where a proof mass is placed on the free end of a cantilever Under an external vibration, the beam bends, and the piezoelectric layer is alternately subjected to tension and compression due to the mechanical motion These stress generate an alternating output voltage across the electrodes. Figure 1 5 A typical piezoelectric transducer with cantilevered beam structure Similar to electrostatic transducer, one adva ntage of the piezoelectric transducer is its ability to generate high voltages (volts to tens of volts) which makes the interface circuit design much easier. Another advantage is that the piezoelectric transducer e during operation. However, from a miniaturization and integration standpoint, piezoelectric devices are more difficult to

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24 implement on the microscale or to integrate with microelectronics, because of fabrication difficulties of microfabricating high qual ity piezoelectric materials 1.1.3 Electrodynamic Vibrational Energy Harvester variation of the magnetic flux through a coil induces electromotive force (EMF) or voltage The variation of the magnetic flux can be either caused by relative motion of the coil to a fixed magnetic field (motional induction), the change of the magnetic field (transformer induction), or both. An electrodynamic transducer utilizes motional induction for mechanical to electrical energy conversion As shown in Figure 1 6 when there is relative movement between the coil and the magnet, V emf is induced due to the magnetic flux change. If an electrical load is connected, electrical current will flow in the direction governed by Figure 1 6 Illustration of motional induction An example electrodynamic harvester architecture is shown in Figure 1 7 which is a reson ant mass spring damper system. There is a magnet and a coil moving with respect to each other according to the outside vibrations. The magnetic flux in the coil is time varying and therefore introduces a voltage, which generates current when an

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25 electrical load is connected. As a result, the mechanical power from vibration sources is converted to the electrical power delivered to the load. Figure 1 7 Prototype of an electrodynamic transducer. Electrodynamic transduction has attracted increasing research attention [20] [21] [22] [23] [24] [25] There are a variety of modern high performance permanent magnetic materials available that can provide high magnetic fields and thus good coupling between the mechanical domain and electrical domain. Another benefit of th e electrodynamic transducer compared with electrostatic transducer is that no separate voltage supply is required. However, one main challenge of electrodynamic energy harvesting is the small output voltage level (few hundreds of mV to several volts) due to the limited practical size, which makes the power electronic circuit design more difficult. How to make efficient interface circuit s at these low voltage levels becomes a serious challenge which will be addressed in detail in Chapter 2. Another challen ge is to achieve high performance micro magnets that can be integrated into small scale systems [26] Magnetic thin films tend to provide poorer magnetic properties, while bulk magnetic materials with superior performance are d iff icult to integrate on silicon. The advantages and disadvantages of the discussed transducers are summarized in Table 1 1

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26 Table 1 1 Summary of three conversion mechanisms. Type Advantage Disadvantage Electrostatic Easy to integrate with standard CMOS process External voltage source for startup Mechanical stops required High output impedance Piezoelectric No external voltage source for startup Difficult to integrate with standard CMOS process High output impedance Electrodynamic No external voltage source for startup Low output voltage Difficult to integrate with standard CMOS process 1.2 Vibrational Energy Harvesting Interface Circuits Power electronic i nterface circuits are often required, because the output voltage and current from the energy harvester are rarely compatible with the load. As the power processing stage of the energy harve sting system, t he interface circuit is desired to extract the maximum power from the harvester and also transfer the maximum power to the load The circuit should also provide a regulated output voltage to be compatible with the load A typical vibrational energy harvesting circuit consists of a rectifier (ac/dc converter) and a step up converter (dc/dc converter), as shown in Figure 1 8 The output from the transducer is typically an ac voltage, but e lectronic loads almost always require a dc voltage for their operation thereby the ac/dc converter is used to convert the ac voltage to a dc voltage. When the dc voltage is not high enough for load operation, an additional step up dc/dc converter (boost converter) is added to boost it to a higher level. For most cases, the load requires a fixed dc voltage at various input

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27 voltage and load current, a control circuit is therefore designed to achie ve output voltage regulation through a feedback loop Figure 1 8 A typical energy harvesting interface circuit 1.2.1 Circuit Design Challenges Since electrodynamic vibrational energy harvesters generally produce relatively low ac voltage (tens of mV to several volts) and low output power (hundreds of W to several mW) [7] [27] specialized power electronic circuits are necessary. Moreover, t he power m anagement circuits must be able to efficiently function with challenging operational conditions such as low input voltages, intermit tency of available power, and small physical size, which are further discussed below. One of the challenges is the low inpu t voltage level, which makes circuit design difficult, especially for the front end a c/dc converter as it interface s with the harvester directly. Considering the low output voltage of vibration harvesters, conventional diode based rectifiers are often not practical because of their forward bias diode voltage drop (hundreds of millivolts) Another option is to use a transformer to increase the ac voltage before the diode bridge. However this is also problematic because the low operational

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28 frequencies (<< 1 kHz) would require a very bulky transformer that would be massive compared to the size of a typical energy harvester. Another major challenge is the stand alone operation. Energy harvesting circuits must be able to operate independently from the other com ponents in the application because the access to external power is not guaranteed. Most reported energy harvesting circuits are self powered by their load side energy storage elements. However, in real applications, the ambient vibrational energy may be ve ry intermittent. When the harvester is not vibrating, the standby power draw of the circuit may drain the energy reservoirs. In this case, an additional startup circuit or a backup energy storage element is required to wake up the interface circuit after l ong periods of inactivity from the harvester, which increases the power, complexity, and size of the overall circuit The third challenge for the ener gy harvesting interface circuit is the desire to achieve high performance but in a small form factor for size compatibility with compact energy harvesting systems. The overall circuit should be physically small, so that it performance interface circuits becomes difficult due to the low output voltage and power from small size harvesters. While integrated circuits can be made quite small, the low voltages and low operating frequencies found in energy harvesting systems often require the use of large value and large sized passives. For example, several commercial energy harvesting products are available [28] [29] [30] but their large volume s ( hundreds of cm 3 ) are not suitable for size limited applications. 1.2.2 Literature Review Interface circuit s play an important role in energy harvesting systems They are designed to provide efficient power conditioning and storage of the incoming energy,

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29 while consuming as small size and quiescent power as possible. A number of papers have been published [31] [32] on energy harvesting circuits (for all types of environmental energy) Therefore, i t prudent to limit the scope of the literature review to the circuit s that are dedicated to vibration to electrical energy conversion considering the circuit requirement may vary significantly among different types of input energy Furthermore, because of the substantial differences in voltage levels and output impedances between electrodynamic vibrational harvesters and both electrostatic and piezoelectric vibrational harvesters, these different transducers demand different circuit architectures. The primary focus of this work is targeting electrodynamic vibrational energy harvesters, which present the most challenging low voltage levels. Note that although some circuit architecture s or blocks have been reported for piezoelectric h arve s ters or other applications such as RFID s certain concepts or architectures can be adapted to electrodynamic vibrational harvester circuit s as well. Based on the circuit functionality the following review is categorized into to three parts: 1) ac/dc rectification 2) dc/dc voltage conversion 3) direct ac/dc voltage conversion 1.2.2.1 Ac/dc Rectification Passive Ac/dc Rectification The most common and simple ac/dc rectification circuit is the full wave bridge rectifier which consist of four passive p/n diodes as shown in Figure 1 9 To be implemented on silicon, the passive diodes can be replaced by diode connected transistors [33] or cross coupled MOSFETs [34] However, the input operating voltage s of these MOSFET based full wave rectifiers are low limited by the threshold voltages of

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30 the transistors. Moreover, the conversion efficiency is also sensit ive to the threshold voltage, because the voltage drop across the transistors will increase significantly when they are not fully turned on. Some techniques, such as floating gate [35] and boot strapping [1 26], have been repor ted to solve the bottleneck caused by the transistor threshold voltage at the expense of additional fabrication or circuit cost. In comparison, Schottky diodes can offer relatively stable forward bias voltage drop as low as around 0.2 V. Low voltage Schottky diodes normally suffer from high reverse current but it is not fatal to low voltage low power energy harvesting circuits. However, 0.2 V voltage drop still leads to a t least 20% power loss assuming the input amplitude is 1 V. Addi tionally, Schottky diode s are not compatible with CMOS fabrication process easily which increases the size and cost of the circuit. Figure 1 9 Conventional full wave bridge rectifier Active Ac/dc Rectification Another option is to replace the standard diodes with active diodes. An active diode can be constructed to mimic the behavior of an ideal diode to overcome t he forward bias voltage drop discussed before. As shown in Figure 1 10 an active diode consists of a comparator and a switch. The two terminals of the switch are equivalent to the anode and cathode terminals of a diode. The c omparator monitors the voltage

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31 between the two terminals. When the anode voltage is higher than the cathode, the comparator outputs turns on the switch. Otherwise the switch is turned off. Figure 1 10 Active diode A ctive diode based ac/dc rectifiers (or called synchronous rectifiers) are widely reported [1 28 32 41, 42 shuo ] to achieve higher efficiency by reducing the conduction loss The active diode approach potentially enables very low turn on voltages and low reverse leakage characteristics. While the comparator requires some external power, this power consumption is usually quite low depending on the current through the switch device (i.e. MO SFET) Since the on resistance of the switch MOSFET is usually much smaller than the equivalent resistance of a passive diode the active diode can provide a more efficient rectification. However, active diodes suffer from one major drawback, that is exte rnal power supply is required for the comparator. In an energy harvesting system, this requires a continuous supply of energy as well as a mechanism for self starting (boot strapping) from a completely discharged state. Fortunately, the power consumption i s quite low; commercial vendors, and even lower power comparators may be designed by eliminating unnecessary functions found on commercial ICs.

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32 Peter s et al. designed a two stage a c/dc converter using a MOSFET based active bridge rectifier followed by an active diode to control the direction of current flow, as shown in Figure 1 11 [36] The resulting conduction voltage drop was about 10 times lower than that of a conventional diode bridge, and the overall efficiency was > 95%. However, these results were based on testing results under light load conditions (load than twice the MOS threshold voltage in order to turn on the MOSFETs in the first stage (1.25 V in their work). Figure 1 11 Two stage rectifier reported in [36] ( Copyright 2010 IEEE ) Lam et al. presented a cross coupled structure that reduced the minimum input voltage to 500 mV [37] Active diodes were used to replace the bottom two MOSFETs of the previous synchronized rectifier, as shown in Figure 1 12 Since the entire input voltage was exerted across the gate and source terminals, the resulting minimum input voltage was reduced to the threshold voltage of the MOSFET (0.5 0.7 V). Efficienc y with input voltages of 1.5 V or greater was reported to be 60 90%. Cheng et al. designed an active diode based voltage doubler [38] for vibrational energy harvester, where the output voltage is twice the input amplitude, as shown in Figure 1 13 Implemented by discrete components, the circuit is a ble to work at input as

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33 low as 5 mV, but requires external power supply for the comparators in the active diodes. The measured power efficiency is above 80% for input amplitude of 250 mV or higher. Figure 1 12 Full wave active rectifier topology reported in [37] ( Copyright 2006 IEEE ) Figure 1 13 Voltage doubler presented in [38] 1.2.2.2 Dc/dc Voltage Conversion To provide a dc voltage that is compatible with the load electronics, a dc/dc voltage converter is often required in addition the ac/dc rectification circuit. The simplest and most old fashioned dc/dc converter is linear regulator [39] However, the linear

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34 regulator is not favorable for vibrational energy harvesting systems, because it can only provide the output voltage that is lower than the input. Furthermore, a li near regulator suffers from poor efficiency when there is a large voltage drop between the input and the output To achieve regulated output while maintaining decent efficiency switch mode dc/dc converters are widely used Most of the reported dc/dc volta ge converter in energy harvesting applications belong s to one of two categories: switch mode dc/dc converters with capacitor (or called charge pumps ) a nd switch mode dc/dc converters with inductor as presented below. Capacitor Based Switch mode Dc/dc Conv erters Switched capacitor Dc/dc converters ( commonly called charge pumps ) have been reported to provide voltage regulation and conditioning in energy harvesting syste ms [1 77 82]. However the application is limited in low voltage low power vibrational sy stems, because it has relatively large output impedance and requires complicated control circuits. Meanwhile, using on chip capacitors can save significant area, but the voltage regulation accuracy is poor due to the large parasitic capacitances, which can be as large as 50% of the useful capacitance [1 48]. Delgado et al. designed a 300 mV energy harvesting system using 0.5 m CMOS technology [40] The circuit consists of a low voltage charge pump and an oscillator. The low voltage operation is achieved by programming the floating gate of the transistors, so that the transistor threshold is reduced. The oscillator is programmable to allow the chang e on the operating frequency, as shown in Figure 1 14 [40] However, programming gate requires special process Fowler Nordheim tunneling an d hot electron injection which makes it not attractive in cost constraint applications.

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35 Furthermore, the charge transfer of charge pump circuit in ultra low voltage applications has relatively low efficiency. Figure 1 14 Ring oscillator with floating gate PMOS reported in [40] ( Copyright 2010 IEEE ) Lu et al. improved the charge transfer capability of a charge pump with a new tree topology, as shown in Figure 1 15 [41] The front end stage operates exactly like an ac/dc voltage doubler, while the back end stage uses the boosted voltage to output an even higher voltage. The tradeoff is that to minimize the capacitor array size, the operating frequency must be very high (3 MHz to 100 MHz), which dramatically increases the switching loss of the circuit and also introduces electromagnetic inter ference (EMI) to the system. Figure 1 15 Tree topology charge pump reported in [41] ( Copyright 2010 IEEE )

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36 Inductor Based Switch mode Dc/dc Converters Switch mode dc/dc converter using inductor is one of the most popular dc/dc conversion topologies in energy harvesting systems for its high efficiency, accurate regulation and low power consumption [42] [43] [44] [45] [46] [42] The output voltage of the switch mode dc/dc converter is regulated through dynamically charging and discharg ing an inductor. Since high Q on chip inductor design is still challenging, the requirement of using a discrete inductor adds both cost and size of the circuit. Cao e t al present ed a feedforward and feedback dc/dc converter for vibration al energy harvesti ng system, as shown in Figure 1 16 [23] The circuit adjusts the duty cycle of switching pulses based on both the input voltage through a feedforward loop, and the output voltage through a feedback loop. Fabricated in a 0.35 circuit is reported to have better adjustability than conventional PWM converter with only feedback loop especially at large input voltage. Figure 1 16 Feedback and feedforward PWM dc/dc converter presented in [23] ( Copyright 2007 IEEE )

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37 Carlson et al. [45] demonstrated a circuit that can boost dc input voltage from 20 mV. A novel control circuit is designed to use peak current regulation and achieve efficient operation in this low input voltage range, as shown in Figure 1 17 The output regulated through a controlled switching pulse with variable frequency and 50% duty cycle. The low power consumption is achieved by a one shot pulse design for switching MOS (M2) for powe r saving With input range of 50 250 mV at 10 60 70% efficiency. However, this circuit requires 600 mV supply to cold start from sleep mode and drive the switches. Figure 1 17 Boost converter diagram reported in [45] ( Copyright 2010 IEEE ) 1.2.2.3 Direct Ac/dc Voltage Conversion In direct ac/dc voltage converter s [47] [48] there is no separate ac/dc rectification, that is both the ac/dc and the dc/dc functionalities are realized simultaneously. The idea is based on the alternating operation of the two switch mode

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38 dc/dc converters at the negative half and the positive half of the ac input. They have the benefit of lower input voltages and relatively high efficiency due to lower circuit complexity. However, the circuit usually contains more than one inductor, leading to a larger volume and circuit cost Dwari et al. reported a 400 mV direct ac/dc voltage converter for electrodynamic harvesters without any bridge rectification, as shown in Figure 1 18 [47] The circuit consists of a boost converter in parallel with a buck boost converter, which are operated in the positive half cycle and negative half cycle, res pectively. The negative gain of the buck boost converter is used to boost the voltage of the negative half wave of the transducer output to positive dc voltage. This direct ac to dc converter avoids the conventional bridge rectification and therefore has higher efficiency. However, in this design, the coil of the magnetic harvester acts as the inductor of the power converters, so the circuit and harvester must be carefully co designed to maintain a good performance of the system. In addition, a self starti ng circuit using a battery is required for the proper function of the system. Figure 1 18 Direct ac/dc converter reported in [47] ( Copyright 2010 IEEE )

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39 As a general summary recent research es on energy harvesting interface circuits ha ve resulted in remarkable advances in power efficiency, scaling, and low voltage capabilities S tate of the art interface circuits reviewed are summarized in Table 1 2 covering ac/dc conversion, dc/dc conversion and direct ac/dc conversion Table 1 2 State of the art low voltage energy harvesting circuits. Ref Function Architecture Powered by Efficiency Minimum Input Process ( m) Cheng [38] Ac/dc Active diode based voltage doubler External Supply 80% @ 0.25V 5 mV Off chip Peters [36] Ac/dc Two stag es: Negative voltage converter and an active diode Output 90% @ 50 kHz 350 mV 0.35 Lam [37] Ac/dc cross coupled structure Output 60%~90% @ 1.5V 500 mV 0.35 Dwari [47] Direct Ac/dc Boost converter in parallel with a buck boost converter. External Supply 61% 400 mV Off chip Delgado [40] Dc/dc Floating gate programming and Charge pump Output N/A 300 mV 0.5 Lu [41] Dc/dc Charge pump with tree topology Output N/A 280 mV 0.065 Cao [23] Dc/dc Feedback and feedforward PWM Output NA NA 0.35 Carlson [45] Dc/dc Digitally control at DCM Output 52% 20mV 0.13 Based on the literature review the conclusion can be made that nearly all circuits designed for vibrational energy harvesting are either powered by an external power supply or the output. Additionally, there are very few papers on developing a complete interf ace circuit, including both ac/dc and dc/dc conversion stages, let alone a complete vibrational energy harvesting system.

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40 1.2.3 Input powered Energy Harvesting Circuit In previous literature review, most energy harvesting circuits use active components ; thereby requir e some source of power Using an external power supply has been reported [38] [47] but it is not practical for energy harvesting applications, because the energy harvesting system is anticipated to be the sole power source In other word s in end applications, external power is likely not available. Thus, in most state of the art energy harvesting circuit designs, output powered converters (also called self power ed converters) are used [22] [49] [45] [50] [51] [52] [53] [54] where the power is supplied from the output voltage. As shown in Figure 1 19 output powered converters are powered by load side energy storage elements such as super capacitors or rechargeable batteries. Output powered ac/dc converters have one major drawback for energy harvesting systems. That is they tend to consume power even when the system is not harvesting any en ergy. If there are short intervals between energy harvesting cycles, the system has sufficient energy in the storage element and will start up and function normally. This rom the energy harvester, these energy storage reservoirs will eventually be drained. In this case appropriate measures must be taken to ensure the entire energy harvesting ve circuitry may be required in the circuit design to provide this cold start functionality, and this passive circuitry may require higher input voltages. If such circuitry does not exist, the time between charging cycles must be short enough so that the e nergy storage element has sufficient energy to startup energy harvesting circuits Additionally, there

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41 must be sufficient power available on the load to maintain continuous operation of the ac/dc converter. Figure 1 19 Block diagram of conventional energy harvesting circuits To solve the problem of output powered energy harvesting circuits, an input powered energy harvesting circuit is shown in Figure 1 20 which consists of two sub circuits: an input powered ac/dc converter and an input powered dc/dc converter. Both sub circuits have input controlled standby mode and zero standby power. Therefore th e overall interface automatically turns on/off depending on the input voltage level and consumes power only when the input is high enough for harvesting. This feature eliminates the need for pre charging a load and allows for indefinitely long int ervals be tween charging cycles, thereby the circuit can cold start without any additional startup sub circuit required. Figure 1 20 Block diagram of the input powered energy harvesting circuits

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42 1.3 Vibrational Energy Harvesting Systems A typical energy harvesting system can be broken down into three key components: the transducer, the interface circuit, and the load as shown in Figure 1 21 The transducer is an energy conversion device that couples the energy from a source domain (mechanical, solar, thermal, etc.) to the electrical domain. The role of the interface circuit is to extract a maximum amount of ener gy from the energy harvester and make the energy usable to the load. This may include voltage rectification, voltage regulation, and other power management functions. The load may comprise power consuming electronic devices (circuits, sensors, actuators an d etc.) and/or energy storage elements (batteries, capacito rs, super capacitors and etc.). Figure 1 21 Block diagram of an energy harvesting system. 1.3.1 System Design Challenges To design a high performing energy harvesting syst em, all function blocks should be considered together as one cooperative system, because the performance degradation of any one component or any in compatibility of one component with another will make the s ystem less efficient. Sometimes the improvement of one component may come with a performance loss elsewhere For example, the interface circuit would benefit from higher input voltage and power for larger output power and higher efficiency. However, to ge nerate higher voltage and power, the transducer size

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43 might need to become large r and the power density of the whole system may be sacrificed. T he design of an energy harvesting system is very different than the design of a traditional battery powered system. Whereas battery powered systems are typically limited by energy, energy harvesting systems are typically limited by power (energy is theoretically limitless). Thus, maximizing the output power is the main goal of an energy harvesting system. energy the mechanical to electrical efficiency is not the primary concern. The output power however, is dependent on the size and power density of the energy harvest ing transducer and the efficiency of the associated interface circuitry, both of which should be optimized simultaneously. 1.3.2 Literature Review There now exist an enormous number of publications on energy harvest ers and energy harvesting circuit interfaces, but much fewer studies that combine harvesters and interface electronics to create fully functi oning energy harvesting systems [55] [56] [57] [58] Of those combined electromagnetic harvester/electronics systems, many have one or more shortcomings as discussed below Rahimi et al. reported an electromagnetic energy harvesting system, as shown in Figure 1 22 [56] The system, including an electromagnetic harvester and a full wave rectifier, is realized in a system on package with a small volume comparable to the size of a C Type battery. The harvester has two coils, one of which is used to power the interface circuit, leading to an unnecessary increase of the harvester size. Moreover, without any dc/dc circuit in the system, it cann ot provide a regulated output voltage and therefore are not readily compatible with modern electronic devices.

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4 4 Figure 1 22 Energy harvesting system reported in [56] ( Copyright 2012 IEEE ) A vibration power generator system [23] was proposed as shown in Figure 1 23 consisting of a n electro dynamic vibration power generator and an efficient interface circuit implemented on a minute printed circuit board (PCB). The circuit employs an ac/dc converter and a PWM boost converter. Although the output is regulated, the circuit is actually powered by the load. Therefore their stored charge can be easily depleted during states of non activity and an extra startup or bootstrapping procedure is often required. Figure 1 23 Energy harvesting system reported in [23] ( Copyright 2007 IEEE ) Another magnetic energy harvesting system is proposed in [55] to scavenge the low amplitude, low frequency, and non periodic vibrations on bridges. A frequency

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45 increased generation technique operating at 2 Hz is employed through a bi state low frequency mechanical structure. The circuit design, shown in Figure 1 24 however is powered by the output, and there is no voltage regulation in the system. Figure 1 24 Energy harvesting system reported in [55] ( Copyright 2009 IEEE ) 1.4 Research Objectives Based on the rapidly expanding literature involving energy harvesting systems, t he focus of this research is to explore the design and implementation of the power electronics interface for low voltage, stand alone vibrational energy harvesting systems that employ electrodynamic energy harvesters. The two primary reasons for focusing on just the electrodynamic harvester, as opposed to all transduction techniques, are as follows. First, some comparative studies argue that piezoelectric energy harvesters are more promising than their electrodynamic counterparts, because although the electrodynamic transduction may offer higher power densities, the output voltage of electrodynamic harvesters is typically too low to be efficiently rectified [59] This ongoing for this work. Second, the electrical characteristics of electrodynamic harvesters are very different than piezoelectric or electrostatic harvesters. The output impedance of the latter two are

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46 capacitive in nature and typically very high impedance. They t ypically output higher voltages and lower currents. Conversely, electrodynamic harvesters generally exhibit much lower net output impedance, and the impedance is often resistive in nature or in some cases modestly inductive. Electrodynamic harvesters typi cally output lower voltages and higher currents. As a result, it is anticipated that optimal interface circuits will require different approaches for these different classes of energy harvesting transducers. This research will focus on energy harvesting i nterface circuits with low input voltage range (1 V to 3 V) low power consumption and small physical size (tens of cm 3 ) designed for compatibility with electrodynamic energy harvesters. Previously discussed design challenges will be i nvestigated and new designs will be proposed to optimize the energy harvesting interface circuits for low level vibrations. The output power, power efficiency and minimum operable input voltage level will be examined for all the proposed designs. The optim ized interface circuits will be designed to harvest as much energy has possible from the harvester and transfer the gathered energy to energy storage elements with minimum power losses. Meanwhile, the circuits will be fully input powered to achieve zero st andby power when the harvester is not harvesting any energy. Note that the design and optimization of the vibrational energy harvester structure and/or the vibration source are explicitly outside the scope of this work. 1.5 Dissertation Organization The disse rtation consists of seven chapters. Chapter 1 introduces the background and the state of the art vibrational energy harvesting circuits and systems, as well as the goal and contribution of this research. Chapter 2 describes the design,

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47 implementation, and characterization of three input powered ac/dc converters. Chapter 3 demonstrates a closed loop input powered dc/dc converter. In Chapter 4, t he ac/dc converter and the dc/dc converter are combined to form a complete input powered interface circuit. Chapter 5 presents an energy harvesting system model, including a resonant electrodynamic transducer and the proposed interface circuit Chapter 6 demonstrates a fully functional, self sufficient energy harvesting system on real human movements utiliz ing a unique non resonant electrodynamic transducer and the interface circuit to charge a rechargeable battery. Finally, Chapter 7 summarizes the work and also describes the future direction s of this research.

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48 CHAPTER 2 INPUT POWERED AC/DC CONVERTERS In man y real world use cases, the power generation from a v ibrational energy harvester is intermittent T herefore when the harvester is not vibrating, the interface circuit standby power may drain the energy reservoirs and the whole system may no longer be able problem wherein active ac/dc converters are directly powered by the input ac signal generated from the energy harvester, instead of load sided energy storage elements in conventional en ergy harvesting circuits. This input powered approach eliminates the need for external power supplies, reduces the pin count of the circuit interface and avoids standby power consumption when the input amplitude is too low for energy harvesting. Three to pologies are commonly used in ac/dc converters for vibrational energy harvesting systems: the half wave bridge ac/dc converter, the full wave ac/dc converter, and the voltage doubling ac/dc converter (also called voltage doubler). In this chapter, the inpu t powered feature is implemented on these three ac/dc converter topologies. The design, fabrication, and experimental result are discussed thoroughly for each topology in the following three sections. In the final summary section, their performances are co mpared and the recommendation is given for electrodynamic vibrational energy harvesting applications. 2.1 Half wave Ac/dc Converter 2.1.1 Circuit Design A c onventional half wave ac/dc converter ( commonly called a peak detector) is composed of a diode and a capacitor. In this design, an active diode is used for its lower

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49 turn on voltage and lower reverse leakage current, as compared to a junction based diode [60] [61] [62] [63] Figure 2 1 shows the schematic of the half wave ac/dc converter using the active diode Th e comparator in the active diode is powered by its input (V rect ). Therefore the comparator consumes power only when the input is high enough for energy harvesting. The half wave converter operates only in the positive half cycle of the input waveform, and thus the dc output is the ac input amplitude (V IN ) of the positive cycle at open load. Figure 2 1 Schematic of the input powered half wave ac/dc converter The active diode comprises a comparator and a PMOS switch, as shown in Figure 2 2 The two terminals of the PMOS are equivalent to the anode and cathode terminals of a diode. The comparator detects the voltage difference between these two terminals and determines when to turn on or off the PMOS. When the anode voltage is higher than the cathode, the comparator output is low and PMOS switch is turned on to charge the loa d; otherwise the PMOS is turned off and the reverse current is blocked. T he MOSFET switch sizing has a large impact on the performance of the voltage doubler. A larger MOSFET has lower turn on resistance, and therefore can reduce the conduction loss. Howe ver, a large transistor increases both the silicon area and the

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50 gate capacitance. The increase of the gate capacitance makes the comparator design more complicated to maintain fast speed an d low power consumption. In the implementation here, the W/L ratio of the PMOS is 1500/1 in m based on the tradeoff. Note that the PMOS bulk is connected to the anode, so that the body diode helps pre charge the load before startup of the circuit. Also the parallel connection of the body diode and the active diode avoids conduction of any reverse current. Figure 2 2 Schematic of the active diode A simple and low power comparator is designed to control the PMOS switch in the active diode. Differing from previously reported comparators used in active diodes [60] the comparator here is directly powered by its negative inp ut (V ). As a result, when the input voltage is not sufficiently high, the comparator automatically turns off and consumes no power. The schematic of the two stage input powered comparator is shown in Figure 2 3 The first stage includes a differential transistor pair (M3, M4) and a latch (M1, M2). The differential pair tracks the differential voltage signal, and the latch amplifies the signal by positive feedback. Both the power supply (VDD) and the bias voltage (V bias ) ), which is the output of the full wave rectifier (V rect ). The second stage acts as an inverter. M6 and M8 are used to reverse and

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51 sharpen the output signal, and M7 is added to reduce the comparator offset. This stage is powered by the positive input (V+) rather than the negative input (V ), but the inverter only consumes dynamic power when switching. Therefore the second stage draws no static current from the positive input and does not consume any power when the converter is in standby mode. This is important because when used in an energy harvester system, the positive input (V+) to the comparator is also the ac/dc rectifier output (V out ). Figure 2 3 Schematic of the input powered comparator An external diode D1 is added here to block the leakage current from the substrate to the n well when the input is the negati ve half wave, because of the bidirectional output from the harvester. In comparison, the full wave ac/dc converter in the previous section does not need this diode, because the full wave rectifier stage has converted all the negative half waves into positi ve ones. In the half wave ac/dc converter, this additional diode will not induce any voltage drop of the dc output drop will impact the rail rail voltage headroom of th e comparator in the active diode

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52 T herefore a low forward voltage diode is preferred to minimize the input threshold of the converter. 2.1.2 Circuit Implementation The circuit was fabricated i n silicon by On Semi 3M 2P 0.5 m CMOS process and packaged in DIP40 package. Figure 2 4 shows the circuit layout where the active area for the chip is 0.012 mm 2 (196 m 62 m). A 220 F aluminum electrolytic capacitor (C1) and a NSR0320 Schottky diode [ 64] (D1) are used as the off chip discrete components. Figure 2 4 Layout of the half wave ac/dc converter 2.1.3 Measurement Result Using a 20 Hz input sine wave from Agilent 33120A function generator, the output voltage, output power and power efficiency of the half wave ac/dc converter can be characterized Figure 2 5 (A) gives the output voltage at different input amplitude when the load increasing load resistance bec ause the lower voltage drop of the PMOS due to the smaller load current. The measurement is up limited to 3 V input by the breakdown voltage of the fabrication process. Accordingly the output power at different input and load conditions is shown in Figure 2 5 (B) where the output power can be up to 8.2 mW

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53 at 3 V input and 500 load The maximum output power occurs when the load is around 500 (A) (B) Figure 2 5 Measurement result of the half wave ac/dc converter (A) Output voltage (B) Output power Measurement of the power efficiency is straightforward, because the input signal is the only power source. The efficiency is given by where V out,RMS is the RMS value of the output voltage, R load is the load resistance, V in (t) and I in (t) are the instantaneous input voltage and current, respectively, and T is the period of one dc cycle. The block diagram of the test setup is shown in Figure 2 6 A vibrational energy harvester or function generator provides an ac input source to the ac/dc converter, which generates a dc output by charging the output capacitance (C1). The input curren t waveform is measured with a Tektronix TCP312 current probe and TCP300 amplifier. The input voltage waveforms (V in ) and full wave rectifier output (V rect ) are displayed and characterized for time average input power using a Tektronix TDS5104B digital phos phor oscilloscope. The input ac current (I in ) is first sensed by TCP312 current probe and then amplified by TCP300 current probe amplifier, so that it

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54 can be displayed on TDS504B oscilloscope as well. A Fluke 189 multimeter is used to measure the rectifie d output voltage (V out ), so that the power efficiency can be estimated. Figure 2 6 Test setup of input powered ac/dc converter Figure 2 7 presents the measured power efficiency when the input is a 20 Hz sine wave with voltage amplitude of 1.5 V. The peak efficiency of 84% is measured Figure 2 7 Power efficiency of the half wave ac/dc converter with 1.5 V, 20 Hz sine wave input. The power efficiency becomes poor at light load because less power is delivered to the load compared with the load ind ependent power consumption of the circuit. For

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55 example, the power consumption of the comparator keeps almost constant at various loads. Consequently when the output current decreases, the percentage of the comparator power loss becomes dominant. 2.2 Full wave Ac/dc Converter 2.2.1 Circuit Design Figure 2 8 shows the circuit block diagram of a full wave ac/dc converter. It consists of two stages: a MOSFET based full wave rectifier stage and an active diode stage. The full wave rectifier converts the negative half waves of the ac input (V in ) from the harvester into positive ones (V rect ), which charge the load through the active diode to obtain a dc output (V out ). The active diode ensures current does not flow from the load back toward the source, since the active full wave rectifier has no inherent current rectification capability. Figure 2 8 Circuit diagram of the input powered active ac/dc converter The MOSFET based full wave r ectifier consists of four MOSFETs (M1 M4) in a bridge connection [38] as shown in Figure 2 9 M1 and M2 are PMOS, while M3 and M4 are NMOS. When V in is positive, M1 and M4 are conducting. Therefore the current flows from the input to the output through M1 and back to the input throug h M4.

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56 Otherwise when V in is negative, M2 and M3 are conducting. Thus the current flows from the input to the load through M2 and back to the input through M3. The MOSFETs are sized large enough to reduce turn on resistance, so that both the voltage drop and power consumption of the rectifier are minimized. However, care must be taken because large MOSFETs suffer from high leakage, low gate oxide breakdown voltage and large area. In this work, the W/L ratio of 750/1 in m is used for all the MOSFETs. Sinc e the MOSFETs are used as switches, the threshold voltages of these MOSFETs determine the startup voltage of the circuitry. Thus when the input is higher than their threshold voltage (0.6 V), the voltage drop is very small due to the low turn on resistance Figure 2 9 Schematic of full wave rectifier T he full wave rectifier stage cannot charge the load directly because the reverse current is not blocked; current may flow back into the en ergy harvester when the output voltage is higher than the input voltage. Therefore the second stage active diode is added to control the current direction (i.e. block reverse current) with minimum voltage drop and power loss. The design is exactly the same with the half wave converter as discussed before, except that the external diode is not needed, because the input to the

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57 active diode is now all positive. Although two halves of the available input power are utilized compared with the half wave converter, there is additional power loss due to the full wave rectification stage, which compensates the other half of the input power. The input powered design is also implemented on this converter such that the comparator power supply (VDD1) is connected to V IN through an external diode (D1). Another power supply (VDD2) is connected to the dc output of the converter (V OUT ). The schematic of the input powered comparator has been presented in Figure 2 3 and will not be repeated here. the voltage drop and power consumption induced by the four MOSFETs in the rectifier stage become significant when these MOSFETs are not fully turned on at low input amp l itude. 2.2.2 Circuit Implementation The input powered ac/dc converter is implemented in the On Semi 3M 2P 0.5 m CMOS process and packaged in a DIP40. Figure 2 10 shows the chip photo and chip micrograph. The total area for the ac/dc converter is 0.026mm 2 (130m 200m) as shown in Figure 2 10 (A) As shown in the enlarged layout image in Figure 2 10 (B) most of the die area is occupied by the four large switch transistors in the full wave rectifier and the PMOS switch in the active diode. 2.2.3 Measurement Result For characterization purpose s, a 20 Hz, 1 V amplitude sine wave from Stanford Research Systems SR780 signal analyzer is used as a baseline input waveform to mimic the output of a vibrational energy harvester. The function generator has an output ectrolytic capacitor is connected at the output as the storage element.

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58 2.2.3.1 Function Test Figure 2 11 (A) shows the input (V in ) and output (V rect ) waveforms of t he full wave rectifier stage As expected, the rectifier converts the negative part of the input to a positive one. The function of the entire ac/dc converter is also show n in Figure 2 11 (B). Here, the final dc output (V out ) successfully follows the peak of the a c input (V in ) with negligible voltage drop. A B Figure 2 10 Chip photo and microgra p h of the ac/dc converter chip A) Chip micrograph. B) Chip photo. (Photos courtesy of Yuan Rao). A B Figure 2 11 Open load experimental result when input is a sine w ave with 1V amplitude. A) F irst stage only. B) E ntire ac/dc converter

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59 2.2.3.2 Power and Efficiency Figure 2 12 shows the output power of the ac/dc converter at different load resistors and input voltages. When input amplitude is 3 V, the maximum output power is The maximum power point (MPP) happens when th e load impedance matches the output impedance of the ac/dc converter, and thus it may vary when the circuit internal impedance change at different input amplitude. For example, the MPP at 2.0 This behavior is because the circuit impedance is higher at 1 V input due to larger turn on resistance of the switching MOSFETs. Figure 2 12 Output power of the full wave ac/dc converter at various input voltage amplitudes (20 Hz) and load resistances The power efficiency of the full wave ac/dc converter is measured using the similar method with half wave ac/dc converter, as discussed in Section 2.1. 3 Figure 2 13 shows the simulation and experimental results of power efficiency for 1 V amplitude, 20 Hz input sine wave at various load resistances. At first, the power efficiency increases with increasin g load resistance because the increase in output

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60 begins to dominate over the output power. The peak efficiency of 82.4% occurs when Figure 2 13 Experimental measurements and simulation predictions of power efficiency of ac/dc converter for 1V amplitude, 20 Hz input sine wave 2.2.3.3 Test with an Energy Harvester To test the functionality of the circuit with a real energy harvester, the converter output waveform is measured with the ac input supplied from an electrodynamic vibrational energy harvester. The electrodynamic energy harvester [65] a spherical magnet inside a coil wound cavity was shaken by hand, generating a pseudo random voltage ranging from 1.5V to +1.5V A 10 k resistor are connected to the output. As shown in Figure 2 14 the circuit successfully rectified the input waveform with all the positive peaks detected. Figure 2 14 Functional test of the circuit using a vibration electrodynamic harvester

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61 2.3 Voltage Doubling Ac/dc Converter The v oltage doubling ac/dc converter is another commonly used ac/dc topology in vibration energy harvesting circuits, because it has the benefit of higher output voltage compared with half wave and full wave ac/dc converters. Therefore, in this section, a n input powered voltage doubler is demonstrated. The input powered scheme has the same concept with previously discussed full wave and half wave ac/dc converters, that the active part of the circuit, such as the comparator, is powered by the circ uit ac input. As a result, the circuit not only reaps the benefits of the voltage doubling ac/dc topology, but also consumes power only when the input is high enough for harvesting. 2.3.1 Circuit Design 2.3.1.1 Voltage Doubler Figure 2 15 shows the proposed voltage doubler. It consists of two half wave rectifiers: positive side rectifier and negative side rectifier. Each side of the circuit operates on the opposite half cycl e of the input waveform. The load (R LOAD ) is connected across the positive and negative output terminals. So the final dc output is the difference between the positive side dc output (V OUT +) and the negative side dc output (V OUT ), which is twice of the ac input amplitude (V IN ) in the ideal case. A PMOS switch is used in the positive side, and an NMOS switch is used in the negative side, so that the supply voltage requirement of the comparator is reduced. Meanwhile, to avoid reverse current through the body diode, the MOSFET must be connected in a way such that the body diode is oriented as shown in Figure 2 15 The parallel connection of the MOSFET body diode and the a ctive diode also helps pre

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62 charge the load before startup of the circuit. The W/L ratio of the PMOS and NMOS are 1500/1 in m. Figure 2 15 Voltage doubler based on active diodes The positive side comparator is the same as the comparator in the half wave converter and will not be repeated here. The schematic of negative side comparator is shown in Figure 2 16 with W/L ratio in m. Figure 2 16 Schematic of negative side comparator

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63 Differ ing from the NMOS input transistors in the positive side comparator, the negative side comparator uses PMOS input transistors because the minimum input amplitude is close to the positive rail (VDD). VSS1 and VSS2 in the negative side comparator are separated to reduce static power, which will be explained in detail later. The bias voltage is connected to VSS1 to eliminate the need of additional bias circuitry. 2.3.1.2 Input powered Scheme In the input powered scheme, the comparators are powered directly by the ac input (V IN ), as shown in Figure 2 17 The power supply of the first stage (VDD1 or VSS1) is connected to the input through an external diode (D1 or D2, respectively). The d iodes are required because the parasitic diodes between substrate and n well will be turned on when the substrate is not connected to the lowest voltage of the circuit. D1 and D2 are connected in the direction to prevent any reverse current flowing from th e substrate to the circuit. Since the diodes are not in the signal path, no additional conduction losses are induced. Figure 2 17 Voltage doubler with input powered scheme

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64 2.3.2 Circuit Implementation Both circuits were fabricated in the On Semi 3M 2P 0.5 m CMOS process and then packaged together in SOIC16 package. Note that the positive side circuit and the negative side circuits are fabricated in two dies to allow different connections of NMOS bulks (P+ substrates), because the On Semi 0.5 m process does not provide P well mask [66] The microphotographs of voltage doubler chipset are shown in Figure 2 18 The active area for the positive side chip is 0.012 mm 2 (196 m 62 m), while for the negative side chip is 0.013 mm 2 (217 m 60 m). T he total active area for the two chips is 0.025 mm 2 A B Figure 2 18 Microphotograph of voltage doubler chipset A) Positive side chip B) Negative side chip

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65 Figure 2 19 gives the photo of the packaged voltage doubler chipset. On the left is the package outline and on the right is the zoom in photo marked with dimensions. Both the positive side and negative s i de ac/dc converter dies have 1.6 mm by 1.6 mm square shape packaged into one SOIC16 package with outline size of 7.4 mm by 10 mm. The SOI C 16 package accommodates both dies and is covered with a glass lid. There are also some discrete components including two 220 F aluminum electrolytic capacitors (C1 and C2), and two NSR0320 Schottky diodes [64] (D1 and D2). Figure 2 19 Photo of the voltage doubler chipset in SOIC16 package. (Photos courtesy of Yuan Rao) 2.3.3 Measurement Result The complete voltage doubling ad/dc circuit is tested using a signal generator with controllable sinusoidal output voltage amplitude, and then demonstrated functionally with an electrodynamic vibrational energy harvester. 2.3.3.1 Minimum Operating Voltage For systematic characterization purposes, a sine wave from a Stanford Research Systems SR780 signal analyzer is used as a baseline input waveform to mimic the output of a vibra tional energy harvester. The function generator has an output impedance of less than 5

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66 Figure 2 20 shows measured input and output waveforms of the voltage doubler for a 20 Hz sine wave with the minimum operating amplitude. The outputs of positive side and negative side chips (V OUT + and V OUT ) successfully track the positive and negative peak of the AC input separately, so the final dc output is almost twice the inpu t amplitude. The voltage drop is very small compared with the input amplitude due to the open circuit load. As shown in Figure 2 20 the input powered scheme can work down to the minimum operating voltage of 0.7 V, equal to V TH of NMOS in the 0.5 m CMOS fabrication process, as expected. Figure 2 20 Measurement result of minimum input voltage 2.3.3.2 Output Power Figure 2 21 shows the measured output voltage at different load resistances (R LOAD ) and input voltage amplitudes (V IN ). Note that the maximum vol tage amplitude applied to the voltage doubler during the experiment was 3.0 V to avoid damaging the chipset. The results show that the circuit works well down to R LOAD er load resistance (e.g. R LOAD < 500 ) the input powered scheme does n ot work, because the output voltage drops below the minimum supply voltage required for the second stage of the comparators. To increase the comparator rail rail voltage, another cross

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67 connected scheme was reported on the same circuit, but uses a so called [67] Figure 2 21 Measured output voltage versus load resistances In addition to the output voltage, the achievable output power is also important for energy harvesting applications. Figure 2 22 gives the measured output power at di fferent load resistances (R LOAD ) and input voltage amplitudes (V IN ). The extracted output power ranges from tens of microwatts to several milliwatts. Figure 2 22 Measured output power versus load resistances For increasing input voltage amplitudes, the output power increases, as expected. For increasing load resistances, the output power decays because the

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68 re sistive load for maximum energy extraction is normally equal to the net harvester output impedance, and therefore impedance matching techniques [68] [69] are usually needed for the next stage to maxim ize the output power. However, impedance matching is reserved as another topic for later investigation. 2.3.3.3 Power Efficiency The power efficiency of the voltage doubler is the defined as where V OUT+ and V OUT are the positive side and negative side output dc voltage of the voltage doubler. R LOAD is the load resistance, V IN (t) and I IN (t) are the instantaneous input voltage and current to the circuit respectively, and T is the period of one ac cycle. The input current waveform I IN (t) is first measured with a Tektronix TCP312 current probe and a TCP300 amplifier and then displayed on Tektro nix TDS5104B digital phosphor oscilloscope together with the input voltage waveforms V IN (t) The oscilloscope calculates the time average input power from these waveforms. A Fluke189 multimeter measures the output dc voltage V OUT for output power calculat ion. Figure 2 23 shows both t he measur ement and the simulation results of power efficiency at various load, with an input sine wave of 1.5 V amplitude and 20 Hz frequ ency The circuit reaches a maximum power efficiency of 87 load The measurement result is in good accordance to the simulation result. At first, the power efficiency increases with increasing load resistance, because the increase in output powe r outpaces that of the conduction loss. Then beyond certain load resistance, the

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69 begins to dominate over the output power. Note that the efficiency decrease at large loa d is less important because the resulting output power is very low. Figure 2 23 Experimental measurements and simulation predictions of power efficiency with a sine wave of 20 Hz frequency and 1.5 V voltage amplitude 2.3.3.4 Test with an Energy Harvester The voltage doubling ac/dc converter is then functionally tested with an electrodynamic vibrational energy harvester, described in [65] This particular harvester architecture generates an aperiodic voltage waveform. As shown in Figure 2 24 the outputs (V OUT + and V OUT ) successfully track the positive and negative peak of the input (V IN ), which is generated from hand shaking of the harvester. Figure 2 24 Test result with a vibrational energy harvester

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70 2.4 Summary In this chapter, half wave, full wave and voltage doubling ac/dc converters are demonstrated for energy harvesting applications. All the circuits function without any external power supply and have zero standby power when the input is too low for energy harvesting Compared with conventional self powered ac/dc converter, the input powered function eliminates the need for pre charging and allows for indefinitely long intervals between charging cycles, which is critical for energy harvesting systems. All the circuits were implemented in the On Semi 3M 2P 0.5 m CMOS process. The minimum input voltage is 1 V for the full wave ac/dc converter, but only 0.7 V for the half wave and the doubling converters. There are two reasons for the relatively large input threshold vo ltage s of these design s First, the input powered feature requires sufficient input voltage to power the comparator. Second, the input voltage must be higher than the threshold voltage of the CMOS technology to turn on the switch MOSFET. It can be argued that a Schottky diode based passive converter can work at lower input voltage of around 0.2 5 V, compared with the 0.7 V minimum input th reshold of the active diode based converter. For energy harvesting applications, operation at low input voltage is indee d very important. However, at moderate voltage levels (>1 V), the active converter outweighs the passive one for its lower voltage drop. Figure 2 25 shows a simulation result of the voltage drop (i.e. the voltage difference between the ac input amplitude and the dc output voltage ) of a Schottky diode based half wave ac/dc converter and th e active diode based half wave ac/dc converter. The SPICE model of a low voltage Schottky diode [64] an ideal 20 Hz sine wave ac input and 10 k resistor load are used in the simulation

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71 Compared with relatively constant volt age drop (~0.24 V) of the Schottky diode based converter, the active diode based converter has much smaller voltage drop when the input amplitude is from 1 V to 3 V. As the input increases after the PMOS has been fully turned on, the current flowing throu gh the switch increases, yielding a slightly larger voltage drop. Note that w hen the input is less than the PMOS threshold (i.e. 0.9 V in 0.5 r voltage drop due to the weak turn on of the PMOS. Figure 2 25 Simulation result of the voltage drop between the ac input amplitude and the dc output voltage. Considering the three different input powered ac/dc converters, c riteria must be set topology for vibrational energy harvesting applications. As the front end interface circuit in an energy harvesting system, the goal of ac/dc converter is to extract as much power as possible from the harvester. There a re many factors determining the maximum power that the converter can extract, such as the converter topology, voltage drop of the switching transistor and the impedance matching of the harvester and the converter /load Therefore, to determine the best can didate for

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72 vibrational energy harvesting application, the output power of three topologies are measured and compared at the exact same input and load conditions. A B C Figure 2 26 Output power comparison of full wave, half wave and voltage doubling ac/dc converters. A) V in = 1 V B) V in = 2 V C) V in = 3 V

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73 The measurement result of the output power is shown in Figure 2 26 where all circuits are input powered by a 20 Hz sine wave from Agilent 33120A function generator In all cases, t he voltage doubler is shown to extract the m ost power from the simulated harvester source for input amplitude s ranging from 1 V to 3 V and load resistance ranging from 500 4 Meanwhile, the half wave ac/dc converter outperforms the full wave ac/dc converter in both output power and power efficiency. Another important parameter to compare ac/dc converters is the power efficiency. From the system point of view, power efficiency may not as critical as the output power, because system designers care more about the maximum power that can be extracted by the front end circuit. However, power efficiency of the ac/dc must be optimized t o achieve decent system efficiency, and the two m etrics of power efficiency and maximum power performance are linked. Figure 2 2 7 compares the power efficiency at various load resistance when the inp ut is a 20 Hz sine wave with voltage amplitude of 1.5 V. The result shows that the voltage doubler has a higher efficiency than both the half wave and full wave ac/dc converters, due to its high output dc voltage and less power hungry circuit components (e .g. swtiching MOSFETs). Figure 2 2 7 Power efficiency with the input amplitude of 1.5 V

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74 There are several reasons why the voltage doubling ac/dc converter (i.e. voltage doubler) out performs full wave and half wave topologies in vibrational energy harvesting applications. First, the single diode rectification used in each side of voltage doubler offers improved low power efficiency than the four diode full wave rectifier, especially w hen the diode forward voltage drop is comparable with the input amplitude [70] Second, in vibrational energy harvesting systems, the input ac voltage is often much lower than the desired output dc voltage on the load, and there fore the voltage doubling effect of voltage doubler can reduce the voltage amplification burden of a subsequent step up converter. Another important benefit is the reduced circuit complexity and power loss associated with the fewer number of diodes compare d with full wave topology, particularly if active diodes are used.

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75 CHAPTER 3 INPUT POWERED DC/DC CONVERTER In Chapter 2, i nput powered ac/dc converters have been discussed. The ac/dc converter converts the ac input to a dc voltage. However, c onsidering the low output voltage of electrodynamic vibration energy harvester s the dc output from just the ac/dc converter is always too low (i.e. hundreds of mV to a few volts) to be compatible with the load electronics. For example, to charge a 3.7 V rechargeable battery the open load voltage of the circuit must be higher or at least equal to 3.7 V. Moreover, this dc voltage is not regulated, which means it may fluctuate with the changing output voltage from the harvester. Therefore, a dc/dc converter is needed to provi de a dc voltage that is not only stable, but also high enough to charge the load (e.g. rechargeable batteries). This chapter presents a switch mode step up dc/dc converter with the goal of achieving Regulated constant output voltage (i.e. 3 V ). Large input voltage range (i.e. 0.1 V 2 V) High efficiently at light load (i.e. 100 A 10 mA) Compared with previously reported dc/dc converters in energy harvesting applications [71] [72] [73] the dc/dc converter developed here not only provides a regulated output but also adopts an input powered architecture that eliminates static power consumption when the circuit is in standby mode. Pulse Skip Modulation ( PSM) control method is implemented in the dc/dc controller for its decent light load efficiency and low circuit cost. F abricated by the On Semi 0.5 m 3M 2P CMOS process the circuit was successfully characterized and then bench top tested.

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76 3.1 Circuit Design In this section, a pulse skip modulation (PSM) control scheme of the dc/dc converter is fi rst introduced, and then the block diagram of the dc/dc converter is provided. The transistor level design of each circuit block in the dc/dc controller is the discu ssed in detail. 3.1.1 Pulse Skip Modulation The basic function of a dc/dc converter is to convert the dc input voltage to another dc output voltage with a larger or smaller magnitude. A switched mode dc/dc converter is one of the commonly used dc/dc topologies f or its high efficiency and small size. It converts the input voltage by temporarily storing the input energy in an inductor and then releases it at a d ifferent output voltage level. When the output voltage is higher than the input, the circuit is called a boost converter, or step up converter Figure 3 1 shows the basic schematic of a boost converter, which consists of an inductor (L), a switch, a diode (D) and an output capacitor (C). Figure 3 1 Basic schematic of a boost converter When the switch is closed, the inductor absorbs input energy, yielding an increase of the inductor current, as shown in Figure 3 2 (A) Then, when the switch is opened, the accumulated inductor energy is transferred to the output capacitor through the diode, because it is the only path that the inductor current can flow through. Since

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77 the voltage across the inductor is related to the rate of the current change, not the input voltage, a higher output voltage is generated. A B Figure 3 2 Boost converter circuit during two operating intervals A) W hen the switch is closed B) W hen the switch is open A controller is always desired in dc/dc converter to produce a stable, regulated output voltage for various input voltage s and load current s As shown in Figure 3 3 the dc /dc controller forms a feedback loop, allow ing control signal (i.e. the switching pulse) to change with the feedback signal from the out put and an external reference voltage Figure 3 3 Basic boost converter with a dc/dc controller There are three commonly used control methods to realize dc/dc controllers: Pulse Width Modulation (PWM), Pulse Frequency Modulation (PFM), and PSM (Pulse Skip Modulation). PWM control has fixed switching frequency and regulates the output voltage through to heavy load

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78 because of its fixed switching loss. PFM (Pulse Frequency Modulation) control uses a fixed duty cycle and regulates the output voltage through adjusting the switching frequency. Since t he switching loss can b e reduced with the load current, PFM typically is popular for power sav ing mode at light load PSM however, uses both fixed duty cycle and fixed switching frequency. It regulates the output voltage through connecting or disconnecting the switching pulse s with the switching transistor. An example of PSM control is shown in Figure 3 4 Whenever the feedback signal is below the reference, the control switching pulse is applied so that the inductor starts charging and discharging energy, generating a higher output voltage unti l it reaches the reference. Although the regulation resolution of PSM is limited compared with PWM and PFM it i s chosen in this work for its decent efficiency at light loads and low circuit complexity. Figure 3 4 An example of PSM control scheme The most important parameter to characterize a dc/dc converter is its power efficiency, the ratio between the total output power and the total input power. Ideally, all the input power will be transferred to the output, so power efficiency should be 100%. However, in real converters, the efficiency cannot reach 100% because of power losses, which can be classified as conduction loss, switching loss and fixed loss.

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79 Conduction loss refers to th e loss of switching transistor on resistan ce, diode forward voltage drop, inductor winding resistance and capacitor series resistance. Switching transistor and diode parasitic capacitance charge and discharge, however, mainly cause switching loss. Another loss is due to controller standby current a nd leakage current of the diode, transistor s and output capacitor Since this loss is relative ly constant in spite of the input voltage and load current, the PSM architecture has a fixed loss. All above loss mecha nisms limit the power efficiency of a dc/dc converter in real applications. 3.1.2 Circuit Diagram The circuit diagram of the input powered boost converter is shown in Figure 3 5 where an inductor L, a diode D, a switching transistor M 1 and an output capacitor C OUT together form a basic boost converter. An on chip dc/dc controller, highlighted by the dotted lines, is powered from the input (V IN ), much like the ac/dc converter s in Chapter 2. This input powered feature eliminates the need for external power supplies, reduces the pin count, and avoids standby power consumption, all of which are critical in energy harve sting applications. T he dc/dc controller uses PSM to achieve a regulated output dc voltage (V OUT ), in the presence of variation in the input voltage and load current A resistor ladder (R 1 and R 2 ) senses the output voltage level (V IN ), and generates a fee dback voltage V FB to be compared with the reference voltage V REF When V FB is lower than V REF the error comparator output (V C ) goes high to activate a n on chip ring o scillator via a level shifter. The oscillator generates a switching pulse with fixed duty cycle and frequency which drives the switching transistor M 1 through an on chip buffer Transistor M 1 cycles on and off to transfer power from the input to the output until V FB is higher than V REF at

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80 which point V C goes low and thus both the level shifter and the oscillator are disabled. M 1 remains off until sometime there after the load (R LOAD ) discharges the capacitor C OUT and again V FB drops below V REF the process restarts. In PSM, the inactive period increases a t light load (i.e. small load current) which helps maintain low overall power consumption and a relatively high efficiency of the converter. Note that a NMOS transistor M 2 is added in to the resistor divider ladder used to cut off the leakage path during standby mode. When V A is below tge NMOS threshold voltage, which is around 0.7 V in the 2 turns off and prevent s any leakage current flowing through the resistor divider. To avoid discharging the load when the circuit is in standby m ode, the comparator in the controller circuit is input IN ), as shown in Figure 3 5 Although the oscillator, the level shifter and the buffer are powered or partially powered by the output voltage (V OUT power when they are not active. T he detail explanation will be presented in the following sections on the transistor level design of each block. Figure 3 5 Block diagram of the input powered boost converter

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81 3.1.3 Error Compar ator As shown in Figure 3 6 a two stage latching comparator is designed, with the transistor size listed in Table 3 1 The NMOS differential pair (M3 and M4 ) in the first stage is chosen for its lower input common mode range, with a large size to reduce the offset. The cross coupled PMOS (M1 and M2) uses the minimum size transistors to make the latch status swap easily when the input changes. The PMOS transistor pair is further weakened by adding another two diode connected PMOS transistors (M9 and M10). The bias current is controlled by the input signal EN through NMOS tr ansistors (M5 and M8). The pull up PMOS (M6) in the second stage is sized large enough to make sure the output (V OUT ) is close to VDD when it is turned on. NMOS transistor M7 has the same size with input transistors (M3 and M4), to reduce the systematic of fset from the second stage. Both EN and VDD are connected to the dc input (V IN ) of the boost converter, as shown in Figure 3 5 allowing the circuit to enter standby mode with no static current when EN is too low to turn on the bias transistors, or when VDD is not providing enough headroom for the comparator to work. Figure 3 6 Schematic of the error comparator

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82 Table 3 1 Transistor size of the error comparator The system offset is simulated by DC analysis by Spectra Spice in Cadence. The result is shown in Figure 3 7 where the systematic offset is less than 10 mV for EN an d VDD ranging from 1 V to 3 V. However, the random offset, caused by parameter variation, fabrication mismatch, temperature fluctuation and etc, is not included here. It may be predicted by Monte Carlo simulation, but the accuracy highly depends on the ava ilable model from the fabrication technology, which is beyond the scope of this work. A B Figure 3 7 DC analysis of the comparator offset. A) EN=VDD=3 V. B) EN=VDD=1 V 3.1.4 Level Shifter One of the main con duction losses in the dc/dc converter is the conduction loss of the switching transistor (i.e. M 1 in Figure 3 5 ). It become s even worse in the input powered design, where the switching pulse swing is up limited by the input V IN and thus Transistor M1, M2, M8, M9, M10 1.5/0.6 M3, M4, M7 15/0.6 M6 9/0.6 M5 3/0.6

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83 M1 may not be fully turned on when V IN is low. To solve this problem, a level shifter is introduced to reduce the conduction loss of the swit ching transistor (M 1 ) by increasing the voltage swing of the switching pulse on its gate. Figure 3 8 gives the schematic of the level shifter with the transistor size in Table 3 2 D2 by cross remains unchanged at VSS. As a result, the input (V IN ) and the output (V OUT ) digital signals have the same frequency and phase, except that the voltage swings change from the input VDD (VDD1) to the output VDD (VDD2). The two inverters (M1 and M3, M2 and M4) are powered by the VDD1, generating sharp digital pulses to control the gates of M8 and M7, separately. Figure 3 8 Schematic of the level shifter The latch stage and the output inverter (M9 and M10) are powered by VDD2, which will be connected to the boost converter output (V OUT ), as shown in Figure 3 5 Although these circuits are not strictly power when there is no input switching pulse. Therefore, the level shifter can enter

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84 sleep mode with n o static power when the input (VDD1) is too low to supply enough rail rail voltage, or there is no input pulse from the error comparator. IN ) is relatively low (i.e. VDD1 is close to 1V), strong NMOS tra nsistors (M7 and M8) and weak PMOS transistors (M5 and M6) are sized to make sure the drain of M7 can be pulled down to VSS when VIN is high. Simulation result shows that the level shifter can work with VDD1 ranging from 1 V to 3 V, and VDD2 ranging from 3 V to 4.5 V. Table 3 2 Transistor size of the level shifter 3.1.5 Voltage Controlled Oscillator Figure 3 9 shows the schematic o f the on chip voltage controlled oscillator (VCO), which can generate the switching pulses to drive the switching transist or. The oscillator is composed of a series of current starved inverters and NAND gate back coupled to provide an unstable state that leads to oscillation. The three input NAND gate has one input (EN_OSC) that will be connected to the level shifter output. W hen the feedback voltage is higher than the reference, EN_OSC goes low, and t he oscillator stops oscillating, which turns off the switching MOSFET (M 1 in Figure 3 5 ) Once the harvester output voltage is high enough, EN_OSC goes high and the oscillator resumes oscillation. Transistor M1, M2, M9 6/0.6 M3, M4, M10 3/0.6 M5, M6 1.5/12 M7, M8 30/0.6

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85 Figure 3 9 S chematic of the voltage controlled oscillator The schematic of the current starved inverter and NAND gate is presented in Figure 3 10 with transistor size listed in Table 3 3 The control voltage (V m ) sets the current flowing through current starved logic gates which subsequently controls the delay of each stage, allowing tunable oscillator frequency. Transistor length in the inverter is sized large (15 ) for achieving long enough delay of gate. A B Figure 3 10 Current starved logic gates in the ring oscillator A) Inverter. B) NAND gate 3.1.6 Switching MOSFET and Buffer To reduce the conduction loss, the switching NMOS transistor (M 1 ) in the boost converter ( Figure 3 5 ) must be as large as possible to reduce the on resistance.

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86 However, larger MOSFETs suffer from larger area, increasing switching loss and lower speed. Based on the tradeoffs, the size of M 1 is chosen to be 900/0.6 in Table 3 3 Transistor size of the current starved logic gates To improve the dr iving capability, a voltage buffer is inserted between the oscillator and the switching MOSFET. As shown in Figure 3 11 the voltage buffer consists of three inverters (INV1 to INV3), where INV1 is a m inimum size inverter (NMOS: 1.5/0.6 PMOS: 3/0.6 ), and INV2 and INV3 are 16 times and 256 times of INV1 respectively. The buffer is powered by the dc output of the dc/dc converter, as shown in Figure 3 5 pulses. Figure 3 11 Schematic of the voltage buffer 3.2 Circuit Implementation The dc/dc controller circuit wa s fabricated in silicon using the On Semi 3M 2P 0.5 m CMOS process. Figure 3 12 shows the die micrograph where the total active area is about 0. 05 mm 2 ( 430 m 115 m). The oscillator and the voltage buffer, occupy most of the active area. Transistor M1, M2 3/15 M3 1.5 / 15 M4, M5, M8 6 /0.6 M6, M7 3/0.6

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87 Figure 3 12 Micrograph of the dc/dc controller die The dc/dc controller die, with a size of 1.6 mm by 1.6 mm, is then packaged in a SOIC 24 package, as shown in Figure 3 13 The outline of the final packaged controller chip is about 10 mm x 7.4 mm. Figure 3 13 Photo of the dc/dc controller chip (Photos courtesy of Yuan Rao) Besides the on chip circuit, there are some other discrete components, summarized in Table 3 4 They are designed to optimize the boost converter around the possible ope rating condition of the system, as discussed below in detail. Normally the inductor used in portable DC DC converter is tens to hundreds H. Larger values of inductors have smaller current ripple and therefore higher efficiency at light load. However, the transient performance of the power converter will be degraded

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88 with increasing inductance value. In addition, a larger inductance leads to a larger volume which is not desirable in energy harvesting applications where system volume is limited A high Q inductor is preferred to realize a high efficiency boost converter. Normally an inductor is selected having a quality factor ( Q ) in excess of 50 at the switching frequency. In this work, a 22 H vertical mount coil inductor [74] with Q of about 300 at 100 kHz,,is chosen based on the switching frequency, current ripple, and inductor size. Table 3 4 List of discrete components in the system prototype The output diode (D) in the dc/dc converter impacts the power efficiency of the boost converter directly. The diode ) can be expressed as where V F (I OUT ) is the diode forward voltage drop at the output current I OUT V OUT is the output voltage of the boost converter. Hence, a low V F Schottky diode is preferred in the also limits the circuit efficiency because of the leakage current flows fr om the load when the diode is reverse biased. In the experiment, several Schottky diodes are tested and Type Name Implementation Inductor L 1 Diode D NSR0320 [75] Capacitor C OUT F Resistor R 1, R 2

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89 an ultra low voltage drop (V F =0.24 V at 10 mA dc) Schottky diode NSR0320 [75] is selected considering all the factors. Th e output capacitor (C OUT ) determines the output voltage ripple and the circuit response time. A large output capacitance reduces output voltage ripple, but also increases the circuit time constant and the capacitor volume. Taken into account these trade of fs, C OUT is selected to be The resistor divider in the dc/dc feedback loop must have negligible power loss to the overall converter Therefore, R 1 and R 2 should be much higher than the load resistance, so that power loss in the resistor divider is negligible compared with the output power. The larger the resistance of the divider is, the smaller the current flows into the resistor divider during normal operation. Meanwhile, the value of R 1 and R 2 should make sure that the feedback voltage V FB stays in the input common mode range of the error comparator. Considering the tradeoff betwe en power loss and resistor size and V FB range, R 1 = 2 2 are selected in this design. 3.3 Experimental Result The input powered dc/dc boost converter pe rformance is bench top characterized using a dc power supply (Agilent E3630A) as the input source. The reference voltage (V REF ) is set to 1 V by another dc power supply (Agilent E3630A), to achieve 3 V regulated output (V OUT ). Note that this reference volt age will be replaced by an on chip bandgap reference circuit in future designs. 3.3.1 Function Test Figure 3 14 shows the general function and the input controlled standby operation of the boost converter. The screenshot includes waveforms of dc input (V IN ), inductor current (I L ), switching signal (V OSC ), and dc output (V OUT ). At first, when V IN is

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90 zero, the circuit is in standby mode. Therefore there are no switching pulses and no current flowing through the inductor. However when the input jumps to 1.5 V, the oscillator starts generating the switch ing pulse, and inductor current is close to a triangular waveform. ly determined by the output capacitor Under these conditions, the input dc voltage is successfully boosted up, and V OUT is stabilized at 3 V due to output regulation. When the input voltage drops, the circuit enters standby mode, and V OUT gradually decreases via discharge through the load resistor ( R LOAD = Figure 3 14 Screenshot of input voltage, inductor current, switching signal and output voltage of the boost converter The relationship between the oscillator switching frequency and the control signal (V m ) is also measured. The measured VCO frequency can be up to 228 kHz when V m is grounded compared with 340 kHz in the simulation, as shown in Figure 3 15 The difference is due to the parasitic resistance and capacitance introd uced in the measurement. By adjusting V m the optimum switching frequency can be achieved which depends on the voltage gain, inductor size and the load condition. Note that w hen

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91 V m is higher than 2 V, or EN_OSC is below 1.5 V, the oscillator stops oscillat ing with no standby power consumption. Figure 3 15 Simulation and measurement result of VCO output frequency 3.3.2 Power Efficiency In dc/dc converters direct measurement of loss is complicat ed but measurement of the power efficiency of the boost converter is straightforward because all the input and output signals are dc. As mentioned before, powered efficiency of a boost converter is defined as the ratio between total output power and total input power, given by where V IN and I IN are the input dc voltage and current, V OUT is the DC value of the output voltage, and R LOAD is the load resistance all of which are measured by Fluke 189 digital multimeters directly. The measured power efficiency with a variety of load currents is shown in Figure 3 16 For a 3 V regulated output, maximum efficiencies of 86% and 90% are achieved for inputs of 1.5 V and 2 V, respectively. However, t he power efficiency at both light load and heavy load is poor The efficiency at light load is low because of the switching loss and the fixed loss of the

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92 controller circuit. While at heavy load, power loss is mainly contributed by conduction loss, which is proportional to the square of output current. Therefore whe n the output current increases, the power efficiency increases because of the reduced impact of the switching loss and the fixed loss. Note that the circuit can still function with an input as low as 1 V, but the efficiency drops quickly due to increased t urn on resistance of the transistors. Figure 3 16 Measured power efficiency of the boost converter at different loads for regulated 3 V dc output 3.4 Summary An input powered boost converter is presented with input controlled standby mode and zero standby power. This circuit eliminates the need for pre charging and allows for indefinitely long intervals between charging cycles, which is critical for energy harvesting systems. The chips are im plemented in ON Semi 3M 2P 0.5 m CMOS process and packaged in a SOIC 24 package The measured results show the system cold starts at 1 V amplitude input and works properly with input sine waveforms with amplitudes ranging from 1 V to 3 V and frequencies r ange from 1 Hz up to 100 kHz. The system enters standby mode when the input drops below 600 mV with zero standby

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93 power consumption. For reasonably low 1.5 V input amplitude and 3 V regulated output the net circuit efficiency is up to 86 %.

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94 CHAPTER 4 COMPLETE INPUT POWERED INTERFACE CIRCUIT A complete interface circuit for vibrational energy harvesting systems typically consists of an ac/dc converter and a dc/dc converter. Input powered ac/dc converters and the dc/dc converter have been discussed in Chapter 2 and Chapter 3, separately. Here, a complete input powered interface circuit, combing ac/dc converters and the dc/dc converter, is presented and investigated. In this chapter, the circuit design is first introduced. Then the bench top measurement setup and experimental result with an ac input from a function generator are discussed in detail. 4.1 Circuit Design The block diagram of the circuit is shown in Figure 4 1 and the corresponding bonding diagram is in APPENDIX B The complete energy harvesting interface circuit consists of an ac/dc stage and a dc/dc stage, converting the energy from the input source to the load. V S represents the ac input source with an internal impe dance of R S and a constant voltage (CV) load to represent the load (i.e. rechargeable batteries). T wo ac/dc converters are used in the ac/dc stage: a voltage doubler, which serves as the primary rectifier in the power path and a n auxiliary half wave ac/dc converter, which is used to provide a more stable supply voltage for the dc/dc converter stage The positive side of the voltage doubler converts the ac input to a positive dc voltage (V P ) which is then regulated by the dc/dc stage The negative side vo ltage doubler generates a negative dc voltage, which is connected to the ground (VSS) of the dc/dc stage. Therefore, the voltage difference between the dual outputs of the voltage doubler is the actual dc input seen by the dc/dc converter. Choosing voltage doubler as the front end ac/dc converter in vibrational interface circuit has the benefit of higher

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95 output power, and also lower voltage conversion burden to the dc/dc stage, as discussed in Chapter 2. Figure 4 1 Block diagram of the complete input powered interface circuit Although the output from the voltage doubler is dc at open load due to the capacitor s (C 1 and C 3 ), there will be an increasing ripple on th is dc voltage w hen the cir cuit is loaded and the load current (I LOAD ) increases. The ripple frequency is much lower than the switching frequency of the dc/dc controller, and therefore it will not cause any stability problem. However, the voltage sag, caused by the load dependent ri pple, makes the input powered design challenging, because the dc/dc controller requires sufficient rail to rail voltage to function. To overcome the voltage sag and ripple that arise on V P with heavy load currents, a half wave ac/dc converter is added as a n auxiliary rectifier, which generate s a low ripple supply voltage (V A ) for the dc/dc controller. The more stable voltage V A improves the overall voltage by providing a load independent supply to the dc/dc converter.

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96 mentioning that the complete interface circuit is input powered. All the ac/dc converters in t he ac/dc stage are powered by their ac input in a straightforward A instead of the dc input V P supplies the dc/dc controller in this stage. However it is still an input powered converter, considering that the power of V A is also from the input side voltage source ( V S ). 4.2 Experimental Result In this section, the complete interface circuit is characterized usin g an ideal 20 Hz sine waveforms from a function generator (Agilent 33120A) to mimic the output of a low frequency electrodynamic energy harvester. 4.2.1 Measurement Setup The measurement setup of the interface circuit is shown in Figure 4 2 The ac input is from a function generator (Agilent 33120A), and the reference voltage (V REF ) to the dc/dc controller is provided by a dc power supply ( Agilent E3616A ). The input voltage v in (t) is measured by an oscilloscope (Tektronix TDS5104B). The input current i in (t) is first sens ed by a current probe (Tektronix TCP312) and then amplified by a current amplifier (Tektronix TCP300) Both the input voltage and the input current waveforms are displayed on the same oscillosco pe, so that t he time averaging input power can be calculated by the mathematic function of the oscilloscope Meanwhile, the output from the positive side ac/dc converter (V P ) and the half wave ac/dc converter (V A ) are me asured by the other two channels of the oscilloscope. T he output dc current from the voltage doubler (I P ) and the half wave ac/dc converter ( I A ) are measured by Fluke189 digital multimeters. Another two multimeters are used to measure the dc output voltage s from the negative side voltage doubler and the dc/dc converter. The

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97 load is from an electronic load (BK Precision 8500), where constant resistance (CR), constant current (CC) and constant voltage (CV) loads are selectable. Figure 4 2 M easurement setup of the complete input powered interface circuit 4.2.2 Minimum Operating Voltage To illustrate the benefit of the minimum input threshold by adding the auxiliary half wave ac/dc converter the circuit is measured with a 20 Hz, 1.2 V amplitude sine wave input ( v in which represents a medium load ( I LOAD =3.7 mA ) condition As shown in Figure 4 3 the auxiliary ac/dc converter output (V A ) has negligible voltage ripple and a steady dc voltage of 1. 1 V ( 92 % of V in ). Conversely, the positive side voltage doubler output (V P ) has a large ripple with an aver age value of 0. 8 V (only 67 % of V in ). This illustrates the improved stability of t he voltage V A at medium to heavy load conditions.

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98 Figure 4 3 Measurement result at 1 the input is a 20 Hz, 1 .2 V amplitude sine wave Measurements show that at open load the entire circuit interface turns on when the input amplitude is above 1 V and turns off when the input amplitude drops below about 600 mV. When the circuit is off, there is no measurable standby power consumption. Figure 4 4 presents a screenshot of the startup process at open load when the input is a 20 Hz, 1.2V amplitude sine wave. The startup time is about 500 ms, which depends on the load capacitance s (C 1 and C 2 ). With no load current, V A and V P have almost the same startup wavefo rm and negligible voltage ripple. Figure 4 4 Circuit start up process at open load when input is a 20 Hz, 1.2 V amplitude sine wave

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99 Further measurement shows that, with a 3.7 V CV load, the interface circuit functions with minimum input voltage of 1.2 V pk In comparison, in a prior circuit implementation without the half wave ac/dc converter [76] [77] the minimum input threshold was 1.5 V pk 4.2.3 Output Power and Efficiency As shown in Figure 4 5 the measured output power increases with increasing input amplitude V in,pk when the charging a 3.7 V CV load. The power delivered to the load ranges from 1.1 m W for a 1.2 V in,p k input up to 22.6 mW for a 3 .0 V in,p k input. Note input amplitude to 3 .0 V. Figure 4 5 Power delivered to 3.7 V CV load at different inp ut voltage amplitude According to the measurement setup shown in Figure 4 2 t he power efficiency of the ac/dc stage ( ), the dc/dc stage ( ) and the overall interface circuit ( ) can be calculated as

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100 where v in (t) and i in (t) are instantaneous input voltage and current (ac), V P V N and I P are the output voltage and current (dc) of the voltage doubler V A and I A are the output voltage and current (dc) of the half wave ac/dc converter, V OUT and I OUT are the output voltage and current (dc) of the system The time T is the duration of measurement, chosen to b e larger than at least 10 times the input period. Figure 4 6 shows the bench top measurement result of circuit power efficiencies of the ac/dc stage, the dc/dc converter, and the overall interface circuit at different input voltage amplitudes. The over all efficiency is above 6 0% for input voltages >2.5 V pk In this design the overall efficiency is limited by both the ac/dc and the d c/dc stage. For instance, with a 20 Hz sine wave with input amplitude of 2. 6 V and regulated dc output of 3.7 V, the system achieves an overall efficiency of 61 % when delivering 16.7 mW of output power; the efficiency of the ac/dc stage and the dc/dc stage are 84 % and 7 3 %, respectively. Figure 4 6 Power efficiency of interface circuit vs. input voltage amplitude for regulated 3 .7 V dc output

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101 The ac/dc converter efficiency increase s with the input amplitude because conduction loss decreases with smaller R ds (on) of switching transistors (e.g. switc h MOSFETs in the rectifiers). For the boost converter, the power efficiency at light load is poor because of the switching loss and the fixed loss of the controller circuit. However, at medium heavy load, power loss is primarily conduction loss of the swit ching MOS (e.g. M 1 ), which is proportional to the square of output current. Therefore when the output current increases, the power efficiency increases because of the reduced impact of the switching loss and the fixed loss. 4.3 Summary This chapter combines the circuits introduced in Chapter 2 and Chapter 3, to build a complete, input powered interface circuit for electrodynamic vibrational energy harvesting systems The complete interface circuit, consisting of an ac/dc stage and a dc/dc stage, is able to co nvert ac input from into usable dc voltage level The ac/dc stage includes two ac/dc converters: a voltage doubling ac/dc converter and a half wave ac/dc converter. The voltage doubler is chosen for its relatively higher output power, whereas the auxiliary half wave converter helps provide a load independent power supply to the dc/dc controller. By implementing input powered design on both stages the entire interface circuit requires no external power supplies and features zero standb y power when the inpu t amplitude is less than 600 mV When the input amplitude is above 1.2 V, the circuit starts to charge 3.7 V constant voltage load with output power range of lithium ion polymer battery at an average power of 1.1 mW to 22.6 mW. With a 20 Hz sine wave input the system achieves an overall efficiency of 61 % when delivering 16.7 mW of output power when the input amplitude is 2. 6 V and regulated dc output is 3.7 V

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102 CHAPTER 5 RESONANT ELECTRODYNAMIC ENERGY HARVESTING SYSTEM MODELING The ultimate goal of an energy harvesting system is to deliver the maximum power to an electrical load. A reliable and quantitative system model that accurately represents all three blocks (harvester, interface circuit, and load) is therefore highly desirable, so that the entire energy harvesting system can be designed as a whole, as opposed to independent design of each block. The energy harvesting system depends on many different physical behaviors, for example the electromechanics of the harvester, the electrical behavior of the inter face circuit, and the electrochemical response of a storage battery. An appropriate system model must be able to accurately model all of these different physics in one common framework, Equivalent circuit representations are used in this work to model the system for three reasons. First, since electrical power delivery is one of the primary design goals, circuit representations are a natural choice. Second, circuit models can represent every block in the system. For instance, the power converter is already an electrical circuit with standard circuit elements. Most of the electrical loads can be simplified using an equivalent circuit model. An equivalent circuit network can also represent the coupled electromechanical behavior of the transducer. The third rea son is that circuit simulation tools can be leveraged allowing system level analysis and optimization, even with complex power electronic circuits. In this chapter, the reduced order models of the resonant electrodynamic transducer, the interface circuitr y and the electrical load are first introduced. Then parameter extraction methods for the harvester and the interface circuit are explained with examples. The system level model is then presented and simulated. Conclusions

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103 are made based on the comparison between Simulation Program with Integrated Circuit Emphasis SPICE simulations and experimental measurements. 5.1 Reduced order Models 5.1.1 Electrodynamic Energy Harvester Model The schematic of a resonant type electrodynamic energy harvester is shown in Figure 5 1 where a permanent magnet of mass is connected to a frame through a spring with stiffness and surrounded by a coil. A dash box represents the mec hanical damping with damping coefficient b. When external vibration occurs, the frame moves with velocity of amplitude .The relative motion of the mass relative to the coil with velocity amplitude of causes magnetic flux change. When the load is connected across the coil, a current will flow to the load and the power is delivered. Figure 5 1 Schematic of a resonant electrodynamic energy harvester The electrodynamic energy harvester is a multi energy domain transducer that operates in both the mechanical domain and the electrical domain, which makes analysis and co simulation with the interface circuits difficult. To solve this problem, a lumped ele ment model (LEM) equivalent circuit will be used to model the behavior of the resonant type vibrational energy harvester with discrete circuit elements. Many

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104 modeling methods have been reported for better understanding of the system dynamics and optimizati on of electromechanical transducers. Of these, LEM is a simple and effective method for modeling transducers across multiple coupled energy domains [78] [79] The LEM technique is appropriate when t he wavelength or diffusion length of the physical phenomena is much larger than the characteristic length scale of the transducer [78] In this work, the operating frequency of the harvester is 42 Hz, which corresponds to an e lectrical wavelength of 7137 km and an acoustic wavelength of 8.2 m. Clearly the device size (~few centimeters) is much less than either of these, so LEM is a valid method for modeling the transducer behavior. LEM allows the multiple energy domain systems to be represented with equivalent circuit elements. Each energy domain is represented by a pair of conjugate voltage, and the flow variable is current. In the me chanical domain, the effort variable is force and the flow variable is velocity. In both energy domains, the product of the effort and the flow is power (units of Watts). In each energy domain, every energy storage or dissipation mechanism is categorized into one of three types: generalized kinetic energy storage, gener alized potential energy storage or generalized energy dissipater. Kinetic energy is represented by a generalized inductor, so that t he energy obtained at a non zero flow is associated with a non zero velocity of the inductor. Similarly, poten t ial energy is represented by a generalized capacitor, and the energy stored at a non zero effort is associated with a non zero displacement of the capacitor. The energy dissipater is represented by a

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105 gen eralized purely passive resistor which absorb s energy dissipated in any flow or effort conditions. Therefore, in the electrical domain, kinetic energy is stored in the inductor, while in the mechanical domain kinetic energy is stored in the proof mass. Potential energy in the electrical domain is stored on a capacitor, while the compliance of a spring stores the potential energy in the mechanical domain. In the electrical domain, energy is dissipated across a resistor as Joule heating whereas in the mec hanical domain, energy is dissipated through friction and damping. Using this method, a mechanical mass can be modeled by an inductor, a mechanical compliance can be modeled by a capacitor, and a mechanical damper can be modeled by a resistor. Table 5 1 summarizes the variables and elements in energy domain conversion. Table 5 1 Variables and elements in energy domain conversion. Electrical Domain Mechanical Domain Effort variable Voltage ( ) Force ( ) Flow variable Velocity ( ) Current ( ) Stored kinetic energy Mass ( ) Inductor ( ) Stored potential energy Compliance ( ) Capacitor ( ) Dissipated energy Damping coefficient ( ) Resistor ( ) The LEM to describe the behavior of the electrodynamic energy harvester is shown in Figure 5 2 This equivalent circuit model [80] contains two domains: the mechanical domain (on the left) and the electrical domain (on the right) that are coupled by a gyrato r, which represents the electrodynamic transduction between these two

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106 domains. The gyration resistance of the gyrator is equal to the transduction coefficient and therefore given by where is the average flux density and is the total length of the coil [80] Figure 5 2 LEM of the electrodynamic energy harvester. The input vibration is assumed to be a sinusoidal excitation of the harvester. This excitation is modeled as a constant amplitude ac current source with amplitude where equals to the velocity amplitude of the frame ( ). On the left side of the gyrator, the proof mass is represented by an inductor with inductance and thus the energy stored in the inductor ( equals the kinetic energy of the mass ( A capacitor represents the mechanical compl iance with capacitance of so that the energy stored in the capacitor ( equals the energy stored in the spring ( A resistor represents the mechanical damping of the device and equals to the damping coefficient b, to model the energy d issipated in the mechanical domain. Note that the capacitor and the resistor are connected in series because the spring and the damper share the same velocity.

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107 On the right side of the gyrator, R coil and L coil represent the coil winding resistance and the coil winding inductance, respectively. The load is represented by an appropriate equivalent circuit, which in this case will be the harvester interface circuit and will be described later. More details of the electrodynamic LEM model can be found in [80] Figure 5 2 represents the harvester in both the mechanical and electrical domains. If the circuit can be simplified to the electrical domain only, the analysis will become easier because of its better interaction with an electrical circuit simulator. To do this, the circuit elements in the mechanical domain can be reflected across the gyrator f rom the mechanical to the electrical domain. The reflected impedance ( of an element is given by In this calculation method, a resistance remains a resistance. However, a capacitor becomes an inductor and vice versa. Meanwhile, a parallel connection becomes serial connection and vice versa. Additionally, the relationship between the source and reflected sources are given by where is the flow variable and is the reflected effort variable. So the mechanical velocity source now becomes a voltage source Figure 5 3 shows the circuit model after reflecting all the left hand side components of Figure 5 2 to the right hand side. Now common circuit techniques can now be applied. For example, Figure 5 4 sho ws the Thvenin equivalent circuit model. T he imaginary part of the source impedance is represented by whereas the real part is represented by

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108 Figure 5 3 LEM reflected into electrical domain. Figure 5 4 Thvenin equivalent circuit of electrodynamic harvesters. The equivalent voltage source ( ) and source impedance ( ) in the harvester model are given by The simplified equivalent circuit model in Figure 5 4 in the electrical domain now makes circuit analysis more straightforward. For example, the maximum power transfer to the load can be achieved when the load impedance which is also the input impedance of the interface circuit, equals to the co njugate complex as

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109 Therefore when this optimal load ( ) is connected to the transducer, the maximum power ( ) can be derived by 5.1.2 Interface Circuit Model The transistor level interface circuits presented in previous chapters are simplified here for system modeling, to make the system a nalysis much easier. In the simplified equivalent circuit model, only the primary behavioral functions and main loss contributors are considered. 5.1.2.1 Ac/dc Converter The interface circuit presented in Chapter 4 uses a voltage doubler in the main signal path. Since the voltage doubler actually consists of two half wave ac/dc converters, the half wave converter is first discussed before the voltage doubler. The half wav e ac/dc converter, introduced in Chapter 2 ( Figure 2 1 ), consists of an active diode and a storage capacitor. The simplified behavioral circuit is shown in Figure 5 6 where V S and Z S are the Th venin equivalent voltage and impedance of the harvester. In the active diode, both the comparator and the switch MOSFET are

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110 idealized. An additional series voltage source V D is added equivalent to the comparator offset. The switch MOSFET is replaced by an ideal switch, but at the expense of another series resistor R D to represent the on resistance. Figure 5 5 A half wave ac/dc converter The circuit can be further simplified by using an ideal diode D (i.e. zero voltage drop) to replace both the ideal comparator and the ideal switch. As a result, the final equivalent circuit model of the active half wave rectifier is given in Figure 5 6 Figure 5 6 Equivalent circuit model of the half wave ac/dc converter Since the voltage doubling ac/dc converter consists of two half wave ac/dc converters with one for the positive half wave input and one for the negative half wave input, the equivalent circuit model is a straightforward extension, as shown in Figure 5 7

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111 Figure 5 7 Equivalent circuit model of the voltage doubling ac/dc converter 5.1.2.2 Boost Converter The boost converter ( Figure 3 5 ) in the energy harvesting circuit can be simplified as shown in Figure 5 8 (A) where R g and V g represent the output voltage and impedance from the previous ac/dc stage, respectively However, the conduction losses contributed by the inductor (L), the diode (D 2 ) and the switch should not be ignored. The boost converter, including inductor winding resistance (R L ), diode on resistance (R D ), diode forward voltage drop (V D ) and switch MOS on resistance (R on ), is as shown in Figure 5 8 (B) The boost converter can be modeled by analyzing two steady states of the circuit [81] : when the switch is on and when the switch is off. When the switch is on (closed), the diode is reverse biase d, and the inductor current (i L ) is flowing through the switch, as shown in Figure 5 9 (A) Similarly, when the switch is off (open), the diode is on. The inductor cu rrent is flowing through the diode and charge the capacitor, as shown in Figure 5 9 (B)

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112 A B Figure 5 8 PSM boost converter A) An ideal converter. B) A converter including conduction losses. Using the principles that dc components of the inductor voltage and the capacitor current are equal to zero, the two states in Figure 5 9 can be drawn together in Figure 5 10 where D is the duty cycle of the switching pulse. More detail description on how to combine two states into one circuit can be found in [81] The circuit model i s further simplified in Figure 5 11 with all resistors added together. Assuming the circuit works in CCM, the equivalent resistor (R eq ) and voltage source (V eq ) are

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113 A B Figure 5 9 Boost converter circuit A) When the switch is on. B) W hen the switch is off. Figure 5 10 Equivalent circuit model of the boost converter. Note that the simplified boost converter model is only valid for continuous conduction mode (CCM), where the inductor current never goes to zero during switching. In the discontinuous mode (DCM), that the i nductor current may goes to zero

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114 for some period of time at each switching cycle, the model is much more complicated, and beyond the scope of this work. Figure 5 11 Simplified Equivalent circuit model of the boost converter. 5.1.3 Load Model Depending on the type of electronic load, the load in an energy harvesting system can be modeled as an equivalent resistor, a constant current sink, or even a constant voltage source. An equivalent resisto r can represent a purely dissipative load when the behavior of storage elements is not considered. A current sink representation is limited to applications where the load has constant current dissipation and its impedance is much larger than the circuit o utput impedance. When a rechargeable chemical battery is included in the system, the output voltage remains relatively stable; therefore a constant voltage source can represent an ideal battery. However, a real battery model [82] is far more complicated than a constant voltage source, because the actual battery voltage is dependent on the stored energy. In this work, a resistor is used to model the load. 5.2 Model Parameter Extraction 5.2.1 Electrodynamic Energy Harvester Parameters A resonant electrodynamic energy harvester is used as an example for parameter extraction and measurement. The side view and top view photos are shown

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115 in Figure 5 12 with outline dimensions marked. The harvester has a typical cantilever beam structure: an aluminum cantilever beam (0.6 mm thick) with two NdFeB magnets attached at its tip that serve as the proof mass. A copper coil (AWG 28 copper magnet wire) with ~400 turns surrounds the magnet but is fixed to the base frame, which is made from a 3D printer. In operation, the base is vibrated, and the magnets on the cantilever tip move relative to the coil. A B Figure 5 12 Photos of the resonant electrodynamic energy harvester prototype A) S ide view photo. B) T op view photo. (Photos courtesy of Yuan Rao) Using this electrodyn amic harvester, the parameters in the lumped element model ( Figure 5 13 ), including the mass coil inductance coil resistance spring constant transduction coefficient and damping coefficient are extracted through a series of experiments, as discussed below. Figure 5 13 LEM model of electrodynamic energy harvester

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116 5.2.1.1 Spring Constant ( ) The spring constant (also called stiffness) is calculated by measuring the dis placement of the tip magnet when a mechanical force is applied, with the equation of Therefore, the slope of the curve between the force and the displacement yields the mechanical spring constant of the beam. The test setup includes a xyz micropositioner base, a laser displacement sensor (Keyence LK G32) and a digital force gauge (Imada DS2), as shown in Figure 5 14 Before the measurement, both the laser displacement and the force gauge are calibrated to make sure the accuracy of the result. Then the harvester is fixed on the xyz base, whereas the force gauge i s placed vertically on top of the harvester. The initial position is marked when the force gauge tip touches the tip magnet of the harvester, but without noticeable displacement. When the tip magnet is pressured down slowly, the force reading varies at dif ferent displacements. A B Figure 5 14 Test setup for spring constant (A) Calibration (B) Measurement

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117 A curve of displacement vs. force is plotted, as shown in Figure 5 15 over a displacement of 4 mm The slope of the best fit linear line gives the mechanical spring constant of 1390 Nm 1 The beam stiffness is only tested i n one direction (downward), but the stiffness is assumed to be equal in t he opposite (upward) direction. Figure 5 15 Measured mechanical force at different displacement 5.2.1.2 Transduction Coefficient ( ) The electrodynamic transduction coefficient is directly measured by applying a dc current to the coil and measuring the resulting force. When the current versus force curve is generated, the transduction coefficient is equal to its best fit slope, because the relationship between transduction coefficient ( ), mechanical force ( F ) and injected dc current ( ) is The experi mental setup is similar to the spring constant measurement, as shown in Figure 5 16 where the harvester is fixed to a stable table to avoid any noticeable displacemen t caused by external vibrations. A force gauge (Imada DS2) is positioned above the harvester in contact with the tip magnet to measure the resulting upward force. A dc current is applied to the coil from a Keithley 2400 source meter. The corresponding mech anical force is recorded for currents ranging from 10 mA up to 160

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118 mA, up limited by the linear tip displacement range of the beam. The data is shown in Figure 5 17 with the slope of the best fit linear line yielding a transduction coefficient of 3.36 NA 1 Figure 5 16 Test setup of the transduction coefficient. Figure 5 17 Measured mechanical force at different dc current 5.2.1.3 Damping coefficient ( ) The damping coefficient is ususally derived by the damping ratio which provides a mathematical means of expressing the l evel of damping in a system relative to the critical damping of a second order system. The relationship between them is

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119 where the corresponding critical damping coefficient b c is defined as Since the spring constant ( has been measured natural frequency ( ) are known, the damping coefficient can be calculated by For an under damped system in the time domain the damping ratio can be found expe rimentally by the logarithmic decrement method. The logarithmic decrement is the natural logarithm of the ratio of the amplitudes of any two successive peaks in displacement in the transient step response of the system Generally, peaks separated severa l periods away are used for better estimation, such that where is the amplitude at time is the amplitude of the peak periods away, and is the number of successive positive peaks. The damping ratio is then calculated as I the logarithmic decrement method becomes less accurate as the damping ratio increases (i.e. ). When the system is over damped (i.e. ), this method is no longer valid. In actual measurements with the electrodynamic harvester the decay of the mechani cal transient response of the harvester is estimated by measuring the decay of

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120 the electrical voltage waveform across the two terminals of the coil The result is valid only with the assumption that is constant and therefore the electrical voltage is eq ually proportional to the mechanical displacement The harvester is open circuited to avoid any electrical influence on the mechanical property. A n initial deflection is applied to the tip magnet and then removed suddenly ( flick test ), initiating a step response on the vibrating structure. The induced voltage waveform across the coil is measured by an oscilloscope (Agilent DSO X 2004A), as shown in Figure 5 18 From t he transient waveform, the damping ratio is estimated by using Equation (5 1 4 Note that n is chosen to be large (n=20) for error reduction. Figure 5 18 Resulting transient waveform from the flicker test Using the above method, the damping ratio is calculated to be 0.26 and yielding of 0.01 1 is obtained by Equation (5 1 5 ). The natural frequency can also be derived from the measurement, since where is the damped natural frequency and is the measured period of the transient waveform. From the measured waveform, is 0.024 s and thus is 260. The

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121 underdampe d natural frequency is therefore 41 Hz. Knowing the damping ratio ( ) spring constant ( and natural frequency ( ) the damping coefficient can be calculated by Equation (5 1 3 ), which is 0.113 for the test harvester. 5.2.1.4 Mass ( ) The mass in the harvester model refers to the effective mass of the structure, including not only the mass of the magnet, but also the mass contributed by the beam, There are two ways to measure the mass of the magnet. One way is measure the mass of the magnet and the beam using a digital balance before the harvester is assembled. When the tip magnet mass is much larger than the beam mass, the effective mass can be defined as mass [83] In most cases, due to the large mass of the magnet, the mass of the beam can be neglected. The effective mass is therefore equal to the mass of the magnet only, as A more accurate approach, however, is to derive the effective mass from its relatio nship with natural frequency and spring constant described by Since the spring constant and natural frequency are already known, the ef fective mass is calculated to be 0.02 kg.

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122 5.2.1.5 Harvester Model Summary So far all the parameters in the harvester transducer model have been extracted and summarized in Table 5 2 Table 5 2 List of extracted parameters of the electrodynamic harvester Name Parameter Value Unit Coil Resistance R c 18.7 Coil Inductance L c 17.3 mH Spring Constant k 1390 Nm 1 Transduction Coefficient K 3.36 NA 1 Damping coefficient b 0.113 NSm 1 Natural Frequency f n 41 Hz Mass m 0.02 kg Using these parameters, the open load voltage and the internal impedance of the harvester can be calculated by Equation (5 3) and Equation (5 4) as For example, when the input acceleration frequency is 41 Hz, the internal impedance can be calculated as Therefore, theoretically the maximum output power ( ) can be extracted from the harvester at the optimum load of

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123 where is the complex conjugate of In cases when the load is purely resistive, the maximum power is generated when is equal to the magnitude of which is about 52 when the acceleration frequency is 41 Hz. In the experi ment, the open load voltage at different input acceleration amplitudes is directly measured by exciting the harvester with vibrations The experimental setup is plotted Figure 5 19 The harvester is mounted on a mechanical shaker (LDS V480), which is driven by a function generator (Agilent 33120A) through a power amplifier (LDS PA100E). A digital oscilloscope (Agilent DSO X 2004A) monitors both the output voltage and the input acceleration amplitude. The acceleration amplitude is measured by an accelerometer (Model 356A16, PCB Piezotronics Inc.), together with a signal conditioner (Model 481A, PCB Piezotronics Inc.). Figure 5 19 Measurement setup of output voltage versus input acceleration amplitude Figure 5 20 plots both the calculation and the measurement result, where the output voltage increases almost linearly with increasing acceleration amplitude. The model calculation well matches the measurement result. The measured output voltage

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124 is up to 2.5 V when the input acceleration amplitude i s 1.7 ( ) and the acceleration frequency is 4 1 Hz. Figure 5 20 Open circuit harvester output voltage versus acceleration amplitude The harvester maximum power is also measured by directly connecting pure resistive load the two terminals of the coil at input acceleration with amplitude of 1 g and frequency of 41 Hz. As shown in Figure 5 21 t he measured maximum power (17.4 mW) occurs at around 5 0 load, close to our calculation result of 5 2 Figure 5 21 Output power versus load resistance The natural frequency corresponds to the frequency at which the maximum open circuit voltage occurs. As shown in Figure 5 22 the measured peak output voltage at open circuit load with 1.5 g input acceleration occurs at about 41 Hz, in agreement with model prediction.

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125 Figure 5 22 Open load harvester output voltage versus acceleration frequency at 1.5 g 5.2.2 Interface Circuit Parameters 5.2.2.1 Ac/dc Converter Parameters The parameter extraction of the interface circuit model is much easier compared with the parameter extraction of the harvester, because there are only tw o parameters required: the comparator offset voltage (V D ) and the turn on resistance (R D ). V D (<10 mV) is much less than the operational input voltages (>1 V), and therefore can be ignored. The simulati on result of resistance R D is shown in Figure 5 23 where R D PMOS is fully turned on (i.e. V GS is between 1 V and 3V). The one to one correspondence between R D and V GS is used in the half wave ac/dc converter model. Figure 5 23 Parameter extraction of PMOS turn on resistance R D

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126 To validate the ac/dc converter model, t he output voltages are compared between the model simulation and the measurement, as shown in Figure 5 24 where the input is a 20 Hz sine wave with voltage amplitude from 1 V to 3 V. The simplified model gives a decent approximation of the real world measurement, with a deviation of less than 7% of the input amplitude. The deviation is caused by the limited accuracy of the SPICE model compared with real measurement. When the input amplitude increases, the deviation increases due to the larger ignored par asitic losses at large current. Figure 5 24 Interface circuit output voltage for model validation 5.2.2.2 Boost Converter Parameters The parameters of the boost converter model include the inductor (L), the inductor winding resistance (R L ), diode on resistance (R D ), diode forward voltage drop (V D ), switch MO S on resistance (R on ) and the duty cycle (D). The inductor and inductor resistance is predetermined once the inductor is chosen. The dc/dc controller utilizes PSM control and therefore the duty cycle of the switching pulse is fixed by the circuit design. A commercial Schottky diode (NSR0320, On Semi) is in used in the dc/dc converter, which has an estimated on resistance and forward voltage drop available in

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127 the datasheet. The on resistance R on of the switch NMOS is simulated at different gate to source vol tage (V GS ), as shown in Figure 5 25 R on NMOS is fully turned on (i.e. V GS is from 1 V to 3V). Similarly to the PMOS switch in the ac /dc converter model, the one to one correspondence of R on and V GS can be used for the boost converter model. Table 5 3 summaries the parameters used in the boost converter model. Figure 5 25 Parameter extraction of NMOS turn on resistance R on Table 5 3 List of parameters of the boost converter Substituting these parameters the dc/dc converter model is simulated and then compared with the real measurement using 1.5 V dc input. The result is shown in Figure Name Parameter Value Unit Source Impedance R g 50 Inductor L 22 Inductor Winding Resistance R L 100 Diode On Resistance R D ~20 Diode Voltage Drop V D 0.24 @10mA V Duty Cycle D 0.5 -

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128 5 26 At light load (i.e. the load current is small), the error largely depends on the the power lo sses induced by parasitic resistance and capacitors of MOS transistors are all ignored in the simplified models. A t heavier load s (i.e. the load current is larger) the model deviates further away from the measurements. This is because when the load curren t increases, the dc/dc converter is closer to the boundary between CCM and DCM. The model becomes less accurate as the inductor current keeps at zero for a longer period of time at each switching cycle. Figure 5 26 Model simulation and measurement result of the dc/dc converter 5.3 Energy Harvesting System Modeling The electrodynamic harvester model, the interface circuit model and the load model, as well as their parameter extraction, ha ve been discussed thoroughly in the previous sections. The entire system model, which combines all the reduced order models, is shown in Figure 5 27 For simplicity, only a voltage doubler model is included in the ac/dc stage as the half wave converter since the input power supply is not in the main signal path. R D1 and R D2 represent the on resistances of the positive side diode

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129 (D 1 ) and negative side diode (D 2 ), respe ctively. At the back end of the overall system, a resistor R LOAD models the constant resistive load. Figure 5 27 Complete energy harvesting system model As mentioned before, the ability to develop an equivalent circuit model makes it possible to simulate the entire system in circuit simulation tools, allowing a convenient performance analysis of the system. To validate the modeling strategy, the system model is simulated in Cadence Spect re SPICE, using the parameters extracted previously. Meanwhile, the prototype system was measured by putting the harvester on top a shaker with an input acceleration of 1 g at the system resonance of 41 Hz. At open circuit load, a sine wave with 1.5 V ampl itude is first measured. The harvester is then connected to the interface circuit and a resistor load, while maintaining the same input acceleration. The measured output voltage and power on the load are compared against the SPICE simulation results, a s shown in Figure 5 28 B oth the simulation and the measurement results follow exactly the same trend. Although the voltage mismatch in between is about 10% to 20% of the measurement result, possibly due to the limited accuracy of the interface model and the SPICE models, the model gives a decent prediction of the measurement result

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130 A B Figure 5 28 Comparison of measurement and model simulation result of entire system at various load. A) Output voltage B) Output power In Figure 5 28 (B), the model simulation result show s a peak power delivered with 500 optimum load, and t he same result is obtained experimentally. This optimum resistive load is quite different from the optimum resistive load of 5 2 from Equation (5 21) The reason is that the interface circuit changes the net output impedance of the system (i.e. harvester + interface circuit). Based on the experimental measurements, it can be inferred that the effectiv e output impedance of the enti re energy harvesting system is ~ 500 Ohm. Hence the circuit adds an additional ~ 450 Ohm parasitic loss to

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131 the system. Based on the impedance of the interface circuit and its associlated power losses, the maximum system output power in Figure 5 28 (B) is 3.3 mW, much lower than the maximum har vester output power of 17.4 mW in Figure 5 21 In addition, the system model is validated with different input acceleration amplitude at 41 Hz. Figure 5 29 shows the measurement and model simulation result or higher. Both the output voltage and power increase with the increasing input acceleration amplitude in simu lation and measurement results. The voltage error, however, is up to about 20%. For example, the measured output voltage is about 2.46 V at 1.5 g input, compared with 2.92 V from the model simulation. The deviation is a consequence of the limited accuracy of the simplified PSPICE model, because all the circuit components are replaced with the ideal ones. Moreover, the nonlinearities of the harvester parameters, such as the damping coefficient, spring constant, and transduction coefficient, may also contribu te to the difference between the actual measurement and model prediction. 5.4 Summary An equivalent circuit model for the resonant electrodynamic energy harvesting system is developed in this chapter. The system model comprises three reduced order models: an electrodynamic harvester model, an interface circuit model and a load model. The re duced order models are first introduced and parameter extraction methods are discussed in detail with examples. Then the entire system model is given and verified by experimental results. The model simulation and real measurement result of an electrodynami c energy harvester system are shown in close agreement under certain conditions. Since the system varies significantly with different har vester and

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132 interface parameters, t he model provides a straightforward way to predict the optimal system conditions even before the entire system is measured. Therefore, the ability of using equivalent circuit model to represent the multi domain energy harvesting system makes the system level simulation and analysis feasible and convenient using circuit simulation tools. A B Figure 5 29 Comparison of measurement and model simulation at various inp ut acceleration amplitude. A) Output voltage B) Output power

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133 CHAPTER 6 NON RESONANT E NERGY H ARVESTING SYSTEM FOR H UMAN MOVEMENTS There is a large amount of mechanical energy created by people in their daily lives, such as walking, jogging, cycling, climbing stairs, tapping a foot or even breathing. If effective and self contained energy harvesting systems were availa ble, this freely available human energy may be effectively harnessed and put to good use. The previously described resonant energy harvesting system (Chapter 5) is not well suited for low frequency and multi dimensional human movements, because it responds to only one dimensional vibrations over a narrow frequency range. This chapter presents a fully functional, self sufficient non resonant energy harvesting system for harvesting energy from human motions The system targets natural human movements as t he primary energy source, with the long term vision of supplying power to portable, wearable, or even implanted electronic devices. It features a unique omnidirectional, electrodynamic (magnetic) energy harvesting transducer along with the input powered in terface circuit, which together charge s a thin film rechargeable battery from human movements. The complete system is implemented, demonstrated, and characterized using real human activities, including walking, jogging, and cycling. The system is shown to successfully generate electrical energy from these human induced movements, convert the induced ac voltage to a dc voltage, and then boost and regulate the dc voltage to charge the battery. The interface circuit has been discussed in Chapter 4 and will not be repeated here. The remainder of this chapter is organized as follows. First the detail system design of the harvester and the energy storage are provided, followed by the complete

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134 system prototype implementation. Then, the system characte rization and demonstration are presented. In the end, the summary is made. 6.1 System Design Figure 6 1 presents the block diagram of the energy harvesting system. The sys tem consists of an omnidirectional electrodynamic (magnetic) energy harvester, the input powered interface circuit (Chapter 4) and a Li ion polymer rechargeable battery. The kinetic energy from human movements is converted to electrical energy through the harvester, producing a quasi chaotic time varying voltage. However, the output voltage from the harvester is relatively low and not able to charge the battery directly, so it is fed to the interface circuit, which rectifies and regulates the input to a con stant dc voltage (3.7 V) to charge the battery. By using input powered interface circuit (Chapter 4), the complete system does not require external power supply .Moreover, the system in the absence of vibrational inputs and input stimulations. Figure 6 1 Block diagram of the self sufficient energy harvesting system 6.1.1 Energy Harvester Compared with mechanical vibration sources, human induced motions are challenging for energy harvesting design because of their low frequency (1 10 Hz), aperiodic, and time varying characteristics. The commonly used high Q resonant type

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135 energy harvesters, which are based on an under damped, single degree of freedom, mass spring damper system, are generally not well suited for human movement energy harvesting [8]. The reason is that these resonant systems are optimized to achieve maximum output power within a small frequency range under one dimensional, oscillatory accelerations. Moreover, it is difficult to tune the resonant frequency to the low frequencies of human motions (1 10 Hz) and maintain high quality factor, especially all while maintaining compact device dimensions. Another major restriction of conventional resonant harvesters is that they are typically designed for only one rectilinear degree of freedom, while normal human movements occur in three dimensions and involve a high degree of rotational, rather than oscillatory, motions. The above challenges motivate the use of a non resonant harvester architecture, which can respond over a broad range of vibration frequencies and amplitudes, and the use of a multi directional architecture that can respo nd to motions in multiple axes. Our research group ha s previously reported a unique, omnidirectional electrodynami c energy harvester design [65] which is replicated with modifications in the system reported here. As shown in Figure 6 2 the harvester structure is fabricated in two symmetric hemispheres using a Nylon plastic material from a 3D printer Both the upper and the lower half are wrapped with ~1400 turns of 34 AWG copper wire with a (diameter=3 cm) with a permanent magnet ball (diameter=1.27 cm, Grade N40 NdFeB) inside. The harvester has a total volume of about 39 cm 3 and weighs 6 8 g. In operation, the magnet ball moves chaotically within this spherical housing when subjected to external vibrations/motions. The motion of the magnet ball induces a

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136 time varying magnetic flux in each of the surrounding coils, thus generating a voltag e is connected a current will flow through the coil thus converting mechanical energy into electrical energy. In the experiments, the two coils are c ounter wound and connected i n series to impr ove power generation. A B Figure 6 2 Non resonant electrodynamic energy harvester A) Photo. B) 3 D schematic (Photos courtesy of Yuan Rao). Figure 6 3 shows the screenshot of the open circuit voltage waveform by gently hand shaking the constructed harvester. The voltage waveform is pseudo random with frequency content ranging from 1 6 Hz an d amplitude up to 3.5 V. Note that the voltage and frequency will change significantly with different input acceleration. Figure 6 3 Example open circuit output voltage waveform when hand shaking the harvester

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137 6.1.2 Energy Storag e The selection of a proper energy storage element depends on the application requirements, including consideration of size, safety, cost, performance, lifetime, environmental concern, etc. Rechargeable batteries and super capacitors have been the most co mmonly used energy storage elements in energy harvesting systems. Super capacitors are the most efficient and tolerant to temperature change, shocks, and vibrations, but their energy dens ity is lower than batteries [7] As a re sult, for the same capacity, super capacitors are usually larger than batteries. Therefore, in this design where the system size is limited, a rechargeable lithium ion (Li ion) battery is used as the energy storage element. A Li ion battery is chosen here not only because of its small size, lightweight, and good energy density, but also because it has no memory effect and slow self discharge rate. However, Li ion batteries should be handled with caution, especially in high temperatures, because they can ea sily ignite or explode. A commercial Li ion polymer rechargeable battery with nominal voltage of 3.7 V and maximum capacity of 65 mAh is used [84] The battery has a smaller size (23mm 12mm 4mm) and a longer life time (up t o 500 cycles charge/discharge) than conventional rechargeable cells. It comes with a self protection circuit to avoid over charge or over discharge, and therefore no additional battery management circuit is needed in the system design. 6.2 System Prototype Figure 6 4 depicts the prototype system, which has a cylinder structure with 6.3 cm height and 3.8 cm width, leading to a total volume of about 70 cm 3 and a total weight

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138 of 81 g. The interface chip, the discrete components, and the rechargeable battery are all assembled on two double sided circular PCBs with radius of 1.41 cm. Figure 6 4 Photograph of the system prototype (Photos courtesy of Yuan Rao). The interface chi ps, capacitors, diodes, and inductor are soldered on the top PCB (PCB1) as indicated in Figure 6 5 whereas the button battery and the rechargeable batter y are mounted on the back PCB (PCB2). These two US quarter size PCBs are mounted on top and bottom surfaces of the harvester to make the whole structure firm when exposed to external vibrations. (A) Top side view ( B ) Bottom side view Figure 6 5 Photograph of the double sided circuit PCB boards (Photos courtesy of Yuan Rao).

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139 voltage of 1.5 V is used to provi de the reference voltage (VREF) for the dc/dc controller, because there is no on chip reference circuit. Note that this battery is connected to the gate of a MOSFET, and therefore the power consumption from this battery is negligible. In future designs, th is battery can be replaced by an on chip bandgap reference circuit. 6.3 System Demonstration 6.3.1 Measurement Method The energy reclamation performance of the complete energy harvesting system is then measured when subjected to real human activities. In the experim ents, the for two reasons. The first reason is that more motion/vibrations are expected at these locations, and thus more energy can be harvested [85] Another reason is that to attach activities, which is important for future applications. Because different pe ople may have different gait s in daily activities, the ge nerated energy will vary from person to person. Therefore, in the experiment, the same person (me) is tested for all the activities, so that the result is comparable and fair among different locations and movements. The system energy reclamation is measure d for three types of movements: walking, jogging and cycling. Walking and jogging are carried on a treadmill, while cycling is performed on a stationary cycling machine. By doing so, the speed of each activity is accurately controlled and replicated. Each activity type is tested for 10 minute duration and is repeated by attaching the system to the different parts of the body. To quantify the harvested energy, the battery voltage is measured before the activity and at 1 minute intervals during the activity. Comparing the voltages to the separately

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140 as shown in Figure 6 6 the total energy delivered from the system to the battery is estimated. Figure 6 6 6.3.2 Delivered Energy Figure 6 7 (A) shows the estimated energy delivered to the battery versus time when the system is mounted on the an kle for jogging, walking and cycling. Comparing these three activities, the most accumulated energy of 142 mJ is delivered after 10 minute s of jogging. This result is within our expectation because jogging generates larger vibration accelera tions than walking and cycling. Similarly, Figure 6 7 (B) plots the energy delivered during jogging when the system is mounted at ankle, arm and wrist of the human body, among which the most significant energy of 142 mJ is detected at the wrist. Note that in both figures the energy delivered increases almost linearly with the time of human movements, because the speed of each human activity keeps constant. 6.3.3 Average Power The average harvested power can be estimated by calculating the slope of th e lines in Figure 6 7 Table 6 1 summarizes the measurement results. The avera ge power

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141 resulting a power density of 3. 3 3 Due to the relatively small upper b ody vibrations, no significant power is measured at arm during walking and cycling, and the same for the wrist during cycling. This is attributed to the energy harvester not generating sufficient ac voltage amplitude for the interface circuitry to function A B Figure 6 7 Energy delivered to the battery A) F rom human ankle B) F rom jogging

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142 Table 6 1 Measured average power delivered to the battery during human movements (Photos courtesy of Yuan Rao). 6.4 Summary An enormous number of research publications on energy harvesters and energy harvesting circuit interfaces have been reported, but much fewer studies combine harvesters and interface electronics to create fully functioning energy harvesting systems. For har vesting power from human movements, the list is even shorter. Therefore, in this chapter, a complete self sufficient energy harvesting system including a magnetic harvester, an integrated self powered interface circuit, and a rechargeable battery is demons trated and characterized. The harvester employs spherical magnetic harvester structure, favorable for harvesting multi directional vibrations from human movements. The interface circuit includes both an ac/dc stage and a dc/dc stage, and provides a regul ated dc output voltage for battery charging. Additionally, the input powered feature of the interface circuit eliminates the standby power consumption of the system when there is no activity or the input vibration amplitude is too low for successful energy extraction. Because the Ankle Wrist Arm Jogging (4mph) Walking (2.5mph) N/A Cycling (22mph) N/A N/A

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143 input vibration energy is always intermittent, a Li ion rechargeable battery is used as the energy storage element. The system successfully scavenges and converts mechanical energy from ordinary human movements to electrical energy for charging a battery. The the battery during jogging when the system is mounted on the ankle. The total volume of the system is 70 cm 3 and therefore the net power density i s about 3. 3 3

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144 CHAPTER 7 CONCLUSIONS AND FUTURE DIRECTION S The main focus of this dissertation is to develop input powered energy harvesting circuits, including ac/dc converters and a dc/dc converter, for electrodynamic vibrational energy systems. The des ign considerations and experimental results of the interface circuit were explored in detail. The energy harvesting systems using these interface circuits were demonstrated and the power density was characterized in real applications. In this chapter, the research contributions are listed in the first section. The second section gives the summary of the work. Suggestions on the possible future directions are given in the final section. 7.1 Research Contributions The following is the list of the contributions of this research: Developed self sustaining input powered energy harvesting interface circuits including ac/dc converters and the dc/dc converter, intended for electrodynamic vibrational energy harvesters Compared different topologies of ac/dc converters, concluding the voltage doubler offers advantages for vibrational energy harvesters. Developed a complete energy harvesting system model for resonant type vibrational energy harvester with interface circuits and load, and validated this model experimentally. Fabricated a complete, fully functional, self contained energy harvesting system and demonstrated successful energy reclamatio n from real human movements. 7.2 Summary of Research There has been a significant increase in the research on devices and circuits for harvesting vibrational energy in recent years, such as harvesting mechanical energy

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145 from human induced vibrations. For these vibrational energy harvesters, most work has been focused on the electromechanical energy conversion, i.e. the transducer. To realize function systems, however, small scale, efficient, self powered interface circuits are also important. For harvesting vi brational energy in many real world scenarios standby power of the interface circuits may drain the energy storage element and cause a startup problem after a long idle time if there is a long interval between two harvesting cycles. To solve this problem was developed on ac/dc converter s (Chapter 2) and a close loop dc/dc converter (Chapter 3). Then they were combined to form a complete input powered interface circuit (Chapter 4) All designs were implemented on silicon with On Semi 0.5 m CMOS fabrication process and bench top characterized using a 20 Hz sine wave from function generators. To provide a 3.7 V regulated output voltage the complete interface circuit functions with input amplitude ranges from 1.2 V to 3.0 V wit h maximum efficiency of 61 % at input amplitude of 2. 6 V When the input voltage amplitude drops below 0.6 V, the complete circuit automatically enters standby mode with no standby power consumption. Compared with state of the art vibrational energy harvest ing circuits [30 40], the input powered interface circuit achieves zero standby power through simple circuit implementation, while maintaining reasonable efficiency. The deficiency of the input powered design however, is the limited minimum input voltage due to the required r ail rail voltage of the circuit, which constrains the application of the circuit. A resonant electrodynamic (magnetic) energy harvesting system model was introduced (Chapter 5), providing an approach for better understanding of the sy stem.

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146 The energy harvesting system model is based on the reduced circuit model of each function block: a resonant electrodynamic transducer, an interface circuit (Chapter 4), and the load. To investigate the accuracy of the model, an electrodynamic energy harvesting system was built and characterized on a shaker. The system model with extracted parameters was simulated, and the result was compared with actual measurement result. The output voltage error is within 20% with the input acceleration amplitude be tween 0.7 g to 1.5 g. Finally, a fully functional, self sufficient motional energy harvesting system was demonstrated (Chapter 6). It features a unique non resonant electrodynamic transducer and a complete input powered interface circuit (Chapter 4) which together charge a thin film rechargeable battery from human movements. The system is powered by the input energy and thus allows the system to automatically turn on/off depending on the input vibration level. The complete system was implemented, de monstrated, and characterized on human body using real human activities, including walking, jogging, and cycling. The system successfully scavenges and converts mechanical energy from daily human movements to electrical energy for charging a 3.7 V battery. A maximum average power delivery of 234 jogging when the system prototype is mounted on the ankle and therefore the net power density is about 3. 3 4 3 7.3 Future Direction s The applications of energy harvesting systems are hig hly constrained by the power they can supply. In most situations, the power densities of energy harvesting systems are for too small to meet the power density demands of today

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147 electronic devices. Therefore, future directions of this research inc lude continued reduction of the system volume and continued optimization of the output power. To reduce the system volume, both the transducer and the electronic circuits should be shrunk in size. The magnetic transducer size is lower limited by t he minim um operation input voltage of the circuit Therefore by migrating the circuit to a more advanced CMOS technology the voltage requirement for the harvester is less restricted, allowing a smaller harvester size in the system. For example, the minimum input threshold of ac/dc converters in this work is around 0. 7 V, which is determined by Meanwhile, by using adv anced twin well process, the entire interface circuit, including the ac/dc converters and the boost converter, can be implemented on the same substrate inside a single die, thereby a smaller circuit size can be expected. Similarly, both the transducer and the electronic circuits can be further optimized to achieve higher output power of the system. The electrodynamic transducer design can be improved by optimizing the structure and the coil windings. The interface circuit can extract more power from the har vester if the minimum input threshold voltage is further reduced Furthermore, impedance matching technique can be implemented to achieve ma ximum output power of the system, especially when the transducer type and the load conditions have been specified. T he maximum power point tracking (MPPT) technique [68] [69] [86] provides a possible way in future vibration energy harvesting field, to match the impedance between the harvester source and the interface

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1 48 electronics, through adjusting parameters of the circuit, such as the switching frequency and duty cycle of the dc/dc converter. Above all, from the application point of view, to compete with widely used batteries, the bo ttle neck of this research is the relatively large size and small output with the continuous effort on optimizing the system, including reducing the size and maximizing the output power, special applications may be motivated in the future, such as wireless sensors or biomedical devices.

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149 APPENDIX A PUBLICATIONS Powered Active AC/DC Converter With Zero Standby Power for Energy Harvesting 4446, 2010. Powered Energy Harvesting Interface Circuits 1999, 2011. Powered Energy Harvesting Interface Circuits with 3524 3533, 2011. pp. 2844 2851, 2013. Y Sufficient Energy Harvesting 101 104 2012.

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150 APPENDIX B CHIP BONDING DIAGRAM Figure B 1. Bonding diagram of the voltage doubler chip T able .B 1 Pin list of the voltage doubler chip Pin # Pin Name Comment 1 NA NA 2 VSS_N Negative side doubler VSS 3 VDD_N Negative side doublerVDD 4 NA NA 5 VSS_P Positive side doubler VSS 6 VDD_A Auxiliary converter VDD 7 VCMP_A Comparator output of Auxiliary converter 8 VDD_P Positive side doubler VDD 9 VOUT_A Auxiliary converter output 10 VOUT_P Positive side doubler output 11 VCOMP_P Positive side comparator output 12 VIN_P Positive side doubler input 13 VCOMP_N Negative side comparator output 14 VIN_N Negative side doubler input 15 VOUT_N Negative side doubler input 16 NA NA

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151 Figure. B 2 Bonding diagram of dc/dc converter chip

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152 Table. B 2 Pin list of the dc/dc converter chip Pin # Pin Name Circuit Function Comment 1 VOUT1 Half wave ac/dc Converter1 output 2 POS_VDD1 Converter1 VDD 3 VIN Converter input 4 POS_VDD2 Converter2 VDD 5 VOUT2 Converter2 output 6 NA NA NA 7 8 9 VSS Dc/dc VSS 10 VDN1 M1 drain 11 VGN1 M1 gate 12 VG M2 gate 13 VD M2 drain 14 BIAS OSC bias 15 SW M2 gate 16 VDD2 Levelshifter VDD2 1 7 OSC Oscillator output 18 LSHIFT Levelshifter output 19 INV Inverter output 20 COMP Comparator output 21 VDD1 Levelshifter VDD1 22 VREF Reference voltage 23 FB Feedback voltage 24 EN Enable voltage

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160 BIOGRAPHICAL SKETCH Yuan Rao was born in Jiangxi, China. She earned her BSEE and MSEE from Zhejiang University, Hangzhou, China. From 2004 to 2007 s he was a product engineer in Shanghai, China before she came to US in 2008. From 2008 to 20 13 s he was a research assistant of the University of Florida with h er research focused o n developing energy harvesting circuits and systems In the summer of 2013, she received her PhD from the Uni versity of Florida, Gainesville, FL Since 2013 she has been with Texas Instruments as an analog circuit designer working on power converter desig n for consumer electronics.