60-GHz CMOS Micro-Radar System-in-Package for Vital Sign and Vibration Detection

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Title:
60-GHz CMOS Micro-Radar System-in-Package for Vital Sign and Vibration Detection
Physical Description:
1 online resource (109 p.)
Language:
english
Creator:
Kao, Te-Yu
Publisher:
University of Florida
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Gainesville, Fla.
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Thesis/Dissertation Information

Degree:
Doctorate ( Ph.D.)
Degree Grantor:
University of Florida
Degree Disciplines:
Electrical and Computer Engineering
Committee Chair:
Lin, Jenshan
Committee Members:
Eisenstadt, William R
Gu, Qun
Rice, Jennifer Anne

Subjects

Subjects / Keywords:
antenna -- cmos -- heartbeat -- ic -- packaging -- radar -- respiration -- rf
Electrical and Computer Engineering -- Dissertations, Academic -- UF
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Electrical and Computer Engineering thesis, Ph.D.
bibliography   ( marcgt )
theses   ( marcgt )
government publication (state, provincial, terriorial, dependent)   ( marcgt )
born-digital   ( sobekcm )
Electronic Thesis or Dissertation

Notes

Abstract:
The dissertation begins with basic concepts of Doppler radar and motivations of 60-GHz design. Compared to previous works at lower frequencies, the benefits of shorter wavelength are explained by theoretical analysis. State-of-the-art Doppler radar systems are discussed. Chapter 1 also lists challenges associated with high operating frequency such as CMOS circuit performance and loss, package and antenna transition, and strong non-linear Doppler phase modulation from both hardware and signal processing points of view.  Chapter 2 describes the system design considerations such as receiver architecture, sensitivity, and layout floor plan. The detail of each circuit block is investigated in Chapter 3, introducing inductor EM modeling, RF transceiver front-end, IF stages, and flip-chip integration with PCB patch antennas. Chapter 4 shows the experimental results including on-wafer measurement and antenna tests. Detection of small mechanical vibration and human vital sign are successfully demonstrated. In Chapter 5, theoretical analysis is provided in detail to explain the difficulties vital sign detection at 60 GHz. A detection technique monitoring both the fundamental and second harmonic of respiration is proposed to improve the detection accuracy of respiration. In addition, a time-domain signal recovery algorithm is proposed and tested to help the detection of target movement comparable to wavelength. Finally, a circularly polarized sequential-rotation antenna array is implemented on LTCC (low-temperature co-fired ceramic) to increase the antenna bandwidth. As the process and manufacturing variations are often present in mm-wave systems, wide antenna bandwidth is able to cover the possible frequency drift and increase the system yield in mass production.  The summary is provided in Chapter 6. This work demonstrates the first vital sign detection by the flip-chip-integrated CMOS micro-radar at 60 GHz. The shorter wavelength offers significant area reduction and flexibility in system integration. The compact, low-cost CMOS system can be embedded in portable devices such as the smart phone and tablet for daily healthcare and vibration monitoring, as well as deployed in a large sensor network for many other applications.
General Note:
In the series University of Florida Digital Collections.
General Note:
Includes vita.
Bibliography:
Includes bibliographical references.
Source of Description:
Description based on online resource; title from PDF title page.
Source of Description:
This bibliographic record is available under the Creative Commons CC0 public domain dedication. The University of Florida Libraries, as creator of this bibliographic record, has waived all rights to it worldwide under copyright law, including all related and neighboring rights, to the extent allowed by law.
Statement of Responsibility:
by Te-Yu Kao.
Thesis:
Thesis (Ph.D.)--University of Florida, 2013.
Local:
Adviser: Lin, Jenshan.
Electronic Access:
RESTRICTED TO UF STUDENTS, STAFF, FACULTY, AND ON-CAMPUS USE UNTIL 2014-05-31

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UFRGP
Rights Management:
Applicable rights reserved.
Classification:
lcc - LD1780 2013
System ID:
UFE0045145:00001


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1 60 GHz CMOS MICRO RADAR SYSTEM IN PACKAGE FOR VITAL SIGN AND VIBRATION DETECTION By TE YU KAO A DISSERTATION PRESENTED TO THE GRADUATE SCHOOL OF THE UNIVERSITY OF FLORIDA IN PARTIAL FULFILLMENT OF THE REQUIREMENTS FOR THE DEGREE OF DOCTOR OF PHILOSOPHY UNIVERSITY OF FLORIDA 2013

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2 2013 Te Yu Kao

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3 To my p arents and fianc e

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4 ACKNOWLEDGMENTS I would like to express my sincere appreciation to my advisor Dr. Jenshan Lin for his guidance, mentoring, and encouragement throughout my Ph.D. life I am also tha nkful to my Ph.D. committee mem bers, Dr. William Eisenstadt Dr. Jane Gu and Dr. Jennifer Rice for the valuable advice and feedback from many different points of view Bein g a member of Radio Frequency Circuits and Systems Research (RFCSR) group at UF Dr. Lin has al ways been so patient and supportive t o every one of us, providing a very helpful and friendly environment for graduate study. I w ould also like to thank my best colleagues Dr. Yan Yan, Dr. Xiaogang Yu, Dr. Tze Min She n, Dr. Austin Chen, Dr. Minqi Chen, Gabriel Reyes, Jianxuan Tu, Chien Ming Nieh, Ron Chi Kuo, Taesong Hwang, Raul Chinga, Changyu Wei, and Jaime Garnica for the us eful discussion and brainstorming to gether in the lab. In addition, I am thankful to our previous group member Prof Changzhi Li from T exas Tech University for the helpful guidance and discussion. Finally, I would also like to thank Dr. Kenneth O, Dr. Dongha Shim, Dr. Ning Zhang, Dr. Chuyin g Mao, Dr. Hsin Ta Wu, Dr. Wuttichai Lerdsitsomboon, Dr. Tie Sun Rounan Han, Dr. Chieh Lin Wu, Jason Seol, and Yang Hun Yun for their guidance and help in the early stage of my Ph.D. life. For the fabricati on of the 60 GHz micro radar system in package, I would like to acknowledge the single die bumping and flip chip process sponsored by Mr. Terence Collier from CVInc, Ri chardson, Texas, USA. We also acknowledge the 90 nm CMOS chip fabrication by United Microelectronics Corporation (UMC), Hsin Chu, Taiwan R.O.C., and the microwave laminates provided by Rogers Corporation, Rogers, CT, USA.

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5 TABLE OF CONTENTS page ACKNOWLEDGMENTS ................................ ................................ ................................ ............... 4 LIST OF TABLES ................................ ................................ ................................ ........................... 7 LIST OF FIGURES ................................ ................................ ................................ ......................... 8 LIST OF ABBREVIATIONS ................................ ................................ ................................ ........ 12 ABSTRACT ................................ ................................ ................................ ................................ ... 14 CHAPTER 1 INTRODUCTION ................................ ................................ ................................ .................. 16 1.1 Millimeter wave Doppler Radar in CMOS ................................ ................................ .. 16 1.1.1 Doppler Radar ................................ ................................ ................................ ... 16 1.1.2 System Implementation ................................ ................................ .................... 19 1.2 Vibration Detection and Quadrature Architecture ................................ ....................... 21 1.2.1 Optimal and Null Detection Points ................................ ................................ ... 21 1.2.2 Complex Signal Demodulation ................................ ................................ ........ 23 1.3 Vital Sign De tection ................................ ................................ ................................ ..... 23 1.4 Millimeter wave Packaging and Integration ................................ ................................ 26 2 SYSTEM DESIGN AND INTEGRATION ................................ ................................ ........... 29 2.1 Overview ................................ ................................ ................................ ...................... 29 2.2 Sensitivity and Radar Received Power ................................ ................................ ........ 30 2.3 Design Consideration for IF Stage ................................ ................................ ............... 32 2.4 Floor Plan and Flip Chip Transition ................................ ................................ ............ 33 3 CIRCUIT COMPONENT DESIGN ................................ ................................ ....................... 34 3.1 Inductor ................................ ................................ ................................ ........................ 34 3.2 Radar Receiver Front end Design ................................ ................................ ................ 37 3.2.1 LNA ................................ ................................ ................................ .................. 37 3.2.2 Active Mixer and RF VCO ................................ ................................ ............... 39 3.3 Radar Transmitter Front end Design ................................ ................................ ........... 41 3.3.1 Passive Balun ................................ ................................ ................................ .... 42 3.3.2 TX Driver ................................ ................................ ................................ ......... 43 3.4 IF Quadrature VCO and Passive Mixer ................................ ................................ ....... 44 3.5 CMOS Radar Chip Overview ................................ ................................ ...................... 46 3.6 Flip chip Integration and PCB Patch Antenna ................................ ............................. 47 3.6.1 Transition Design and Impedance Match ................................ ......................... 48 3.6.2 Patch Antenna ................................ ................................ ................................ ... 51

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6 4 EXPERIMENTAL RESUTLS ................................ ................................ ............................... 53 4.1 Millimeter wave CMOS Tra nsceiver Measurement ................................ .................... 53 4.2 IF Quadrature Ring VCO ................................ ................................ ............................. 54 4.3 Patch Antenna Test ................................ ................................ ................................ ...... 56 4.4 Radar Transmitted Power Test ................................ ................................ ..................... 59 4.5 Mechanical Vibration Detection ................................ ................................ .................. 63 4.5.1 Quadrature Channel Test ................................ ................................ .................. 63 4.5.2 Sensitivity to Small Vibration ................................ ................................ .......... 64 4.6 Heartbeat and Respiration Detection ................................ ................................ ........... 65 5 ANALYSIS AND IMPROVEMENT ................................ ................................ ..................... 69 5.1 Analysis on 60 GHz Vital Sign Detection ................................ ................................ ... 69 5.1.1 Respiration Detection Improveme nt by Two tone Monitoring ........................ 71 5.1.2 Heartbeat Detection ................................ ................................ .......................... 75 5.2 Proposed Time domain Recovery Algorithm ................................ .............................. 75 5.2.1 Analysis on Quadrature Baseband Outputs ................................ ...................... 76 5.2.2 MATLAB Program Implementation ................................ .............................. 78 5.2.3 Experimental Results ................................ ................................ ........................ 81 5.2.4 Discussion ................................ ................................ ................................ ......... 83 5.3 Broadband Antenna on LTCC System in Package ................................ ..................... 86 5.3.1 Introduction ................................ ................................ ................................ ...... 86 5.3.2 Sequential Rotation Patch Antenna Array ................................ ........................ 88 5.3.3 Vital Sign D etection ................................ ................................ ......................... 93 SUMMARY ................................ ................................ ................................ ................................ ... 95 APPENDIX: MATLAB CODING OF TIME DOMAIN RECOVERY ALGORITHM ........... 97 LIST OF REFERENCES ................................ ................................ ................................ ............. 103 BIOGRAPHICAL SKETCH ................................ ................................ ................................ ....... 109

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7 LIST OF TABLES Table page 3 1 Simulated inductor performance at 60 GHz ................................ ................................ ...... 35

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8 LIST OF FIGURES Figure page 1 1 Illustration of typical vital sign detection using Doppler radar system. ............................ 17 1 2 A 5.8 GHz Doppler radar system integrated on a PCB with two 2 2 patch antenna arrays for TX and RX. ................................ ................................ ................................ ....... 19 1 3 The concept of small displacement detection at optimum and null points of Doppler radar sy stem. ................................ ................................ ................................ ...................... 22 1 4 Comparison of vital sign detection using different radar frequencies ............................... 25 1 5 Typical wire bonding packaging co nfiguration. ................................ ................................ 26 2 1 Block diagram of the 60 GHz CMOS micro radar system including transceiver chip in 90nm C MOS, TX and RX patch antennas, and flip chip integration. ........................... 29 2 2 Sensitivity estimation of the Doppler radar receiver ................................ ......................... 30 3 1 Sim ulation of a 1.5 turn, 95 pH inductor along with the surrounding gro und plane ........ 34 3 2 Microphotograph of the on chip inductors used in t he 60 GHz front end ........................ 35 3 3 Simulation of the isolation between two closely placed inductors as they are in the actual on chip situation. ................................ ................................ ................................ ..... 36 3 4 RX front end (60 GHz to 6 GHz) including the 5 stage LNA, s ingle ended mixer, and 54 GHz VCO (Bias and LO distribution details not shown). ................................ ..... 38 3 5 Microphotograph showing the cascode portion of the layout and vertical access to the power grid. ................................ ................................ ................................ ................... 38 3 6 54 GHz RF LO ge neration and distribution ................................ ................................ ...... 40 3 7 Microphotograph showing the 54 GHz LO distribution network from VCO to up and down convert mixers. ................................ ................................ ................................ .. 40 3 8 TX front end (6 GHz to 60 GHz) using the double balance d up convert mixer, balanced loads (balun), and three stage driver at 60 GHz. ................................ ................ 42 3 9 Lumped element modeled transformer balun with differential to single ended impedance conversion. ................................ ................................ ................................ ....... 42 3 10 IF stage (6 GHz to dc) including the quadrature ring VCO, IF LO buffers, and passive mixers. ................................ ................................ ................................ ................... 44

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9 3 11 Design of the delay cel ls (D 1 and D 2 in Figure 3 10) with two tuning mechanisms (V p and V b ). ................................ ................................ ................................ ........................ 45 3 12 Simulated four output phases (LO I +, LO I LO Q +, and LO Q in Figure 3 10) generated by the quadrature ring VCO. ................................ ................................ ............. 46 3 13 Microphotograph of the 60 GHz CMOS micro radar. ................................ ....................... 47 3 14 Flip chip transition design between the 60 GHz CMOS radar chip and PCB patch antennas on RT/duroid 5870 laminate. ................................ ................................ .............. 48 3 15 Impedance analysis of the transition at 55 GHz before and after the flip chip process. ... 49 3 16 Microphotograph of the flip chip a rea on RT/duriod 5870 surface ................................ ... 50 3 17 Simulated patch antenna s parameter after the flip chip packaging ................................ .. 51 3 18 Simulated patch antenna pattern after the flip chip packaging. ................................ ......... 52 4 1 Measured down conversion gain (60 GHz to 6 GH z) versus RF in put frequ ency ............ 53 4 2 Measured up conversion (6 GHz to 60 GHz) gain compression and P in (diff erential) versus P out (single ended ) curve ................................ ................................ ......................... 54 4 3 Measured single end ed output spectrum of the qua drature ring VCO .............................. 55 4 4 Antenna return loss (S 11 ) measurement. ................................ ................................ ............ 56 4 5 Patch antenna test structure with probing/flip chip G S G S G area zoomed i n. ............. 5 7 4 6 Probe based measurement setup for the broadside radi ation patterns .............................. 58 4 7 Radiation patte rns of th e single patch antenna ................................ ................................ .. 58 4 8 Measured and simulated realized gain spectrums at zenith. ................................ .............. 59 4 9 The final system configuration of the 60 GHz micro radar system in package in cluding the CMOS transceiver chip, two PCB patch antennas, an d dc biasing through blue wires. ................................ ................................ ................................ ............. 60 4 10 Experimental setup for the TX output power of CMOS transceiver chip. ........................ 61 4 11 Photo of the experimental setup to test TX transmitted power of CMOS micro radar chip. ................................ ................................ ................................ ................................ .... 61 4 12 The screenshot of the received power P r = 82.24 dBm on the spectrum analyzer ........... 62 4 13 I an d Q baseband outputs test of the micro radar system ................................ .................. 63

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10 4 14 The experimental results of small mec hanical vibration detection ................................ .... 64 4 15 Heartbeat detection using the 60 GHz radar when the target holds the b reath at 0.3 m away ................................ ................................ ................................ ................................ .. 66 4 16 Measurement results of the respiration detection at 60 GHz as f r = 15 beat/minute, D = 0.3, and m r is slightly varied in the two tests. ................................ ................................ 67 5 1 Theoretical plots of J n ( a ) = J n ( / ) versus vibration amplitude m ................................ 71 5 2 Simulated output s pectrum of vital s ign d etection at 60 GHz ................................ ........... 72 5 3 Vital sign detection results as the person breathes shallowly at 0.3 m in front of th e radar ................................ ................................ ................................ ................................ ... 73 5 4 Vital sign detection results as the person breathes deeply at 0.3 m in front of the rad a r ................................ ................................ ................................ ................................ ... 74 5 5 Non linear input output mapping when the vibration is comparable to at 60 GHz. ....... 76 5 6 Non linear input output mapping of I and Q channels when the vibration is comparable to at 60 GHz ................................ ................................ ................................ 77 5 7 Time domain recovery technique by simply monitoring I and Q baseband outputs when the vibration is comparable to at 60 GHz ................................ ............................. 78 5 8 Illustration of the continuous Flip and Follow operations. ................................ ................ 79 5 9 Simplified flo w chart of the time domai n recovery algor ithm. ................................ ......... 80 5 10 Respiration detection outputs before and after the recovery algorithm i s applied ........... 82 5 11 Recovered respiration peak compared to the orig inal spectrum in Figure 4 16 (B) .......... 82 5 12 Respiration detection outputs before and after the recovery algorithm is applied. The subject inhaled for 2 s, exhaled for 2 s, pause d for 3 s, and repeated the cycl e ................. 83 5 13 CSD spectrum outputs before and after the recovery algorithm is applied ...................... 84 5 14 Duplicate of Figure 5 13 showing the consecutive Follow period s indicated by t he reco very algorithm ................................ ................................ ................................ ............. 85 5 15 Spectrum of the waveform marked by bl ue circle in Figure 5 14 which shows the correct heartbeat rate detection result. ................................ ................................ ............... 85 5 16 Measured and simulated return loss of a single patch PCB antenna as the manufacturing variation is p resent. ................................ ................................ .................... 88

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11 5 17 Layer profile presents the LTCC system in package including 11 metal layers (L 1 L 11 ) and the FR4 board with a slot for CMOS chip. ................................ .......................... 89 5 18 Broadband anten na design on LTCC ................................ ................................ ................. 90 5 19 Measure d and simulated return losses (S 11 ) and TX/RX isolation (S 12 ) of the antenna array. ................................ ................................ ................................ ................................ .. 91 5 20 Realized gain and axial ratio (AR) spectrums of the broadband sequential rotation patch antenna ar ray at zenith. ................................ ................................ ............................. 92 5 21 Radiation patterns of the patch antenna a rray. ................................ ................................ .. 92 5 22 Top view of the fi nal system assembly The flip chip integrated CMOS radar chip and surface mounted bypass capacitors are placed on the other side of LTCC. ............... 93 5 23 CSD output spectrum of vital sign detection using the broadba nd patch antenna array on LTCC ................................ ................................ ................................ ............................ 94

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12 LIST OF ABBREVIATIONS Wavelength ADC Analog to digital conversion BGA Ball grid array BPM Beat per minute CMOS Complem entary symmetry metal oxide semiconductor CPW Coplanar waveguide CS Common source DSP Digital signal processing EM Electromagnetic FFT Fast Fourier transform f MAX Power gain cutoff frequency f T Unity current gain frequency IC Integrated circuit IF Intermed iate frequency LNA Low noise amplifier LO Local oscillator MEMS Micro electro mechanical systems MM wave Millimeter wave NF Noise figure P 1dB 1 dB gain compression point PA Power amplifier PCB Printed circuit board P sat Saturated output power of PA/driver in TX RX Receiver

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13 RF Radio frequency SGH Standard gain horn antenna SiP System in package SoC System on chip TX Tran smitter VCO Voltage controlled oscillator

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14 Abstract of Dissertation Presented to the Graduate School of the University of Florida in Partial Fulfillment of the Requirements for the Degree of Doctor of Philosophy 60 GHz CMOS MICRO RADAR SYSTEM IN PACKAGE FOR VITAL SIGN AND VIBRATION DETEC T ION By Te Yu Kao May 2013 Chair: Jenshan Lin Major: Electrical and Com puter Engineering The dissertation begins with basic concepts of Doppler radar and motivation s of 60 GHz design Compared to previous works at lower frequencies the benefits of short er wavelength are explained by theoretical analysis State of the art Do ppler radar system s are discussed Chapter 1 also lists challenges associated with high operating frequency such as CMOS circuit performance and loss package and antenna transition, and strong non linear Doppler phase modulation from both hardware and sig nal processing points of view Chapter 2 describes the system desi gn consid erations such as receiver architecture, sensitivity and layout floor plan. The detail of each circuit block is investigated in C hapter 3 introducing inductor EM modeling RF trans ceiver front end, IF stage s and flip chi p integration with PCB patch antennas. C hapter 4 shows the exp erimental results including on wafer measurement and antenna tests. D etection of smal l mechanical vibration and human vital sign are successfully demonst rated. In Chapter 5, theoretical analysis is provided in detail to explain the diff iculties vital sign detection at 60 GHz A detection technique monitoring both the fundamental and second harm onic of respiration is proposed to improve the detection accura cy of respiration. In addition, a time domain signal recovery algorithm is proposed and tested to help the detection of target moveme nt comparable to wavelength. Finally, a circularly polarized

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15 sequential rotation antenna array is impleme nted on LTCC ( low temperature co fired ceramic ) to increase the antenna bandwidth As the process and manufacturing variations are often presen t in mm wave systems wide antenna bandwidth is able to cover the possible frequency drift and increase the system yield in mass pr oduction The s ummary is provided in Chapter 6. This work demonstrates the first v ital sign detection by the flip chip integrat ed CMOS micro radar at 60 GHz. The shorter wavelength offers significant area reduction and flexibility in system integration. Th e compact low cost CMOS system can be embedded in portable devices such as the smart phone and tablet for daily healthcare and vibration monitoring, as well as depl oyed in a large sensor network for many other applications.

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16 CHAPTER 1 INTRODUCTION 1.1 Millimet er wave Doppler Radar in CMOS Recent progress on non contact vital sign and vibration detections have been made based on microwave Doppler radar system [ 1 ] [ 6 ]. Compared to other method s of detectio n such as laser based sensor s and interferometers [ 7 ] simple architecture of Doppler radar realized by integrated circuit and system usually makes it a cost effective and low power solution. It is also useful in various situat ions such as longer distance, low visibility and through wall detections [ 8 ] C urrent laser displacement sensor s [ 9 ] [ 11 ] offer excellent precision down to tens of nanom eter s at a high er cost and within short detection range (usually less than 1 m) and a measured d isplaceme nt of utilizing microwave interferometry and sophisticated signal processing methods [ 7 ]. Lately the frequency and amplitude of two tone sinusoidal vibration can be distinguished and accurately measured by using non linear Doppler phase modulation [ 8 ] [ 12 ]. 1.1.1 Doppler Radar Non contact vital sign detection has been found to be the important application of Doppler radar system and drawn increasing attention It proves to be a safe low cost, and effective wa y to monitor the heartbeat and respiration withou t physical contact A typical vital sign detection system using Dopp l er radar is shown in Fig ure 1 1 [ 3 ] where an un modulated signal T ( t ) is tra nsmitted with the amplitude normalized to unity : (1 1) where f and vco are the frequency and phase noise T ( t ) is reflected a nd phase modulated by the target displacement x ( t ) which is the chest wall movement in this case. The baseband output B ( t ) can be expressed as [ 13 ] :

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17 F igure 1 1. Illustration of typical vital sign detection using Doppler radar system. (1 2 ) where is wavelength of T ( t ) variable G s is defined as total system gain, and t i s the total residue phase accumulated in the circuit an d transmission path. Assuming x ( t ) is much smaller than and t is at odd multiples of /2 the system shows approximately linear transfer function near optimal detection poi nt s Fo r example, as t = /2 B ( t ) can be simplified by small angle approximation and expressed as : (1 3 ) As x ( t ) << the baseband output B ( t ) is proportional to the target displacement x ( t ) and it c an be sampled by an analog to digital convertor ( ADC ) for further digital signal processing. Equation (1 3) indicates one of the motivations to design the system at 60 GHz The shorter in the denominator provides a higher system demodulation gain to d istinguish small displacement a t a longer distance away Ideally the linear input output mapping can track arbitrary movement of x ( t ) as long as x ( t ) is much smaller than It is potentially useful to detect small ( m range) vibrati on of objects such as i nsects, acoustic devices, and micro electro

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18 mechanical systems (MEMS) It can serve as a low power, low cost monitor ing network for industrial and security applications. From the viewpoint of total system gain G s which relates (1 1) and (1 2), it i s usu ally determined by antenna gain s, power loss during the reflection and propagation (distance), and transceiver circuits In radar design, moving the operating frequency up to millimeter wave ( mm wave ) range theoretically achieves larger radar received powe r by using less antenna area, which can be seen in the analysis as follows. Generally the antenna gain G increase with frequency for the same antenna effective area A e [ 14 ]: (1 4 ) For a simple flat plate reflector perpendicular to the line of sight (LOS) at far field, the radar cross section is approximately [ 15 ]: (1 5 ) where T is the actual area of the plate. Radar range equation can be used to estimate the received power under far filed condition for simple analysis [ 14 ]: (1 6 ) where P t and P r are the transmitted and received power. G t and G r are the gain of TX (transmitter) and RX (receiver) antennas, and R is the distance betwe en the target and radar. If (1 4 ) and (1 5 ) are plugged into (1 6 ), the received power can be estimated by: (1 7 )

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19 where A t and A r are t he effective area of TX a nd R X antennas. The equation shows apparent advantage of short if other parameters remain the same and air absorption is negligible in a few meters range. Comparing 6 G Hz and 60 GHz radar systems for example, P r at 60 GHz is theo retical ly 10 2 times higher than that at 6 GHz and 1/10 antenna area is used for TX and RX 1.1.2 System Implementation Doppler radar system ca n be implemented by discrete components on printed circuit board ( PCB ) or instrument level for test ing and proof of co ncept s [ 4 ] [ 8 ] [ 16 ] Figure 1 2 shows a 5.8 GHz Doppler radar system integrated on a PCB with two 2 2 patch antenna s array for TX and RX [ 8 ] and it can be used for both n on contact vital sign and vibration detection. Compared to the board level implementation, SoC ( system on chip ) or SiP (system in package) realization are usually desired in terms of cost and system integration. Several s ingl e chip Doppler radar transceivers on CMOS have been successfully developed for non contact vital sign detection [ 3 ] [ 17 ] [ 18 ] To increase the circuit operating frequenc y above tens of GHz III V compound semiconductor (GaAs) is traditionally use d for its superior performance at high frequencies, and SiGe hetrojunction bipolar transistors (HBT) process offers another high perfo rmance silicon based alternative whose f T /f MA X can be nearly 20 0/300 GHz in the 130 nm process Figure 1 2. A 5.8 GHz Doppler radar system integrated on a PCB with two 2 2 patch antenna arrays for TX and RX.

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20 T he state of the art single chip Doppler radar systems are implemented on SiGe based pro cess es to achieve high conversion gain and better noise performance [ 6 ] [ 19 ]. Nevertheless, the economies of scale of CMOS technology with the ability to integrate DSP circuits makes it a very low cost and highly desired platform For a single transistor, f T and f MAX are usually used to evaluate the performance of the device. F T is the unity current gain frequency which is often used to estimate the analog circuit bandwidth at low frequency F MAX i s called the power gain cutoff frequency, representing the transistor limitation at high operating frequency. At the frequencies above f MAX the device is basically passive. A t tributing to the improved transistors performance in scaled CM OS proc ess, f T /f MA X reach es above 120/150 GHz in 90 nm technology and around 200/3 00 GHz in 45 nm technology and it is expected to be further impr oved in more advanced CMOS technologies such as 32 nm and 1 6 nm technologies [ 20 ] Increasing numb er of research work s and significant progress on mm wave CMOS circuit s and system s have been reported for wireless communication radar, and imaging [ 22 ] [ 26 ], and the unlicensed frequency band near 60 GHz is one of the interests Single chip, mm wave Doppler radar transceiver proves to be feasible in CMOS technology [ 21 ] however, vital sign detection by a low cost Doppler radar SiP fully integrated with antennas has not been investigated at this frequency range. Other performance limitations such as TX transmitted power, components loss, and high flicker inherent in CMOS process need to be considered and overcome especially from the radar point of view. In this work, the design considerations in system architecture, circuit components, packaging, and baseband demodulation techniques are studied to realize the radar system at 60 GHz.

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21 1.2 Vibration Detection and Quadrature Architecture As m entioned in the previous Chapter 1.1 of radar signal is a key factor in determining the sensitivity to smal l vibration. It is shown in (1 3 ) that the smaller gives higher level of baseband output B ( t ) and improve s gain of the Doppler radar system. As x ( t ) and other cond itions remain the same, the use of s horter helps distinguish small displacement at a longer distance. A single channel, 94 GHz CMOS Doppler transceiver chip with two horn antennas is potentially t o detect small movement [ 21 ], however the system has limit of null detection point issue which will be explained in the following paragraph s 1.2.1 Optimal and Null Detection Points Figure 1 3 illustrates the concept of small displacement detection using Doppler radar system. When the total residue phase in (1 2 ) is at odd multiples of /2 the vibrating target is at the optim al detection point as x 1 ( t ) As long as x 1 ( t ) is small enough com pared to t he of T(t) the system shows approximately linear transfer function and the baseband outpu t B 1 ( t ) can be easily demodulated by FFT to obtain the spectrum target vibration To detect the displacement (peak to peak) value of the vibration x 1 ( t ) calibration is usually needed to determine G s in ( 1 2 ) at a fixed frequency and distance. Again ideall y the system is able to map any movement of x 1 ( t ) to baseband output B 1 ( t ) with minimal distortion in the approximately linear region. If the displacement of x 1 ( t ) becomes larger to a point that the small angle approximation in (1 3) is no longer valid B essel function is used to analyze the non linear system response assuming x ( t ) is sinusoidal [ 1 ]. The non linear phase modulation effect results in harmonics and intermodulation terms on the output spectrum which will be discus sed in the following chapters

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22 Figure 1 3 The concept of small displacement detection at optimum and null points of Doppler radar system. For illustration purpose x 1 ( t ) and x 2 ( t ) are not to scale. The ratio of harmonics pr oves to be useful to obtain the amplitude of sinusoidal x ( t ) without any distance or gain calibration and the method is not limited to single tone sinusoidal vibration. Recently a two tone mechanical vibration with the amplitude of a few millimeters c an b e accurately measured by 6 GHz Doppler radar system at 1 2 m away [ 8 ] [ 12 ] If higher radar operating frequenc y such as 60 GHz is used, the minimum measurabl e displacement is expected to be further improved as mentioned earlier. Near the null detection point as illustrated by x 2 ( t ) in Figure 1 3 the total residue phase in (1 2 ) is at even multiples of /2 It can be observed that t he baseband output B 2 ( t ) does not contain the fundamental tone of the original vibration The alternating optimal and null detection point s occurs every / 8 as the distance d 0 varies t [ 13 ] and t he issue becomes more p roblematic for the mm wave Doppler radar as is inversely prop ortional to t he radar frequency. Double sideband architecture [ 18 ] or f requency [ 28 ] tuning is possible to alleviate the null detection point issue but it also become s more and more impractical as is getting small

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23 1.2.2 Complex Signal Demodulation To solve the null point issue without extra tuning, c omplex signal demodulation (CSD) [ 5 ] is utilized in the 5.8 GHz CMOS quadrature receiver [ 3 ] In phase ( I ) an d quadrature phase ( Q ) baseband outputs can be generated by the quadrature architecture of the receiver circuits and the CSD baseband output can be software reconstructed in real time : (1 8 ) Since exp ( j t ) has a constant magnitude, the method eliminates the effect of total residue phase t when Fourier transform is applied to S ( t ) to obtain the CSD spectrum. CSD makes the Doppler radar detection independent of the residue phase shift which is mainly de termined by the d istance d 0 bet ween the radar and target The quadrature architecture is essential to solve null detection point issue for the mm wave Doppler radar design. I / Q generation is one of the design challenges for the mm wave system. It can be re alized by a RF quadrature VCO in a direct conversion system [ 21 ], but the loss (power consumption) and mismatch of mm wave quadrature LO distribution needed to be concerned [ 29 ] Quadrature separati on is also possible in the RF signal path using a current domain method or a poly phas e filter as introduced in [ 26 ] [ 30 ], however high flicker noise of the CMOS active mixers used in the system is not preferable as th e vital sign output is nearly dc (1 2 Hz). In this work, I / Q generation is designed at IF stage (6 GHz) by utilizing a compact ring oscillator and it is able to drive two large pass ive mixers for low flicker noise requirement 1.3 Vital S ign Detection Many successful human heartbeat and respiration detection s by Doppler radar system s have been demonstrated [ 1 ] [ 3 ] [ 13 ] [ 18 ] [ 27 ] [ 31 ] at the frequency range from a few to hundreds of GHz As mentioned in the previous sections, the increa se in frequency reduces the component

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24 size and improves the sensitivity to small displacement A 228 GHz heterodyne rada r system reports the detection of vital sign at 50 m away [ 27 ]. However, as the becomes too small compared to target displacement such as chest wall movement, the harmonics of respiration starts to de grade the accuracy of heartbeat detection For the vital sign detection of typical human the optimal radar frequency is found to be a round 27 GHz [ 32 ] In this work, techniques have been deve loped to overcome the challenges and implement the compact SiP for human vital sign detection In addition, the 60 GHz radar frequency is potentially opt imal to detect the vital sign of small animals, for example, which normally have smaller chest wall movement and different ratio of heartbeat and respiration rate Human vital sign sensing is a special case of two tone sinusoidal vibration analyzed in [ 8 ], where the respiration amplitude ( m r 1 6 mm) is normally one order of magnitude larger than that of heartbeat ( m h 0.2 mm). The chest wal l movement due to human respiration and heartbeat can be approximated by a two tone si nusoidal vibration : (1 9 ) where f r and f h are the frequency of the respiration and heartbeat. As presented in Figure 3 of [ 1 ], it shows the detection results of the Ka band Dopp ler radar system using the frequency of 27 GHz at 2 m in front of the targe t The upper right corner is the baseband output signal in time domain and the spectrum clearly shows the vital sign signals. It should be noticed that the magnitude of peaks are n ot always proportional to the heartbeat and respiration amplitude. They are determined by the coefficients of Bessel functions and will be discussed in the later c hapters. Based on the illustration in Figure 1 3 the displacement of respiration is likely t o be out of the linear region depending on of T(t) and results in harmonics on the baseband output spectrum. When the operating frequency of the radar is increased, the heartbeat peak is possibly blocked by

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25 the 3 rd or higher order harmonics of respiratio n, for example, and result in the difficulty of heartbeat detection. Figure 1 4. Comparison of vital sign detection using different radar frequencies. The simulation results of (A) 6 GHz and (B) 60 GHz Dopple r radar system are plotted The radar frequency at 60 GHz is far beyond the optimal carrier frequency [ 32 ] for vital sign detection, and thus further detection techniques need to be developed. The at 60 GHz (5 mm) is comparable to m r and th e n on linear phase modulation becomes much more serious. Figure 1 4 shows a simulated comparison between 6 GHz and 60 GHz detection results after CSD In Figure 1 4 (B ), the relatively small heartbeat peak is overwhelmed by the harmonics of respiration even w ith out the presence of system and environmental noise. In some cases, the fundamental respiration peak (R 1 ) is too small to be distinguished and results in detection failure, which will be discussed in Chapter 5 For this reason, the instrument based 60 GH z millimeter

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26 wave life detection system (MLDS) [ 31 ] measures the heartbeat signal while t he target person is holding the breath, and even the accuracy of respiration detection itself is limited because of the harmonics intermo dulation, and high sensitivity to environmental no ise In this work, further analysis based on Bessel functions and signal recovery techniques are investigated to detect the vibration comparable to 1.4 Millimeter wave Packaging and Integration Increasing t he radar operating frequency poses challenge s in the transition design between chip and antenna The length of typical bonding wire could be several hundreds of micrometers depending on the chip thickn ess as illustrated in Figure 1 5 Figure 1 5 Typical wire bonding packaging configuration. The parasitic inductanc e of the wire is usually in nanohenry range at around 10 GHz. Consequently the parasitic inductance of the traditional packaging configuration is unacceptably high ( nH range) for this frequency range and thus t he research on low loss cost effective transition h as drawn increasing attention in the mm wave ICs O n chi p antennas presented in [ 19 ] [ 33 ] [ 34 ] are alternative s to avoid the lossy mm wave transition as t he antenna size is usually small enough in the mm wave applications. However, the on chip antenna generally suffers from low radiation efficiency d ue to the silicon substrat e [ 22 ] [ 33 ], and thus the limited patch antenna gai n from 10 dBi to 2 dBi is usually reported at 60 GHz. Another drawback of

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27 the on chip antenna is typically the narrow bandwidth which is limited by the r elatively short distance between the top and bottom metal layers With a l ow loss antenna transition a PCB antenna with a superior performance compared to an on chip counterpart can be used, achieving system in package integration. Several custom mm wave packaging techniques [ 22 ] [ 35 ] successful ly achieve the low loss transition by designing supporting structure to horizontally align the chip and antenna, which greatly reduces the parasitic inductance. As shown in Figure 9 (b) of [ 22 ], one technique is to use BGA (ball grid array) tin ball underneath to raise the chip to the same plane of the antenna. I n this case three very short bond wires are used in parallel to minimize the parasitic resistance and inductance effects The parasitic inductance of the antenna transition is estimated to be less than 100 pH at 60 GHz [ 22 ] In addition, bumping and flip chip process widely used in IC industry is known to be another promising solution for mm wave SiP applications without the need of extra supporting structure A nalysis and tests have been made to characterize the mm wave flip chip transition s and techniques such as high impedance compe nsation are propos ed to optimize the transitions based on CPW fed structure [ 36 ] [ 39 ]. This work proposes a compact flip chip transition for mm wave radars, in which TX/RX isolation and optimal LO distribution path are considered. S ingle die bumping and flip chip process provided by C ollier Ventures Inc is evaluated and adopted to integrate th e 60 GHz CMOS radar chip with two closed placed P CB patch antennas The issues including TX/RX isolation, i mpedance m atching, and loss of the structure will be investigated and presented in Chapter 3.6 The Doppler radar SiP in this work demonstrates the first vital sign detection by the CMOS flip chip integrated radar at 60 GHz. The shorter offers significan t area reduction and

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28 fl exibility in system integration, and the CMOS flip chip implementation provides the low cost potential for mass production. It can be readily embedded into one of the smartphone functions, for example, making it a pervasive first aid tool for no n contact vital sign monitoring. The system can also be applied to a large sensor network for many other applications

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29 CHAPTER 2 SYSTEM DESIGN AND IN TEGRATION 2.1 Overview Figure 2 1. Block diagram of the 60 GHz CMOS micro radar system including transceiver chip in 90nm CMOS, TX and RX patch antennas, and flip chip integration. Fig ure 2 1 shows an overview of the system including the CMOS radar chip (0.96 mm by 2.35 mm) and two PCB patch antennas on RT/duroid 5870 lam inate (31.3 mm by 45 m m) for TX and RX. RF pads on chip and the metal traces on the laminate (zoom in area) are designed for mm wave flip chip integration. The integrated transmitter containing two VCOs, an up conversion mixer, a balun, and a driver is des igned to transmit an un modulated, 60 GHz continuous wave (CW) signal through the TX antenna. The signal is reflected and phase modulated by a small vibration of the target due to Doppler effect, and then received by the RX antenna. For the integrated rece iver, the weak recei ved signal is amplified by a 60 GHz LNA and down converted by the same VCOs. Since the same VCOs are used for TX and RX and the phase noise are correlated for short distance detection, this range correlation effect [ 13 ] [ 40 ] results in significant reduction of VCO phase noise at radar baseband output and thus free running VCOs can be adopted in the system.

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30 2.2 Sensitivity and Radar Received Power Sensitivity estimation provides an id ea of margin that certain RX received power is targeted to achieve requir ed output signal to noise ratio ( SNR re ). For vital sign applications, the frequency of interest is near dc ( 1 Hz) which results in several situations quite different from typical com munication systems. As shown in Figure 2 2, the noise figure ( NF ) F 4 and F 5 are estimated to be as high as 60 dB [ 3 ] due to flicker noise, and total NF is around 31.2 dB even with a high gain (35 dB) first stage. The FFT observ ation time window ( TW ) at baseband is usually about 20 s to obtain enough cycles and maintain good spectrum resolution bandwidth ( RBW = 1/ TW = 0.05 Hz). The use of small RBW significantly reduces the overall noise level at output. However, a low sampling f requency ( f s ) of ADC around 50 Hz is normally chosen to have a reasonable FFT bin size (TW f s ) for real time computation on portable devices. After sampling, the actual noise level near 1 Hz increases due to aliasing since f s is far below baseband output bandwidth B ( 1 MHz ) and flicker noise corner. Experiment results in [ 3 ] shows the noise level is often dominated by the folded white noise when f s is low, and the radar RX sensitivity ( S ) is estimated as (in dB scale): (2 1) Figure 2 2. Sensitivity estimation of the Doppler radar receiver. F 4 and F 5 are estimated to have high noise figure ( NF ) at 1 Hz due to flicker noise.

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31 where kT is thermal noise floor per H z at input, and SNR re is the requirement at output. Here kT RBW (dBm) + NF wh (dB) represents the white noise level before aliasing and NF wh is around 7.1 dB dominated by the first two stages in Fig ure 2 2 The corresponding sensitivity is 117 dBm at SNR re = 20 dB. It should be noticed that if f s is increased or RX has a higher flicker noise corner, the folded flicker noise might become dominant and degrade the sensitivity. The estimation of radar received power is nontrivial since it involves radar cross section of human body, and sometimes the short range results in near fiel d detection where (1 7 ) is invalid. However, the maximum possible received power can be estimated by assuming an infinite perfect reflector and using [ 14 ] : (2 2) where G t and G r are both 5 dB, P t is set at 0 dBm, and the target is at 2 m away. The maximum received power P r is then calculated to be 70 dBm at 60 GHz while the travel distance R is 4 m based on the image theory. In the real case, the received power is lower since the actual target has smaller reflection area and other sources of loss are present. The estimation above reveals possible margin between the received power and sensitivity, a nd it indicates the need of a high gain LNA for noise suppression and minimized flicker noise of baseband circuit blocks. For the detection within a few meters, moderate tran smitted power around 0 dBm in (2 2 ) is targeted to reduce TX power consumption, as long as the received power meets sensitivity requirement. Gain stages may be placed at RX to satisfy ADC input specifications. In fact, Doppler radar transmitting un modulated radar signal allows TX operating in nonlinear region to have high efficiency. F or comparison, generally one stage of a LNA in 90 nm CMOS achieves a gain around 7 dB with a power consumpt ion less than 15mW at 60 GHz [ 41 ] [ 43 ], on the other hand, state of art PAs achieving 20 dB gain and 10 dBm saturated power (P sat ) usually require

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32 more than 150 mW [ 41 ] [ 45 ]. Thus, a TX driver amplifier operating near P sat is adopted in the system for the short range detection. 2.3 Design Con sideration for IF Stage Quadrature channels at RX are required for the use of CSD to eliminate null detection points. I / Q separation is realized at the IF stage of heterodyne receiver instead of a direct conversion topo logy, which avoids the loss, mismatch and power consumption of 60 GHz I / Q separation and distribution Because the phase noise is correlated at short detection range and significantly reduce d by range correlation effect [ 13 ] [ 40 ] a f ree running quadrature ring VCO can be used at IF to provide a compact and power efficient choice for driving two large passive mixers. The compact ring oscillator without the use of inductors is able to drive the large passive mixers and achieves a wide t uning range (2 8 GHz). Passive mixers are used for low flicker noise because this noise from mixer is not cancelled through range correlation. Flicker noise is one of the major considerations for the Doppler radar application since the baseband outputs is very close to dc (0 2 Hz for heartbeat and respiration). The flicker noise can be represented by [ 46 ]: (2 3) w here K fk is a process dependent constan t and C ox is the total gate cap acitance of the transistor. Based on the equation larger transistors tend to give lower flicker noise at a fix frequency. Thus t wo passive mixers with large transistor sizes are used in I and Q path to minimize the flicker noi se presented in the baseband [ 3 ]. The wide tunable IF provided by the ring VCO compensates the possible RF drift due to mm wave circuit modeling uncertainty and improves the system robustness. In addition, the IF frequency chosen roughly a decade away from the RF LO (54 GHz) makes the LO feed through

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33 be effectively attenuated by the output tank of the RF mixer. At the output of IF stage, two sets of baseband am plifiers are also integrated on the CMOS chip, enabling the base band outputs B I (t) and B Q (t) to be directly sampled for software CSD [ 5 ]. 2.4 Floor Plan and Flip Chip Transition For the system floor plan shown in Figure 2 2 the mixers are easily reached by the RF VCO and single ended antenn as are adopted and placed closely on the same side of the chip to minimize loss and power consumption of 54 GHz LO dist ribution It also prevents the dc /baseband connections from interfering w ith antennas. The proposed G S G S G transition achieves impedan ce match and provides enough TX/RX isolation to reduce the direct coupling of signal from TX. By using the 150 m pitch and 50 impedance interface, on wafer (for chip) and on board (for ante nnas) probing measurement can be conducted separately. The anten nas and flip chip transition were designed at the measured optimal operating frequency of the chip after fabri cation, which helps to ensure the system performance against the modeling uncertainty, under estimation of parasitics, and manufacturing variation

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34 CHAPTER 3 CIRCUIT COMPONENT DE SIGN 3.1 I nductor Lumped element modeled inductors are extensively used throughout the 60 GHz RF portion of the system for their low area consumption compared to transmission lines [ 41 ]. To achieve h igh effective quality factor Q [ 47 ] and compact layout, different designs of inductors are evaluated by 3 D EM simulation which captures important mechanisms such as skin effect, proxim ity effect [ 48 ], substrate loss and return current path [ 49 ] In the 90 nm CMOS process, the inductor is implemented on the top metal 9 (M9) layer, surrounded by thick ground (GND) walls stacking from substrate all the way to M9 to minimiz e the parasitic resistance and induc t ance on the chip ground. Figure 3 1 shows a 95 pH inductor at 60 GHz along with the interaction between the ground structure and the inductor itself. As excited from port 1, the return current I r seeks the lowest impeda nce path back to the source [ 49 ] [ 50 ] and contributes negative mutual inductance. The displacement current I d due to capacitive (E filed) coupling flows through the substrate and back to the source ground. Figure 3 1. Simulation of a 1.5 turn, 95 pH inductor along with the surrounding ground plane. The return current ( I r ) and displacement current ( I d ) illustrate the EM interactions bet ween the ground plane and the inductor.

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35 T able 3 1. Simulated inductor performance at 60 GHz Number of Turns Inductance (nH) C p (fF) R s ( ) Q 1 124.2 17.2 1.7 18.6 2 122.1 10.4 2.4 15.7 3 nm CMOS process wi th 9 metal layers. If the ground plane is moved closer to the inductor, the inductance density drops and parasitic capacitance to the ground (C p ) increas es which degrade the Q of the inductor Thus there is no metal ground underneath the inductor in the s tructure. Increasing the number of turns achieves the same inductance with smaller footprint and thus C p as listed in Table 3 1. However, the series resistance (R s ) of 2 turn inductor is higher than that of the 1 turn inductor, which is largely due to the current crowding effect at the high frequency which can be observed in the current distribution plot similar in [ 48 ]. The Q of the single turn inductor is 20% higher than that of the two turn inductors, despite the increase in area and C p helps lower the impact of C p Considering the tradeoff between the area and Q of the inductor, the L shaped, 1.5 turn structure as in Figure 3 1 is adopted in the mm wave transceiver design utilizing the double 47 ] in the circuit simulation Figure 3 2. Microphotograph of the on chip inductors used in the 60 GHz front e nd. (A) 95 pH, L shaped inductor (s imulated Q 18 ) (B) 128 pH circular inductor (s imulated Q 19.5 ).

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36 The 95 pH inductor achieves a simulated Q around 18 at 60 GHz, and the die photo of the L shaped inductor is shown in Figure 3 2 (A). Figure 3 2 (B) presents another circular inductor configuration which is useful in the layout situation that port 1 and port 2 are very close to each other. Similarly the metal ground underneath the inductor is removed, and the single turn structure minimizes the curren t crowding effect mentioned previously to reduce the resistive loss. The circu lar inductor provides the simulated inductance of 128 pH and Q of 19.5 at 60 GHz. This configuration can also be used as a high Q differential inductor for VCO resonant tank if t he center tap is applied [ 52 ]. Figure 3 3 Simulation of the isolation between two closely placed inductors as they are in the actual on chip situation.

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37 Minimizing t he coupling between adjacent inductors facilitates a compact chip layout and increases the flexibility of the system floor pla n. As previously mentioned the stacked M1 through M9 ground walls surrounding the inductor reduc es the parasitics on the ground, a nd they also increases the level of isolation between closely placed inductors [ 43 ]. Figure 3 3 shows the 3 D EM simulation of two 95 pH inductors placed closely together (distance 64 m) to emulate the real on chip situation It can be seen from the plot that the L shaped inductor configuration provides the advantage in isolation since the adjacent metal traces of two inductors are orthogonal, which helps to reduce the magnetic coupling [ 44 ] The simulated coupling level between the two inductors is low as S 12 and S 14 are both around 50 dB over the 40 GHz to 80 GHz frequency range. 3.2 Radar Receiver Front end Design In the mm wave r adar application, analysis in Chapter 2.2 indicates the need of a hi gh gain LNA in RX front end. I llustrated in Figure 3 4 as Vin is from the single ended RX patch antenna, the R X front end is composed of a 60 GHz five stage cascode LNA, a single balanced mixer, and a 54 GHz VCO. Transistors are biased at the current dens ity aroun the optimal f MAX and noise performance [ 41 ], and extra parasitic capacitance across the gate, drain, and source due to interconnects was estimated by a 3 port extraction in the 3 D EM simulator and includ ed in t he circuits simulation [ 51 ]. 3.2.1 LNA The 5 stage ca scod e topology with series inductors (L p1 L p5 ) is chosen for its superior perform ance in gain and isolation L g1 and L s1 are 190 pH and 52 pH respectively as matched to the 50 flip chip transition. Stage 2 4 are identical while inter stage conjugate matching was achieved by designing the values of C g2 to C g5

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38 Figure 3 4 RX front end (60 GHz to 6 GHz) including the 5 stage LNA, single ended mix er, and 54 GHz VCO (Bias and LO distribution details not shown). Figure 3 5 Microphotograph showing the cascode portion of the layout and vertical access to the power grid. In this manner the 95 pH inductor in Figure 3 1 can be extensively reused in L p1 L p4 and L d1 L d4 which greatly reduce the custom layout cycle. L p5 and L d5 are both set at 45 pH for the output matching between the LNA and following single balanced active mi xer. As presented i n Figure 3 5 the 1.5 turn, L shape inductor can be arranged to achieve a highly compact layout at the low

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39 coupling level between adjacent inductors as previously mentioned. The 5 stage cascode layout consumes an area of 0.16 mm 2 Simulation shows the LNA provides a 38 dB gain and 5. 2 dB NF at 60 GHz while consuming 38 mA at a 1.2 V power supply. 3.2.2 Active Mixer and RF VCO The single balanced mixer with inductive loads (L d6 = L d7 = 3.4 nH) is adequate for the application based on the simulation. It shows a gain of 0 dB and also serves a s a single ended to differential conversion for the following IF mixers. The 54 GHz LO feed through is far away from the down converted signal on the spectrum and also greatly attenuated by the load resonant tank designed at the IF. A source follower buffe r is n ecessary at the mixer output [ 3 ] to provide low output impedance and drive the large passive mixers. Figure 3 6 shows the LC cross coupled, 54 GHz VCO tuned by accumulation mode varactors [ 52 ] with ps eudo differential LO buffers [ 53 ] to drive up and down convert mixers. The simulated tuning ran ge covers from 51.6 to 54.9 GHz, and the phase noise is 101 dBc/Hz ( at 1 MHz offset) at 54 GHz. The VCO core consumes aro und 18 mW. The first stage of the differential LO distribution buffers utilizes cascode topology to provide better isolation to the core, which prevent s fre quency shift due to possible loading effects Common source (CS) is used in the rest of the stages f or the larger voltage headroom and avoiding t he pole of cascode transistors (between N 2 + and N 3 +). Figure 3 7 shows the microphotograph of LO distribution network design. L 1 to L 12 represent the differential inductors and each of them contains the positive (+) and negative ( ) parts. The d ifferential inductors require the co design of conjugate matching and adequate physical length to reach the next stage. In some cases the inductor length as long as 100 m to 200 m is required.

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40 Figure 3 6 54 GHz RF LO generation and distribution ( A) 54 GHz VCO core ( B) D ifferential LO distribution buffer network to up and down convert mixers. Figure 3 7 Mic rophotograph showing the 54 GHz LO distribution ne twork from VCO to up and down convert mixers. To accurately model ever y trace as a lumped inductor, the physical size has to be smaller than 1/10 of the on chip effective wavelength ( eff ) which can be expressed as

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41 (3 1) where is the wavelength at 60 GHz in free space and r is the relative per mittivity. Since the inductors in the LO distribution network usually have longer physical length and do not require compactness and high inductance density metal ground shielding ( on M1 and M2) can be placed right underneath the inductors to block the high permittivity silicon substrate ( r 12) This is different from the inductor design concept mentioned in Chapter 3.1. By thi s arrangement, t he inductors only see the silicon oxide ( r 4 ) assuming the effect of air above the silicon is neglected for simple estimation and thus r is locally increased which assures eff is around 2500 for the accurate lumped element modelin g It can be observed from Figure 3 7 that the L shaped inductor configuration facilitates the compact LO distribution design connecting the buffer stages, and the GND plane filling increases the isolation between the closely placed components as mentione d in Chapter 3.1. At the power consumption of 11 0 mW, the receiver front end is designed to provide a single ended conversion gain of 38 dB when one of the IF outputs terminated by 50 input 1 dB compression point ( P 1dB ) is at 44 dBm which is much higher than the radar received power estimated in ( 2 2 ). 3.3 Radar Transmitter Front end Design As explained in Chapter 2, t he TX shares the same IF and RF VCOs with RX to utilize ran ge correlation effect [ 13 ] [ 40 ] in the radar system. Figure 3 8 shows the TX front end design A double balanced mixer is used to accommodate the d ifferential signals from the RF and IF VCOs and con vert the 60 GHz outputs to single ended by the passive balun load A three stage driver

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42 ( tuned amplifier ) at 60 GHz following the bal un boosts the power to around 2 dBm at V out and drive s the single ended TX patch antenna. 3.3.1 Passive Balun The passive balun c an be realized by coupled tra nsmission line structure [ 54 ] or transformer based coupled spiral inductors [ 41 ]. The lump element modeled t ransformer balun shown in Figure 3 9 is designed to provide t he balanced output loads for the up convert mixer while minimizing the area overhead. Figure 3 8 TX front end (6 GHz to 60 GHz) using the double ba lanced up convert mixer, balanced loads ( balun ) and three stage driver at 60 GHz. Figure 3 9 Lumped element modeled transformer balun with differential to single ended impedance conversion.

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43 The primary inductor ( L p ) is implemented on the high aluminum pad layer to reduce the loss to substrate, while M9 is used for the secondary inductor ( L s ) to form a stacked coupling structure. Based on the CMOS metal design rules, t he ve rtically stacked transformer shows less magnetic flux leakage and higher coupling coefficient ( k ) compared to a planar transfo rmer The metal ground plane underneat h the transformer is removed similar to that of inductor design in Chapter 3.1 The transformer serves as part of the matching network between the mixer and the TX driver, and the differential impedance Z s and single ende d impedance Z p follows [ 55 ] : (3 2) Here n is conventionally defined as the turn ratio. At a given k which normally ranges from 0.3 to 0.9, varying the value of L p and L s achieves different conversion of the impedances. The L p and L s designed here are 108 pH and 88.5 pH respectively considering the tr adeoff among area, matching, and parasitic loss. The fitting k of the balun is around 0.65 and area i s 0. The area overhead is much smaller than that of the transmission line based balun which usually has a size comparable to wavelength In 3 D EM simulation, the differential to single ended insertion loss of the balun is around 5 dB at 60 GHz which is dominated by the limited mutual inductance. Multi turn inductors may be used to improve k at the cost of higher parasitic loss such as series resistance and parallel capacitance to the ground 3.3.2 TX Driver The three stage TX d river at 60 GHz use s two cascode stages and a CS tuned amplifier as the last output stage. In the simulation, t he LO from IF quadrature VCO is 10.7 dBm (differential ) at 6 GHz, and it is up converted to 60 GHz with the power level boosted to 2 dBm (single ended) by the mixer and driver The output power of the TX driver is designed to operate near its P sat as mentioned Chapter 2.2. The conversion gain from IF to RF is 12.7 dB, and the

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44 driver output is matched to 50 for the f lip chip transition and antenna or on wafer probin g measurement. Overall, the TX front end consumes 35 mW power and 0.13 mm 2 area, which are both less than 10% of the overall system consumptio n. 3.4 IF Quadrature VCO and Passive Mixer The I / Q separation is performed at IF stage as the compact quadrature ring VCO is used to drive two passive mixers for low flicker noi se at baseband outputs in Figure 3 10 d on the Barkhausen criteria [ 56 ] [ 57 ]. The differential I and Q LO signals with two stage resistive load buffers are able to drive large passive mixers, while I LO signals are also needed by the up convert mixer in the TX front end. The buffers (A I and A Q ) have separate power supply (V I and V Q ) to compensate possible I / Q amplitude mismatch due to the different load s. The two passive minimize the flicker noise at the baseband outputs [ 3 ]. Figure 3 10. IF stage (6 GH z to dc) including the quadrature ring VCO, IF LO buffers, and passive mixers.

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45 Figure 3 11 shows the design of the delay cell (D 1 and D 2 ) which includes two tuning mechanisms to improve the precision and tuning range. By only looking into the left hand sid e of the delay cell, the oscillation fre quency can be represented as [ 57 ]: (3 3) where g m is the transconductance of the transistor. G L stands for the total resistive lo ad of M n1 M p1 and M p3 and C L is the total capacitance seen at the output node (V out ). Figure 3 11 Design of the delay cells (D 1 and D 2 in Figure 3 10 ) with two tuning mechanism s (V p and V b ). B y controlling V P turning on M p3 achieves the highest oscillating frequency f os ( G L + g mp3 = g mp1 ) at a fix C L while turning off M p3 results in the lowest f os The accumulation mode varactors at output nodes [ 52 ] [ 58 ] provide extra flexibility to tune C L at the cost of increase d total capacitive load and reduced maximum f os The width (W V ) and number of finger (F V ) are chosen to be the minimum allowed by the process (W V V = 4) to reduce t he impact on maximum f os while the channel length (L V ) is set to be maximum (L V tuning range. In the simulation the tuning range is 63 % (5.2 9.96 GHz) without the varactors,

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46 and is increased to 89% (3.28 8.5 GHz) by adding the vara c tors. Over the entire tuning range, the common mode output level of the ring VCO stays roughly the same at 0.6 V which eases the design of the next stage. The simulated phase noise is 83 dBc/Hz ( at 1 MHz offset) when the f os is at 6.4 GHz, and to tal power consumption of the IF stage is 99 mW. Figure 3 12 shows an example of the quadrature LO outputs (LO I +, LO I LO Q +, and LO Q in Figure 3 10). In the simulation, the oscillating frequency is tuned at 6.375 GHz, and the mode level is maintained around 0.6 V which is about half of the power supply voltage (1.2 V). Figure 3 12. Simulated four out put phases (LO I +, LO I LO Q +, and LO Q in Figure 3 10) generated by the quadrature ring VCO. 3.5 CMOS Radar Chip Overview Figure 3 13 shows the microphotograph of the micro radar in 90 nm CMOS technology. On the left are the 60 GHz RF outputs and on the right are the baseband I / Q signal outputs which can be directly sampled by the oscilloscope or ADC All the dc b iases are accessible from the dc

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47 pins on the top and bottom to determine the optimal bias points. The area of the 60 GHz RF core is 0.73 mm 2 The ar ea reduction of the prototype chip is limited by the nu mber of dc pads. Figure 3 13 Microphotograph of the 60 GHz CMOS micro radar 3.6 Flip chip Integration and PCB Patch Antenna The flip c hip transition and antennas are design ed at the optimal operating frequency (55 GHz) of the transceiver circuits measured in Chapter 4 The flip chip process is able to provide a low loss, impedance matched transition for the mm wave system in package applications. By proper design of the sold er bump, on chip RF pads, and the G S G (Ground Signal Ground) trances on th e PCB the parasitic capacitance and inductance can be estimated by 3 D EM simulator and compensated [ 36 ] [ 39 ] Different from the conventional wire bonding package which is usually 1 2 mm long and introduces unaccepta bly large series inductance (a few nH ) at mm wave frequencies flip chip process places small solder bump between the chip and PCB substrate and the bonding is established by the re flow of solder bump s under heat and pressure. The series inductan ce of the transition is usually smaller than 150 pH at 60 GHz [ 38 ]

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48 3.6.1 Transition Design and Impedance Match Figure 3 14 show s the flip chip transition design between the 60 GHz CMOS radar transceiver chip and PCB pat ch antennas on the RT /duriod 5870 laminate Two bumps are used on each RF signal pad (S) to reduce the series inductance and resistance, and similarly three pairs of bumps are plac ed between the RF ground pads (G) and ground traces on upper metal of PCB, which is also connected to the bottom ground through PCB vias. Since the width of the 50 feed chip G S the taper structure is designed in between for the conversion. The increased inductance due to the taper structure is capacitively compensated by controlling the length d 1 and d 2 to maintain 50 impedance match The flip chip transition can be modele d as a two port network [ 38 ] as shown in Figure 3 14 while port 1 is on the PCB and port 2 is on the chip. Figure 3 14 Flip chip transition design between the 60 GHz CMOS radar chip and PCB patch ant ennas on RT/duroid 5870 laminate. The inductance and capacitan ce values in the two port model are extracted from the 3 D EM simulation at 55 GHz and shown in Figure 3 15 Based on the simulation, t he transition is not solely dominated by the series induc tance since the nearby ground structure contributes

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49 significant amount of shunt capacitanc e, and the resistance is small enough to be neglected. The value of the parasitic inductance and capacitance can be controlled in certain degree by designing the numb er/ dimension of the bump, on chip pad size, and the metal trace s on PCB. Figure 3 15. Impedance analysis of the transition at 55 GHz before and after the flip chip process. Figure 3 15 demonstrat es the impedance analysis bef ore and after the flip chip transition. The TX and RX patch antennas are matched to 50 respectively using the cut on the edge and /4 transfo rmer. Looking into p oint A is the impedance before the flip chip process, and point D is the impedance seen by the chip output after flip chip transition. As shown on the S mith chart from point A B, C, and to D by properly designing the flip chip transition, the parasitic capacitance and inductance can be used to largely cancel the effect of each other The impedance remains very close to the center of the smith chart and achieve adequate match The insertion loss of the flip chip transition is around 1.5 dB at 55 GHz.

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50 off between the antenna bandwidth and surface wave loss [ 22 ] [ 59 ]. In the Doppler radar application, the antenna is not expected to a chieve high bandwidth since single frequency is transmitted and the frequency selection of antennas also removes the undesired harmonics and images. The use of thin PCB substrate also achieves the adequate width of 50 feed line for the flip chip transiti on design and reduces the spurious radiation. A thick FR4 PCB is needed underneath the RF laminate to support the soft RT/duriod 5870 laminate Figure 3 16. Microphotograp h of the flip chip area on RT/duriod 5870 surface. The solder mask in dark green is deigned to control the reflow of solder bumps during the flip chip process. The hole goes through both the RT/duriod 5870 and FR4 supporting board. Figure 3 16 shows the microphotograph of flip chip area on RT/duriod 5870 lamin ate surface In the flip chip process, two solder bumps will be placed on each RF copper trace (G S G S G), and one solder bump will be placed on each dc bias trace. All the copper traces are

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51 partially covered by the green solder mask which is designed to control the reflow of the solder bump s as the heat and pressure are applied. After the chip is attached to the board, the through hole underneath the chip allows the RF circuits contacting the air which helps to maintain the EM characteristics previously simulated. 3.6.2 Patch Antenna Figure 3 17 shows the simulated s parameter of the 55 GHz patch antenna after integrated with the flip chip transition (point D in Figure 3 15 ). Looking into port 1 is the TX antenna and port 2 is the RX antenna. By using the orth ogonal feed lines and G S G S G arrangement, the isolation between port 1 and port 2 reaches 34 dB at 55 GHz even the TX and RX a ntennas are placed closely to each other in the compact floor plan. Minimizing the un modulated signal directly coup led from T X to RX reduces the dc offset in the system. The simulated gain of the single patch antenna is around 5 dBi as shown in Figure 3 18 Figure 3 17 Simulated patch antenna s parameter after the flip chip packaging. S 11 shows t he input matching of the antenna and S 12 represents the isolation between two ports.

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52 Figure 3 18 Simulated patch antenna pattern after the flip chip packaging.

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53 CHAPTER 4 EXPERI MENT AL RESUTLS 4.1 Millimeter wave CMOS Transceiver Measurement A separate test structure which dupli cates the 60 GHz RF core in Figure 2 1 was measured on wafer At a total 190 mW power consumption (1.2 V power supply), the receiver provides more than 30 dB down conversion gain from 52 GHz to 56.5 GHz as plotted in Figure 4 1 The peak down conversion gain is 36 dB from RF to IF (single ended) with the RF VCO operating at 48.4 GHz. The tuning range of the RF VCO is from 48 to 51 GHz. In the first pass, the reduced peak gain and shift ed peak frequency from 60 GHz to 55 GHz are possibly due to several reasons. For example, the modeling of active and passive devices might under estimates some of the parasitic capacitance and inductance at this frequency. Also the current return path assumed in the EM simulatio n setup (Figure 3 1) is different from the complex situation on the actual chip, resulting in the discrepancy between simulated and actual inductance values The input P 1dB of the receiver is measured at 42 dBm. Figure 4 1. M easured down conversion gain (60 GHz to 6 GHz ) versus RF input frequency. The RF input power wa s set at 60 dBm to emulate the weak reflected radar signal.

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54 Figure 4 2 shows the measurement results of the TX front end As the IF input is fixed at 6.6 GHz e mulating the LO signal from the quadrature VCO and the RF VCO is at 48.4 GHz, t he output P 1dB of the transmitter is about 0.3 dBm and P sat is at 1.5 d Bm. In this test structure an extra on chip balun was used to convert the single ended IF input signal t o differential signal for the up convert mixer, and the simulated balun loss was de embedded in the experiment The 6 GH z on chip balun is not used in the final system since the IF VCO already provided the differential signals If the IF VCO power is esti mated to be 10 dBm (differential) in Figure 4 2 the radar TX output power reaches around 1 dBm which is very close to its P sat as previously anticipated in Chapter 3. Figure 4 2. Measured up conversion (6 GHz to 60 GHz ) ga in compression and P in (differential) versus P out (single ended) curve The IF input frequency was at 6.6 GHz emulating the LO signal from the IF quadrature VCO and the RF VCO is at 48.4 GHz 4.2 IF Quadrature Ring VCO A separate test structure of the IF quad rature ring VCO is fabricated and tested to verify the tuning range and output power. Fig ure 4 3 presents the measured spectrum of the ring VCO When both V p and V c in Figure 3 11 are at the highest voltage level ( 1.2 V) M p3 is off and the

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55 two varactor s p rovide the largest C L The lowest f os = 1.8 GHz is measured as shown in Figure 4 3 (A) On the contrary, as both V p and V c are at the lowest voltage level (0 V), M p3 is on and the C L from the varactors is minimized. The highest f os = 7.84 GHz is measured a s presented in Figure 4 3 (C). Figure 4 3 (B) shows an intermediate f os = 6 GHz which can be achieved by tuning V p and V c Amplified by the two stage buffer showing in Figure 3 11, the VCO output power level is around 3 dBm to 1 dBm (single ended) in tho se three cases after a 2 dB cable loss is de embedded. In the experiments, t he total power consumption including the two stage buffers is around 90 mW. Figure 4 3. Measu red single ended output spectru m of the quadrature ring VCO (A) At lowest f os = 1.8 GHz (B) At intermediate f os = 6 GHz (C) At high est f os = 7 4 8 GHz.

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56 4.3 Patch Antenna Test After the optimal operating frequency of the RF transceiver is determined to be around 55 GHz from the measurement, the patch antennas on PCB can be designed accordingly. The design of G S G metal traces with 150 m pitch shown in Figure 2 1 facilitates both the probing measurement of the antennas and flip chip bumping. Figure 4 4 (A) shows an example of the antenna test structure and 67 GHz G S G probe setup. Figure 4 4 (B) plots the measured S 11 of the patch antenna before flip chip pr ocess s imilar to point A in Figure 3 15 The good agreement between 3 D EM simulation and measurement indicates the feasibility to design the antennas right a t the measured optimal frequency of the transceiver chip, even the antenna bandwidth is l imited by the thin RT/duroid 5870 laminate which will be further discussed in Chapter 5.3 Figure 4 4 Antenna return loss (S 11 ) measu rement. (A) Test structure and G S G probe setup (B) M easured and 3 D EM simulated S 11 of the PCB single patch antenna.

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57 A separate patch antenna test structure as shown in Figure 4 5 was used to measure the radiation pattern. T wo identical linearly polariz ed antennas surrounded by ground plane were fabricated on the RT/duroid 5870 laminate for TX and RX, and the zoom in area shows the G S G S G structure which can be used for both probing measurement and flip chip integration. Figure 4 6 shows the experimen tal setup for the measurement of the radiat ion patterns [ 61 ]. The laminate was placed on a probe station and a G S G probe was used to excite one of the antennas. The metallic chuck right underneath the antennas is covered by t he absorber to reduce its influence on the radiation pattern A sta ndard gain horn antenna (SGH) was set on a sliding track which covers an azimuthal range of 25 at the broadside. The probe and the SGH were connected to the network analyzer to measure |S 21 |. A calibration is performed in advance to locate the reference planes at the ends of two cables. Then Friis transmission equation was then employed to calculate the realized gain of the patch antenna, which is written as (4 1) Figure 4 5 Patch antenna test structure with probing/flip chip G S G S G area zoomed in.

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58 where 0 is a free space wavelength, L is the insertion loss of the probe including the interfaces between cable and probe coaxial end an d between antenna and probe tip G t is the gain of the SGH, which has been characterized previously and R is the distance between the SGH and the patch antenna. Figure 4 6 Probe based measurement setup for the broadside radiation patterns. The metallic chuck right underneath the antennas is covered by the absorber to reduce its influence on the radiation pattern. Figure 4 7. Radiation pattern s of the single pa tch antenna. (A) XZ plane (B) YZ plane. The cross polarization shows the 90 polarization mismatch pattern. The measured and simulated gain patterns in XZ and YZ p lane at 55 GHz are shown in Figure 4 7 T he peak gain of the main beam i s about 4.86 dBi. The co polarization pattern shows the results when the orientation of linearly polarized SGH matched to that of the single patch

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59 antenna, and the cross polarization pattern was obtained by rotating the SGH by 90 and result ed in very litt le antenna reception which is ideally zero due to the 90 polarization mismatch The realized gain at zenith agains t frequency is shown in Figure 4 8 The measured results show a good agreement with the simulations. Figure 4 8 Measured and simulated realized gain spectrums at zenith. 4.4 Radar Transmitted Power Test Figure 4 9 shows the photo of the final 60 GHz micro radar SiP assembly incl uding the CMOS transceiver chip tw o PCB patch antennas, and the dc bias through the blue wires B ypass capacitors with large capacitance value (22 F) are used on the PCB to suppress the powe r supply noise The weight of the system shown in the photo is less than 10 gram (0.3 ounce) and can be easily pasted to an upright cardboard facing the target in the experiments. To test the actual power level t ransmitted by the CMOS micro radar chip right before the flip chip transition, Figure 4 10 and Figure 4 11 show the experimental setup including the micro radar with TX antenna gain G t 4 dBi, a horn antenna with G r of 23 dBi, waveguide to coaxial cable a dapters, a 50 75 GHz down convert mixers, and E4448A spectrum analyzer. The spectrum analyzer provides built in LO to down convert the 55 GHz signal to IF.

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60 Figure 4 9 The final system configuration of the 60 GHz micro rada r system in package including the CMOS transceiver chip two PCB patch antennas, and dc biasing through blue wire s. A 55 GHz power source was used to calibrate the loss of the down convert path, and the total loss ( L ) from the adapters, cable, and mixer wa s estimated to be 42 dB. The experiment setup needs to maintain the far field condition, so that Friis transmission equation can be used to calculate the transmitted power. The far field condition is summarized as follows [ 62 ] : (4 2.a) (4 2.b) (4 2.c) where D is the maximum dimension of the antennas. For the single patch antenna on PCB, D t is around 1.8 mm as shown in Figure 4 5, and the horn antenna D r is 40 mm. Based on (4 2), R was chosen to be 0.75 m to ensure the far fie ld condition in the experiment. It should be noticed that in many of the indoor, short distance vital sign detection applications, the distance often results in the near field condition due to the relatively large reflector (human body) at this frequency It is non trivial to use equations to estimate the actual reflected power, for example, due to near field effects such as multi path reflection In these cases, many of the system performance

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61 parameters such as detection distance usually do not have a theo retical value based on calculations as a reference and it mainly relies on experiments to test the system. Figure 4 10. Experimental setup for the TX output power of CMOS transceiver chip. Figu re 4 11. Photo of the experimental setup to test TX transmitted power of CMOS micro radar chip. The distance R is chosen to be 0.75 m to satisfied far field condition. Figure 4 12 shows the screenshot of the received power P r = 82.24 dBm on spectrum ana lyzer. The reso lution bandwidth was lowered from default 3 MHz to 100 KHz to reduce the

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62 noise floor. This helps to distinguish the small received signal on the spectrum at a cost of longer refresh time. Frii s transmission equation including the down conver t loss L can be used to calculate the transmitted power P t : (4 3 ) Figure 4 12 The screenshot of the received power P r = 82.24 dBm on the spectrum analyzer. The resolution bandwidth was low ered from default 3 MHz to 100KHz. By plugging in the known parameters t he transmitted power P t of the micro radar chip is estimated to be about 2.56 dBm. This measured power value is lower than that was designed in Chapter 3.3 ( 2 dBm). Referred to the measured plot in Figure 4 2, th e test indicates the actual IF VCO power fed into the TX up convert mixer might be lower than 10 dBm (differential) which was previously estimated. T he discrepancy might be also due to other unexpected estimation error in e ach parameter of Friss transmission equation. However, from the system application point of view, the test verifies that the TX of CMOS micro radar chip works properly and is ready to be used for the following experiments.

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63 4.5 Mechanical Vibration Detection To verify the I / Q separation by the quadrature receiver and explore the detection range a nd resolution a 0.15 m by 0.15 m flat metal plate was attached to a Zaber T LA60A S actuator and placed at a distance D from the radar. A 1 Hz mechanical vibration wa s generated by the actuator to test the proper I and Q baseband output s at the optimal and null detection points. 4.5.1 Quadrature Channel Test Figure 4 13 show s the detection results directl y sampled by the oscilloscope at D = 0.3 m In Figure 4 13 (A ), the di fferential Q channel s are near the optimal detection point as the baseband output waveforms ( B Q + and B Q ) show the successful detection and the vibration frequency (1 Hz) can be easily read from the time axis. On the other hand, the differential I channels (B I + and B I ) are near the null detection point. T he output waveforms near null detection point have smaller amplitude and do not contain the fundamental tone of the original vibration as explained in Figure 1 3 Figure 4 13 (B) shows the opposite scenari o at a slightly different D which results in the change of total residue phase t Figure 4 13. I and Q baseband outputs test of the micro radar system (A) Q is near optimum detection point and I is near null detection point. (B) I is near optimum detection point and Q is near null detection point.

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64 In this case, the differential I channel s (B I + and B I ) are near the optimal detection point and differential Q channel s (B Q + and B Q ) are near the null detection point. This result indicates the I / Q generation of IF quadrature VCO with passive mixers are wo rking properly, and at least one of the I / Q channels carries the valid detection results for CSD in (1 8) as the detection distance D varies 4.5.2 Sensitivity to Small Vibration In the second experiment to test the detection range and resolution the vibration displacement A was varied a t a fixed D = 0.3 m and the received CSD spectrum was normalized by the largest displacement ( A = 1 mm) as presented in Figure 4 14 (A ). Figure 4 14. The experimental results of s mall mechanical v ibration detection. (A) V arious vibrat ion displacement A at D = 0.3 m (B) S pectrum peak at 1 Hz versus vibration displacement A at various distances. All the data points are normalized to the largest spectrum peak at A = 1 mm, D = 0.3 m.

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65 It shows the vibra tion frequency (1 Hz) can still be detecte d when A is as small as 20 m. Figure 4 14 (B ) displays a more complete comparison through the same normalization. Two observations are made from the plot. First, the received CSD spectrum peak decreases with dista nce as expected. A small vibration with A = 0.2 mm can be detected at D = 2.1 m away. is getting smaller. For example, at D = 0.3 m, as A is reduced by half from 1 mm to 0.5 mm, the CSD spectrum peak drops about 18% ( 1 to 0.82 ); however, as A is reduced by half from 0.2 mm to 0.1 mm, the drop of CSD spectrum peak increases to 42% ( 0.53 to 0.31 ). This implies the detection has an optimal range of target vib ration displacement in term of the detection sensitivity which is worth to be investigated further. 4.6 Heartbeat and Respiration Detection The chest wall movement due to human respiration and heartbeat can be approximated by a two tone sinusoidal vibration a s described in (1 9). In the first test, the CMOS micro radar system was pasted to a ver tically placed cardboard facing the human target The heartbeat signal in Figure 4 15 (A) and (B ) was directly sampled by the oscilloscope as the person sitting on a ch air with his chest wal l 0.3 m in front of the radar The breath was held to avoid the blocking from the harmonics of re s pira tion signal as mentioned in Chapter 1.3. The result shows the robustness of the quadrature architecture to avoid the null detection points, which occur every 1.25 mm ( /4) in detection distance. At the time t = 0 ~ 7 s Q channel was around the optimal point while I channel was near the null point. However, the target entered the null point of Q channel after t = 9 s due to some slight body movement, and I channel started to take o ver the detection. Figure 4 15 (C ) shows the spectrum of detected signal and the heartbeat rate at 69 beats/minute (BPM) after the CSD, which agrees with the result of human counting. The Doppler

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66 frequency shi ft due to body movement also shows up in the spectrum. The total power consumption of the micro radar SiP is 377 mW at 1.2 V power supply. Figure 4 15 Heartbeat detection using the 60 GHz radar when the target holds the breat h at 0.3 m away. The baseband output signal is shown in (A) I channel and (B) Q channel. (C ) CSD spectrum showing accurate heartbeat rate at 69 BPM. The total baseband output noise of the system is band limited by the sampling rate of the baseband ADC. In this case ( Figure 4 15 ) the sampling rate is 50 Hz and the measured baseband noise voltage is around 1 mV rms corresponding to a baseband SNR around 5 (14 dB). This result is consistent with [ 3 ] showing that even the noise figu re of the system could be high around the band of interest due to the large CMOS flicker noise, the vital sign detection is still achievable by proper design of the system. The detection of the respiration and heartbe at simultaneously at 60 GH proves to be challenging due to the severe non linear phase modulation as discussed in C hapter 1.3. As m r is

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67 comparable to or larger than and the modulated phase 4 x ( t ) / in (1 2 ) travels through multiples of 2 and B ( t ) is no longer monotonic during inhale or e xhale. This shows on the spectrum as harmonics and intermodulation which seriously degrades the detection accuracy. Depending on the value or m r in some cases the frequency of the respiration can be distinguishe d on the spectrum, but it is overwhelmed by the harmonics and noise in some other cases The weaker heartbeat signal is usually not able to be read from the CSD spectrum. Figure 4 16 shows two measurement results of the respiration detections as f r is around 15 bea t/minute and m r is slightly varied in the two tests. Figure 4 16 M easur ement results of the respiration detection at 60 GHz as f r = 15 beat/minute D = 0.3, and m r is slightly varied in the two tests. In the top plot of Figure 4 16 the possible respiration p eak around 15 beat/minute (BPM) shows on the CSD spectrum, however there are also many other peaks with larger magnitude which make the detection result in doubt. Sometimes after m r is varied, the main respiration peak

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68 disappears as shown in the bottom pl ot of Figure 4 16 The weaker heartbeat signal at 80 BPM could not be identified in both cases. The 60 GHz vital sign detection difficulty will be discuss in detail is Chapter 5. Theoretical analysis and simulation will be provided to address these issue s and new techniques such a time domain recovery algorithm will be proposed to improve the accuracy of respiration detection.

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69 CHAPTER 5 ANALYSIS AND IMPROVEMENT 5.1 Analysis on 60 GHz Vital Sign Detection The nature of human heartbeat and respiration raise s demodulation issues for 60 GHz vital sign detection as described in Chapter 1.3 and the detection results shown in Chapter 4.5 indicates further theoretical analysis is needed to understand and improve the system Typically the amplitude of chest wall m ovement m r (1 6 mm) due to respiration is at least one order of magnitude larger than m h (0.2 mm) due to heartbea t, and the ratio between m r and (5 mm) no longer satisfies the small angle approximation in (1 3 ). The l arge amplitude difference and stro ng nonlinear phase modulation [ 1 ] significantly increase the complexity of output spectrum and detection difficulty. Human chest wall movement due to heartbeat and respiration can be modeled as a two tone, sinusoida l vibration as in shown in (1 9 ). The baseband output s of I and Q channels can be modified from (1 3): (5 1 .a ) (5 1 .b ) In CSD process, complex S ( t ) is generated by combining I and Q baseband outputs similar to (1 8) and it can be expanded by Bessel function as the summation of its frequency components [ 8 ]: (5 2)

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70 where a r = 4 m r / a h = 4 m h / and J n ( x ) is the n th order Bessel function of the first k ind. The CSD spectrum is obtained by taking FFT of S ( t ), and the constant magnitude of exp ( j t ) no longer affects the detection. The heartbeat and respiration amplitudes with respective to wavelength determine the magnitude of harmonics and intermodulation. A peak at x Hz is proportional to [ 8 ]: (5 3) where k and p are integers satisfying k f r + p f h = x For example, the fundamental respiration peak (R 1 ) located at x = f r ( k = 1 and p = 0) is represented by J 1 ( a r J 0 ( a h ), and the fundamental heartb eat peak (H 1 ) at x = f h ( k = 0 and p = 1) is determined by J 0 ( a r J 1 ( a h ). As illustrated in the Bessel functio n plot in Fig ure 5 1 ( A ), the small m r and m h compared to at 6 GHz (50 mm) stay quite close to the origin on x axis. Ideally any harmonic and in termodulation H x consists of Jn ( a ) with | n | 3 is small enough to be neglected, and this explains the spectrum in Figure 1 4 (A ) showing clear R 1 and H 1 It should be noted that Figure 5 1 only plots J n ( a ) with positive n and J n ( a ) with n < 0 follows the symmetry of Bessel functions [ 8 ]: (5 4) On the other hand, the detection scenario at 60 GHz is quit e different as presented in Figure 5 1 (B ). The heartbeat amplitude m h still stays near the origin on x axis, but the high order terms of J n ( a r ) emerge as m r comparable to resulting in a more complex spectrum. For example, as m r near 5 mm, even J 10 ( a r ) is not negligible and thus generates a prominent peak of R 10 The vital sign spectrum of various m r is simulated in Figure 5 2 with the respiration rate at 15 beat/minute. The follows discuss the 60 respiration and heartbeat, respectively.

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71 Figure 5 1. Theoretical plots of J n ( a ) = J n ( / ) versus vibration amplitude m (A ) At 6 GHz (B ) At 60 GHz Typical value of respiration m r ranges from 1 6 mm, and the heartbeat m h is usually around 0.2 mm. 5.1.1 Respiration Detection Improvement by Two tone Monitoring The fundamental and harmonics of respiration (R 1 R 2 R 3 and etc) can be express as J n ( a r J 0 ( a h ) with n = 1, 2, 3, and etc. Observed from Figure 5 1 (B), a s the value of J 0 ( a h ) is always close to unity due to small a h the respiration harmonics are generally larger than other intermod ulation terms and can be easily id entified on the spectrum in Figure 5 2 However, the detection relying on R 1 is not robust since J 1 ( a r ) shows multiple zero crossing points.

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72 Figure 5 2 Simulated output spectrum of vital sig n detection at 60 GHz (A) At m r = 1.5 mm (B) At m r = 2.1mm (C ) At m r = 4 mm. The f r is at 15 beat/minute. Heartbeat m h is fixed at 0.2 mm with f h = 72 beat/minute As shown in Figure 5 2 (A) and (C ), R 1 vanishes as m r near 1.5 mm and 4 mm, which can be verified by the plot in Figure 5 1 (B) showing the zero crossing points of J 1 ( a r ). Similarly, R 2 disappears as m r near 2.1 mm in Figure 5 2 (B) Observed from Figure 5 1 (B), J 2 ( a r ) approaches local maximum while J 1 ( a r ) is near zero crossing points, and vi ce versa. The important property from the simulation leads to the conclusion that the respiration detection can be improved by monitoring R 1 and R 2 simultaneously. T heoretically the first prominent peak is either R 1 or R 2 as the frequency swept from low to high on the output spectrum, and this detection method is valid

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73 for all values of m r The frequency of other higher order respiration harmonics can be used to distinguish between R 1 and R 2 by division which will be demonstrated in the following experiment s Figure 5 3. V ital sign detection results as the pe rson breathes shallowly at 0.3 m in front of the radar. (A) Time domain waveforms. (B) CSD spectrum. Figure 5 3 shows the vital sign detection of a person sitting 0.3 m in f ront of the radar and breathing shallowly. B I ( t ) and B Q ( t ) are displayed on an oscilloscope ( f s = 25 Hz) and the observation time win dow is 20 s. The results in Figure 5 3 (B ) shows the sallow breath mainly generates the fundamental (R 1 at 13.8 beat/minute ) and second harmonic (R 2 at 27.2 beat/minute), and the detected respiration rate agrees with human counting. The higher order harmonics are not prominent since J n ( a r J 0 ( a h ) with | n | > 3 ar e all small as predicted in Figure 5 1 (B ). From the plot m r can be estimated to be around 1 mm in this test. As the target person breathing deeply at a same rate of 15 beat/minute, Fig ure 5 4 ( A ) and ( B ) shows the I and Q baseband outputs. The observation time window was increased to 50 s

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74 and f s was 10 Hz. It is noted that in B I ( t ) and B Q ( t ), the modulated phase term 4 x ( t )/ in (1 2 ) travels through multiples of 2 due to the large chest wall movement m r resulting in much more co mplex time domain waveforms compared to previous experiment of shallow breath Figure 5 4. V ital sign detection results as the pers on breathes deeply at 0.3 m in front of the radar (A) T ime domain waveform of I channel (B) T ime domain waveform of Q channel. (C ) CSD spectrum The observation time is increased to 50 s, but only 20 s is shown h ere for the comparison with Figure 5 3 Figure 5 4 ( C ) shows the output CSD spectrum where the frequency resolution is improved by the lo nger o bservation time. As discussed earlier in this chapter ideally the prominent peak at 30 beat/minute is either fundamental (R 1 ) or second harmonic (R 2 ) of respiration. Since the

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75 frequency of higher order harmonics around 105 beat/minute (R 7 ) and 135 b eat/minute (R 9 ) are not dividable by 30 beat/minute, it is concluded that the peak around 30 beat/minute is R 2 and fundamental respiration frequency (R 1 ) is at f r =15 beat/minute. However, as presented in (5 3 ), theoretically there are expected to be more prominent harmonics such as J 4 ( a r J 0 ( a h ) and J 5 ( a r J 0 ( a h ) on the spectrum, which are n ot shown in the result of Figure 5 4 (C ). One of the possible reasons is that human respiration and heartbeat movements are not pu rely sinusoidal as modeled in (1 9 ) for simple analysis. The respiratio n signal itself is actually composed of fundamental tones and certain higher order harmonics, which results in the discrepancy of higher order behavior of the Bessel function 5.1.2 Heartbeat Detection The fundamental and harmonics of heartbeat (H 1 H 2 H 3 and etc ) can be expressed as J 0 ( a r J n ( a h ) with n = 1, 2, 3, and etc. Normally the relatively small value of m h n ear the origin on x axis in Figure 5 1 (B ) makes all J n ( a h ) too small to be distinguished as respiration is present. In addition, the heartbeat peaks are also affected by respir ation term J 0 ( a r ). In some rare cases as J 0 ( a r ) is near peak value ( m r small peak of H 1 can be s een on the spectrum as in Figure 5 2 (A ). If J 0 ( a r ) is near its zero crossing points, it makes the alre ady weak H 1 more unlikely to be read from the output spectrum Currently the heartbeat detection at 60 GHz is obtained by holding the breath to avoid respiration harmoni cs on the output spectrum as explained in Chapter 1.3 Detection from the back is an al ternative way to reduce the interference from respiration. 5.2 Proposed Time domain Recovery Algorithm The technique monitoring the first and second harmonic peaks on the spectrum helps to improve the success rate and accuracy of respiration detection. Howeve r, observed in some extreme cases of the experiments, both the first and second respiration peaks may not be

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76 distinguishable on the spectrum due to the environmental noise and system non ideality as captured in Figure 4 16 (B). Especially when the chest wa ll movement has larger deviation from the ideal sinusoidal waveform as modeled in (1 9), the magnitude of each harmonic and intermodulation term may not follow the theoretical prediction by Bessel function curves plotted in Figure 5 1 (B). In another word, the target displacement comparable to or larger than such as respiration movement generates numerous peaks on the output spectrum and t he demodulation based on the recognition of Bessel harmonic peaks on the spectrum is susceptible to any small system non ideality and noise This makes the algorithm improvement in frequency domain difficulty and not robust. One thought is to unwrap the non linear modulated phase term from time domain first, and then the accuracy on frequency domain peak recognition migh t be further improved 5.2.1 Analysis on Quadrature Baseband Outputs Doppler radar system has a non linear input ou t put mapping when the displacement is The relation can be seen by the plot in Figure 5 5 modified from Figure 1 3 O n baseband out put B ( t ) two types of peaks (transition between positive and negative slope ) can be identified. Figure 5 5 Non linear input output mapping whe n the vibration is comparable to at 60 GHz.

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77 Type I peaks are due to the distor tion of nonlinear system mapping (peak A and B), and Type II peak is corresponding to the peak of real displacement (peak C). Plotting both I and Q channels in Figure 5 6 the following relations can be found: 1. As a Type I peak happens on B I ( t ), the sign o f slope on B Q ( t ) remains unchanged (and vice versa). 2. As a Type II peak happens, the sign of slope on B I ( t ) and B Q ( t ) change simultaneously. This indicates the peak is due to the original displacement. 3. For any adjacent Type I peaks (peak A and B) on B I ( t ), the corresponding sign of slope on B Q ( t ) is opposite (and vice versa). Thus if the two types of peaks on B I ( t ) and B Q ( t ) can be distinguished by software, the baseband output can be un folded to show the real target displacement and the peak of intere st such as respiration can be recovered on the final CSD spectrum Figure 5 6 Non linear input output mapping of I and Q channels when the vibration is comparable to at 60 GHz ( B Q ( t ) is not shown in the figure)

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78 5.2.2 MATLAB Program Implementation The time domain signal recovery algorithm based on the above observation was implemented in MATLAB to improve the accuracy of respiration detection. By simply monitoring the s ystem baseband outputs B I ( t ) and B Q ( t ), the algorithm keeps the non distorted portion of wavefo rm from original vibration (in F ollow mode) and recovers the dis torted portion of waveform (in F l ip mode) as demonstrated in Figure 5 7 No other intermediate r eference signals are needed as the algorithm converts B I ( t ) and B Q ( t ) directly to B I ( t ) and B Q ( t ) linear input output mapping to a approximately linear transfer function. Although the system mapp ing after the phase un wrapping is still not completely linear due to the nature of sinusoidal shape the main frequency component of the original displacement x ( t ) can be largely recovered in the final baseband outputs which will be shown in the followi ng sections Figure 5 7. Time domain recovery technique by simply monitoring I and Q baseband outputs when the vibration is comparable to at 60 GHz ( B Q ( t ) and B Q ( t the figure).

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79 Figure 5 8 illustrates a simple technique to perform the continuous Flip and Follow operations in the code It can be observed from Figure 5 7 that the switch between Flip mode and Follow mode is only triggered by Type I peaks (peak A and B), and these two modes are always alternating regardless of the direction of the displacement x ( t ). In Figure 5 8 (A), the algorithm follows every point in segment A 1 B 1 makes no change to the waveform. As soo n as the algorithm detects peak B 1 it starts to vertically flip the first point of segment B 1 C with respect to the previous value and update all the points after this current point. In this manner the algorithm maintains the continuity of the waveform. I t repeats the same Flip operation point by point in segment B 1 C and segment C B 2 as illustrated in Figure 5 8 (B) and Figure 5 8 (C) Since peak C is not a Type I peak, it does not trigger the switch between Flip and Flow modes After the algorithm detect ing Type I peak B 2 the program switch back to Follow mode and the two Type I peaks (B 1 and B 2 ) generated by the non linear system transfer function are eliminated from the original waveform. Figure 5 8. Illustration of the c ontinuous Flip and Follow operations. (A) Follow segment A 1 B 1 (B) Flip segments B 1 C and C B 2 (C) Follow segment B 2 A 2

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80 Figure 5 9 presents the simplified flow chart of the time domain recovery algorithm. After importing B I ( t ) and B Q ( t ) moving average is applied to remove the glitches of the waveform due to noise and reduce the chance of peak misjudgment. In the same loop which go through every point of the waveform B I ( t ) and B Q ( t ) are processed in parallel to increase the efficiency. Figure 5 9 Simplified flow chart of the time domain recovery algorithm. Similarly B I ( t ) and B Q ( t ) represents the original waveforms and B I ( t ) and B Q ( t shown in Appendix

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81 The details of the two sub functions a re presented in Appendix Based on the three relations described in Chapter 5.2.1, once a valid peak is detected on B I ( t ) and there is a consecutive trend (no peak) on B Q ( t ) it is identified as a Type I peak on B I ( t ) which triggers the switch between the Flip and Follow mode on B I ( t ) (and vice versa on B Q ( t ) ) The algorithm makes no change to the Type II peaks, which preserves the original target displacement. 5.2.3 Experimental Results R eferred to the previous outputs in Figure 4 16 (B), the respiration dete ction failed and no prominent fundamental or second harmonic can be read from the spectrum. In the experiment, t he same baseband signal s B I ( t ) and B Q ( t ) of Figure 4 16 (B) were used to test the effectiveness of the time domain recovery algorithm Figure 5 10 compares the baseband outputs before and after the proposed time domain recovery algorithm and as expected the respiration waveform s c lose to the original chest wall movement are recovered on B I ( t B Q It should be notice that from the wavefor ms of recovered B I ( t B Q it verifies that the respiration movement in this cas e is not very close to the single tone pure sinusoidal waveform as modeled in ( 1 9 ). T hus the harmonics and intermodulation terms on the original spectrum in Figure 4 16 (B) do not completely follow the Bessel function analysis in Chapter 5.1. Figure 5 11 shows the CSD spectrum after applying the time domain recovery algorithm Compared to the original spectrum in Figure 4 16 (B), it shows a prominent respiration peak at 15.11 beat/minute, which closely agrees with the results of human counting. This demonstrates the recovery algor ithm can significantly increase the accuracy and robustness of 60 GHz Doppler micro radar detection.

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82 Figure 5 10 Respiration detection outputs before and after the recovery algorithm is applied The respiration waveforms close to the original chest wall movement are recovered on B I ( t ) and B Q ( t by the algorithm Figure 5 11 Recovered r espiration peak compared to the original spectrum in Figure 4 16 (B ) The pro minent peak at 15.11 beat/minute shows the accuracy and robustness of the detection have been improved.

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83 5.2.4 Discussion Figure 5 12 shows the results of recovery algorithm test on different respiration pattern. In the experiment the respiration was controlled by the subject as he inhaled for 2 s exhaled for 2 s paused for 3 s, and repeat ed the cycle. The overall respiration rate should be 60/(2+2+3) = 8 .57 beat/minute. This respiration pattern is no longer single tone, pure sinusoidal movement, and the harmonic and intermodulation terms on the spectrum cannot be predicted by the Bessel function analysis in Chapter 5.1. The experimental results show the r espiration movement can successfully be recovered by the time domain algorithm, even the target has an arbitrary movement pattern. Figure 5 12 Respiration detection outputs before and after the recovery algorithm is applied. The subject inhaled for 2 s, exhaled for 2 s, paused for 3 s, and repeated the cycle. The heartbeat signal can be seen in the interval s between the respiration s Figure 5 13 presents the CSD spectrum before and after applying the recovery algorithm. In Fig ure 5 13 (A), the result shows a relatively low signal to noise ratio (SNR) on the spectrum, and unknown peaks such as the one at 6.6 BPM appear due to the environmental and system

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84 noise. After the time domain recovery algorithm, the prominent respiration peak emerges at 8.65 BPM as shown in Figure 5 13 (B). The SNR on the spectrum is greatly enhanced, which proves to increase the accuracy and robustness of the respiration detection. Figure 5 13. CSD spectrum outputs be fore and after the recovery algorithm is applied. The pro minent peak at 8.65 beat/minute is fairly close the real respiration rate (8.57 beat/ minute), which shows the detection accuracy is improved. From the viewpoint of the recovery algorithm in Figure 5 12 aro und the time period of 7 s to 4 s, 0.5 s to 3.5 s, and after 7.5 s, the program indicates consecutive Follow mode rather than alternating Flip and Follow modes. The result implies the fluctuations of Doppler modulated phase in (1 2) during th ese time per iods is less than indicating the target does not have a large displ acement comparable to As mentioned in Chapter 1.3, the respiration movement is comparable or larger than and thus the small target movement in these consecutive Follow periods only captures the heartbeat signal with some small random body movement. Figure 5 14 duplicates Figure 5 13 to show the consecutive Follow periods in di cated by the recovery

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85 algorithm. One of the time periods marked by the blue circle captures only the heartbea t with small random body movement, corresponding to the paused respiration period. If FFT is applied to the waveform in the blue circle, the heartbeat rate at 72 beat/minute can be obtain correctly as shown in Figure 5 15. This detection result agrees well with the heartbeat rate measured by a wrist band monitor as a reference. Figure 5 1 4. Duplicate of Figure 5 13 showing the cons ecutive Follow periods indicated by the recovery algorithm. The time period marked by blue circle contains only the heartbeat signal with small random body movement. Figure 5 15. Spectrum of the waveform marked by blue circle in Figure 5 14 which shows the correct heartbeat rate detection result

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86 As analyzed in Chapter 5. 1.2, the simultaneous detection of respiration and heartbeat using 60 GHz Doppler radar proves to be difficult due to the intermodulation terms of the Bessel function. However, the discussion above indicates the possible detection in a normal breathing sit uation. In fact, except for some extreme cases such as heavy breathing right after excises, the normal human breathing contains certain time interval between each respiration cycle, especially for the vital sign monitoring during sleep. The time domain rec overy algorithm indicates those which capture only the small movement including heartbea t and body movement. Even if each interval is as short as 2 3 seconds, the heartbeat rate can be extracted from the wa veform fragment as shown in the above experiments In the real time vital sign monitoring, the heartbeat rate detection result can be updated at every time the interval is detected. This method provides a potential solution for the 60 GHz Doppl er micro radar to simultaneously detect the heartbeat and respiration in most of the normal situations. 5.3 Broadband Antenna on LTCC System in Package 5.3.1 Introduction As antenna size shrinks with radar frequency, previous chapters present the 60 GHz CMOS micro r adar with PCB patch antennas to achieve a low cost, fully integrated vital sign sensor. The use of planar microstrip antennas facilitates the mm wave flip chip attachment, and the low profile is suitable for many portable applications such as tablet s and s martphones. For a typical edge fed patch antenna, the choice of dielectric thickness faces the tradeoff between antenna bandwidth and adequate width of 50 feed line [ 60 ]. Although the bandwidth requirement is not critical in terms of using a single tone, fixed frequency radar signal, a wideband antenna is desired to improve yield when modeling and manufacturing variations are present. The LTCC ( low temperature co fired ceramic ) substrate with multiple metal layers is

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87 used to make a better compromise among the bandwidth, radiation efficiency, and proper microstrip width for flip chip transition design For vital sign detection systems utilizing multiple radars facing each other such as for rand om body movement cancellation [ 5 ] [ 63 ], direct coupling from the TX of one radar to RX of another needs to be minimized for interference and DC offset considerations Orthogonal linearly polarized antennas can b e used to alleviate the issue [ 5 ], however, as the number of radars increased, circularly polarized antennas are needed. The right hand circularly polarized (RHCP) antenna at RX provides better isolation against other radars with left hand circul arly polarized (LHCP) TX antennas. A s mentioned in Chapter 3.6.2 and 4.3 the thinnest substrate (12 7 m) available was chosen to minimize the 50 microstrip width ( 360 m) shown in Fig ure 4 5 which sac rifices the antenna bandwidth [ 60 ]. Since the required flip c hip G S G S G pitch is only 150 m, narrow 50 feed line is highly preferable for the flip chip transition design, and it also reduces the spurious radiation which disturbs the antenna pattern. As shown in Fig ure 4 4 and Figure 4 8 of previous chapter the 10 dB return loss bandwidth of the single patch antenna is limited to around 2.4 %, and the simulated 3 dB gain bandwidth is only about 5 % Figure 5 16 shows a case where the measured return loss deviates from the targeted frequency around 55 GHz due to the manufacturing variation of antenna fabrication. In the example, the n arrow 10 dB antenna bandwidth barely covers the t argeted frequency around 55 GHz At the PCB etching precision aroun d 5 % the antenna center frequency shift of 2 3 GHz was observed in the worst case. Taking into account the process variation of CMOS chip, the actual radar operating frequency is possibly to fall out of the optimal performance range of the patch antenna in some worst case scenari os. In this sec tion the circularl y polarized sequential rotation 2 2 patch antenna array is

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88 i ntroduced to increase the gain and bandwidth performance compared to the linearly polarized single patch antenna. Figure 5 16 Measu red and simulated return loss of a single pat ch PCB antenna as the manufacturing variation is present. 5.3.2 Sequential Rotation Patch Antenna Array The multilaye r LTCC substrate is studied for the design of wide band patch antenna. The propose d layer profile is shown in Figure 5 17 The process provides 11 metal layers (L 1 L 11 ) with a thickness of 10 m each, and the relative dielectric constant is 3.9. The antenna elements utilize L 1 and the ground (GND) is on L 6 which forms a to tal dielectric thickness of 328 m to improve the ante nna bandwidth. However, as the surface wave loss and coupling increases with the dielect ric thickness [ 60 ], the design compromise has to be made between bandwidth and radiation efficiency. The microstrip feed line and its GND are on L 1 and L 2 wit h a dielectric thickness of 65 m to keep a narrow feed line width. All GND layers are connected by metallic via fences. For the RF signal, a through substrate via is employed to provide a vertical transition betwe en the chip and the antenna. Six grounded vias are located around the through substrate via to mimic a coa xial transmission line effect [ 64 ] In addition, the LTCC serves as the compact

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89 packaging for the micro radar chip. The c hip and dc bias lines are located at the bottom (L 11 ), reducing the interference to the antenna and making the area usage more efficient. Figure 5 17 Layer profile presents the LTCC system in package including 11 metal layers (L 1 L 11 ) and the FR4 board with a slot for CMOS chip. In addition to increasing the substrate thickness of the antenna, sequential rotation techniques [ 65 ] are exploited to further improve the operation bandwidth. Figure 5 18 (A) shows the configuration of two sequential rotation 22 arrays in LTCC with opposite polarizations for transmitting (LHCP) and receiving (RHCP), respectively. The radiating elements are single feed corner truncated square patches which are sequentially rotated and fed by the sequential phase networks. The zoom in area shows the G S G S G pad s for the flip chip integration which is at the bottom (L 11 ). Figure 5 18 ( B ) shows the detail topology of the multi ports feed network. The input port has a charact eristic impedance of Z 0 = 50 The four output ports are designed to provide balanced signals with equal magnitude and an incremental 90 phase delay. Each output port is connected to the antenna element through a /4 transformer, which results in an inpu t impedance of Z A Two meander lines with electrical lengths of 90 and 270 not only provide

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90 180 phase difference, but also perform the quarter wavelength t ransformation to achieve the impedance matching between the input port and the antennas. Figure 5 18 Broadband antenna design on LTCC. (A ) Circularly polarized se quential rotation arrays (B ) Detail topology of the sequential phase feed network. Thus, the characteristic impedance Z t is determined by (5 5) Figure 5 19 shows the return loss and TX/RX isolation of the sequential rotation array. To allow on wafer measurement, the probing pads and the antenna arrays need to be located on the

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91 same side of the LTCC subs trate. Therefore, additional vertical transitions are used to connect the bottom microstrip lines to the top layer as shown in the inset in Fig ure 9. The effects of the additional transitions are included in both simu lated and measured results. The measure d 10 dB return loss bandwidth achieves 12.8 % Fig ure 9 also shows the simulated isolation around 30 dB at 55 GHz without the additional transitions. Figure 5 19 Measured and simulated return losses (S 11 ) and TX/RX isolation ( S 12 ) of the antenna array. The experimental setup for the radiation patterns is the same as the previous antenna design. The circularly polarized gain and axial ratio were obtained by measuring the complex fields of two orthogonal linearly polarized compo nents followed by post processing [ 66 ]. Figure 5 20 shows the LHCP realized gain and the axial ratio at zenith against frequencies. It is observed that the 3 dB axial ratio bandwidth is about 10%. The peak gain of the array is around 9.7 dBi at 54 GHz with 3 dB gain bandwidth around 12%. The radiation patterns in XZ and YZ plane at 54 GHz are shown in Fig ure 5 21 The measured LHCP gain of the main beam is 9.7 dBi with a cross polarization discrimination of about 27 dB at zenit h. Compared to the single

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92 patch antenna, the wider bandwidth gives the system a better chance to operate in the optimal antenna performance range as modeling and manufacturing variations are present. Besides, as the circularly polarized antenna arrays appl ied in multiple sensor network facing each other, it provides better isolation against multiple interference sources, which is not achievable by the orthogonal linearly polarized antennas. Figure 5 20 Realized gain and axial ratio (AR) spectrums of the broadband sequential rotation patch antenna array at zenith Figure 5 21 Radiation patterns of the patch antenna array. (A) XZ plane (B ) YZ plane.

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93 5.3.3 Vital Sign Detection Figure 5 22 shows the final system assembly attached to an upright cardboard facing the target. The CMOS radar chip is on the other side of LTCC, and DC bias is provided by the FR4 board through the wire and solder bump underneath the LTCC. Compared to the previous SiP shown in Figu re 4 18, the LTCC substrate provides a more area efficient and flexible mm wave integration, since all the surface components including flip chip integrated CMOS chip and bypass capacitors are on the bottom side of the antenna. The entire top side area is used for the antenna array, which potentially can be integrated with an antenna array with more number of elements in the future applications. Figure 5 22 Top view of the final system assembly The flip chip integrated CMOS radar chip and surface mounted bypass capacitors are placed on the other side of LTCC. As a person sitting in front of the radar at D = 0.3 m, the baseband outputs B I ( t ) and B Q (t) as previously shown in Figure 3 13 are directly sampled by an analog to di gital converter (ADC) at a sampling frequency f s for a period of time t and CSD is used to eliminate the null detection points. Figure 5 23 (A ) shows the successful heartbeat detection by the LTCC radar system in package at f s = 50 Hz and t = 20 s, where the person has to hold the breath to avoid the interference by large respiration harmonics The detected heartbeat rate at 71 beat/minute agrees with human counting. The respirat ion detection is plotted in Figure 5 23 (B ) with f s = 10 Hz and

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94 t = 50 s It c an be seen that t he spectrum resolution is improved by the longer measurement time. Due to the nonlin ear Doppler phase modulation some harmonics of respiration signal are prominent, but the fundamental respiration rate at 16 beat/minute can be successfull y read from the spectrum Figure 5 23 CSD output spectrum of vital sign detection using the broadband patch antenna array on LTCC. (A) Heartbeat detection. (B) Respiration Detection

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95 CHAPTER 6 SUMMARY The 60 GHz micro rada r SiP including the CMOS transceiver chip, flip chip integration, and antennas was designed and tested. The system achieves high level of integration by the flip chip transition and small antenna size at mm wave frequency range. The down conversion gain of the transceiver chip is measured to be 36 dB at 55 GHz. Successful detection of small mechanical vibration with a displ acement at 0.3 m away and the detection range reaches 2.1 m if the displacement increases to 200 The quadrature system architecture utilizing CSD solves the null detection point problem o f Doppler radar without extra frequency tuning, which assures the robust detection against detection distance change. The use of 60 GHz radar frequency offers various advantages such as higher sensitivity and smaller antenna size comp ared to lower frequency systems, however, the respiration amplitude comparable to wavelength causes strong non linear phase modulation, and relatively small heartbea t amplitude results in demodulation difficulties T heoretical analysis and simulation of 60 GHz detection are provided to address these issues. Both shallow an d deep breathings are tested in the experiments, and the detection technique monitoring both the fundamental and second harmonic of respiration is proposed. In addition, the signal recovery algorithm is proposed to imp rove the accuracy of vital sign detect ion at 60 GHz The algorithm demonstrates the respiration movement can be successfully recovered in time domain, even it is an arb itrary respiration pattern whose harmonics on spectrum could not be predicted by the simple Bessel function analysis. The ci rcularly polarized sequential rotation antenna array integrated in the 60 GHz SiP is im plemented on the LTCC substrate. The LTCC technology with multiple metal layer structure is able to satisfy the different dielectric thickness requirements of antennas a nd microstrip feed

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96 lines, and it als o provides a compact mm wave pack aging option for the CMOS chip. Compar ed to linearly polarized single patch antenna, the 10 dB b andwidth of the antenna array on LTCC increases from 1.3 GHz (2.4 % ) to 7 GHz (12.8 % ) at 5 5 GHz, and narrow 50 feed line is obtained to rea lize the flip chip transition. As the process and manufacturing variations are often present in the mm wave systems, wide antenna bandwidth is able to cover the possible frequency drift and increase the sy stem yield. This work demonstrates the first v ital sign detection by the flip chip integrated CMOS radar at 60 GHz. The shorter wavelength offers significant area reduction and flexibility in system integration. It can be readily embedded into one of the smartphone functions, for example, making it a pervasive first aid tool for noncontact vital sign monitoring, or ap plied to a large sensor network for different vibration monitoring

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97 APPENDIX MATLAB CODING OF TIME DOMAIN RECOVERY ALGO RITHM clc; clear al l ; close all ; Data = csvread( 'C: \ Users \ Jason \ Desktop \ Measurement \ Full_system_0308 \ scope_255.csv' ,2 ,0); fs = 1/(Data(2,1) Data(1,1)) %Extract sampling freuquency fs. % ---------------------------------------------------------------------% Remove dc component. Q_p and I_p: positive baseband outputs of the Q % and I channels. Q_m and I_m: % negtive baseband outputs of the Q and I channels. Q_p = Data(:,2) mean(Data(:,2)); Q_m = Data(:,3) mean(Data(:,3)); I_p = Data(:,4) mean(Data(:,4)); I_m = Data(:,5) mean(Data(:,5)); Q_p_raw = Q_p; %Store the raw data for later comparison. I_p_raw = I_p; % ---------------------------------------------------------------------% Choose to turn on or off the 3 levels of data smoothing. The simple % moving average acts like a low pass filter to reduce the effect of noise % and prevent the misjudgment of recovery algorithm. Depending on the % quality of raw data, more filters can be turned on to improve the % recovery accuracy. for i = 2:length(I_p) 1 % Low pass filter for I_p I_p(i) = ( I_p(i 1)+I_p(i+1) )/2; end for i = 2:length(I_p) 1 I_p(i) = ( I_p(i 1)+I_p(i+1) )/2; end for i = 2:length(I_p) 1 I_p(i) = ( I_p(i 1)+I_p(i+1) )/2; end for i = 2:length(Q_p) 1 % Low pass filte r for Q_p Q_p(i) = ( Q_p(i 1)+Q_p(i+1) )/2; end for i = 2:length(Q_p) 1 Q_p(i) = ( Q_p(i 1)+Q_p(i+1) )/2; end for i = 2:length(Q_p) 1 Q_p(i) = ( Q_p(i 1)+Q_p(i+1) )/2; end % --------------------------------------------------------------------figure plot(Data(:,1), I_p, r' ); % Plot I_p before the recovery algorithm

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98 hold on ; h_axis=gca; get(h_axis, 'FontSize' ); % displays the default Font size set(h_axis, 'FontSize' ,16); % sets the font size of axis AX = legend( 'B_I(t)' ); LEG = fi ndobj(AX, 'type' 'text' ); set(LEG, 'FontSize' ,16) grid; ylabel( 'Voltage(V)' ); h_ylabel = get(gca, 'YLabel' ); set(h_ylabel, 'FontSize' ,16); xlabel( 'Time(s)' ); h_xlabel = get(gca, 'XLabel' ); set(h_xlabel, 'FontSize' ,16); title( 'Time domain Vital Sign Recovery' ); h = get(gca, 'title' ); set(h, 'FontSize' 16); plot(Data(:,1), Q_p, b' ); % Plot Q_p before the recovery algorithm % ---------------------------------------------------------------------% Find the trend of I_p and Q_p signal and save the value to T rd_Ip and % Trd_Qp. 1 represents the voltage level is rising, and 0 represents the % voltage level is falling. Trd_Ip(1) = 1; %Assign arbitrary trend to the first index for i = 2:length(I_p) if I_p(i) > I_p(i 1) Trd_Ip(i) = 1; elseif I_ p(i) < I_p(i 1) Trd_Ip(i) = 0; else Trd_Ip(i) = ~Trd_Ip(i 1); %If I_p(i) = I_p(i 1), inverse Trd_Ip %to prevent a false trend. It %eliminates the slope = 0 c ase. end end Trd_Qp(1) = 1; %Assign arbitrary trend to the first one for i = 2:length(Q_p) if Q_p(i) > Q_p(i 1) Trd_Qp(i) = 1; elseif Q_p(i) < Q_p(i 1) Trd_Qp(i) = 0; else Trd_Qp(i) = ~Trd_Qp(i 1); %If Q_p(i) = Q_p(i 1), inverse Trd_Qp %to prevent a false trend. It %eliminates the slope = 0 case. end end % --------------------------------------------------------------------% Start of the recovery process.

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99 % I_p_fix and Q_p_fix represent the signal after recovery. Initially % I_p_fix = I_p and Q_p_fix = Q_p; I_p_fix = I_p; Q_p_fix = Q_p; % IF flip == 1, it is in flip status. IF flip == 0, it is in follow status. % Initially set flip = 1. flip_I = 1; flip_Q = 1; % Assign arbitrary initial value to previos peak and slope type as long as % they are not equal to the value 0, 1, and 1. PT_I_Last = 10; Sp_Q_Last = 10; PT_Q_Last = 1 0; Sp_I_Last = 10; for i = 4:length(Trd_Ip) 3 [PC_I PT_I] = PeakCheck(Trd_Ip, i, 1); % Check if there is a valid % peak on I_p. [TC_Q Sp_Q] = TrendCheck(Trd_Qp, i, 1); % Check if there is a % consistent trend on Q_p % If yes, a Type I peak is % identified on I_p. if PC_I && TC_Q && (Sp_Q_Last ~= Sp_Q) flip_I = ~flip_I; % Switch flip to follow stage or follow % to flip stage. if flip_I == 1 I_p_fix(i:end) = I_p_fix(i:end) 2*(I_p(i) I_p(i 1))*ones(length(I_p(i:end)),1); %Flipping I_p and store the data to I_p_fix. end PT_I_Last = PT_I; %Store the current peak and sloep type. Sp_Q_Last = Sp_Q; else if flip_I == 1 I_p_fix(i:end) = I_p_fix(i:end) 2*(I_p(i) I_p(i 1))*ones(length(I_p(i:e nd)),1); end end [PC_Q PT_Q] = PeakCheck(Trd_Qp, i, 1); % Check if there is a valid % peak on Q_p. [TC_I Sp_I] = TrendCheck(Trd_Ip, i, 1); % Check if there is a % consistent trend on I_p % If yes, a Type I peak is % identified on Q_p if PC_Q && TC_I && (Sp_I_Last ~= Sp_I) flip_ Q = ~flip_Q; % Switch flip to follow stage or follow % to flip stage. if flip_Q == 1 Q_p_fix(i:end) = Q_p_fix(i:end) 2*(Q_p(i) Q_p(i 1))*ones(length(Q_p(i:end)),1); %Flipping Q_p and store the data to Q_p_fix.

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100 end PT_Q_Last = PT_Q; %Store the current peak and sloep type. Sp_I_Last = Sp_I; else if flip_Q == 1 Q_p_fix(i:end) = Q_p_fix(i:end) 2*(Q_p(i) Q_p(i 1))*ones(length(Q_p(i :end)),1); end end end plot(Data(:,1), I_p_fix, k' ); %Plot the recovered waveforms. plot(Data(:,1), Q_p_fix, g' ); AX = legend( 'B_I(t)' 'B_Q(t)' 'B_I(t)' 'B_Q(t)' ); hold off ; % ---------------------------------------------------------------------% Start complex signal demodulation [5] CSD1 = I_p_fix + j*Q_p_fix; % Recovered data CSD3 = I_p_raw + j*Q_p_raw; % Use raw data before the waveform smoothing. L = 2^20; %FFT number of point [H1t, W1t] = dtft(CSD1, L); H1 = H1t(L/2+1:L); W1 = W1t(L/2+1:L); [H3t, W3t] = dtft(CSD3, L); H3 = H3t(L/2+1:L); W3 = W3t(L/2+1:L); % -----------------------------------------------------------------------figure; plot (W3/pi*(fs/2)*60, abs(H3), b' ); % Plot the baseba nd spectrum before % recovery h_axis=gca; get(h_axis, 'FontSize' ); % displays the default Font size set(h_axis, 'FontSize' ,16); % sets the font size of axis xlim([0 150]); grid; ylabel( 'CSD Spectrum' ); h_ylabel = get(g ca, 'YLabel' ); set(h_ylabel, 'FontSize' ,16); xlabel( 'Beat/Minute' ); h_xlabel = get(gca, 'XLabel' ); set(h_xlabel, 'FontSize' ,16); title( 'Baseband Spectrum Before Recovery' ); h = get(gca, 'title' ); set(h, 'FontSize' 16); figure; plot (W1/pi*(fs/2)*60, abs (H1), r' ); % Plot the baseband spectrum after

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101 % recovery h_axis=gca; get(h_axis, 'FontSize' ); % displays the default Font size set(h_axis, 'FontSize' ,16); % sets the font size of axis xlim([0 150]); grid; ylabel( 'CSD S pectrum' ); h_ylabel = get(gca, 'YLabel' ); set(h_ylabel, 'FontSize' ,16); xlabel( 'Beat/Minute' ); h_xlabel = get(gca, 'XLabel' ); set(h_xlabel, 'FontSize' ,16); title( 'Baseband Spectrum After Recovery' ); h = get(gca, 'title' ); set(h, 'FontSize' 16); % ---------------------------------------------------------------------function [PC PT] = PeakCheck(Trend, Idx, N) % Idx: center index, % Centered at Idx, check N indexes to the left and N 1 to the right to % determine if it is a valid peak. N>=1. % If it is a valid peak, PC==1. If it is not a valid peak, PC==0. % If the peak type is rising first and then falling, PT == 1. % If the peak type is falling first and then rising, PT == 0. PC = 0; PT = 0; i = 1; while i <= N if (Trend(Idx) == Trend(Idx+ i 1)) && (Trend(Idx) ~= Trend(Idx i)) if i == N PC = 1; if Trend(Idx+i 1) == 0 PT = 1; else PT = 0; end end i = i+1; else PC = 0; break end end % ------------------------------------------------------------------------function [TC Sp]= TrendCheck(Trend, Idx, N) % Idx: center index, % Centered at Idx, check N inde xes to the left and N 1 indexes to the right % to determine if there is a consistent trend. N>=1. % If there is a consistent trend, TC == 1. If there is not a consistent % trend, TC == 0.

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102 % If the slope of the trend > 0, Sp == 1. If the slope of the t rend < 0, % then Sp == 1. % The generate of trend functions Trd_Ip and Trd_Qp has eliminated the % slope = 0 case. TC = 1; i = Idx N+1; while (TC == 1) && (i <= Idx+N 1) if Trend(Idx N) == Trend(i) TC = 1; if Trend(i) == 1 Sp = 1; else Sp = 1; end i = i+1; else TC = 0; Sp = 0; break end end % -----------------------------------------------------------------------function [H, W] = dtf t(h, N) %DTFT calculate DTFT at N equally spaced frequencies % ---% Usage: [H, W] = dtft(h, N) % % h : finite length input vector, whose length is L % N : number of frequencies for evaluation over [ pi,pi) % ==> constraint: N >= L % H : DTFT values (complex) % W : (2nd output) vector of freqs where DTFT is computed % --------------------------------------------------------------% copyright 1994, by C.S. Burrus, J.H. McClellan, A.V. Oppenheim, % T.W. Parks, R.W. Sc hafer, & H.W. Schussler. For use with the book % "Computer Based Exercises for Signal Processing Using MATLAB" % (Prentice Hall, 1994). % --------------------------------------------------------------N = fix(N); L = length(h); h = h(:); %< -for vect ors ONLY !!! if ( N < L ) error( 'DTFT: # data samples cannot exceed # freq samples' ) end W = (2*pi/N) [ 0:(N 1) ]'; mid = ceil(N/2) + 1; W(mid:N) = W(mid:N) 2*pi; % < --move [pi,2pi) to [ pi,0) W = fftshift(W); H = fftshift( fft( h, N ) ); %< --move negative freq components

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103 LIST OF REFERENCES 1. C. Li, Y. Xiao, an power Ka band Theory Tech. vol. 54, no. 12, pp. 44 6 5 4471, Dec. 2006. 2. A. D. Droitcour, V. M. Lubecke, J. Lin, and O. Boric IEEE MTT S Int. Microw. Symp. Digest pp. 17 6 1 78, May, 2001. 3. sen sitivity software configurable 5.8 GHz contact vital sign d Trans. Microw. Theory Tech., vol. 58, no 5, pp. 1410 1419, May 2010. 4. C. Li, and J. Lin, "Non contact measurement of periodic m ovem ents by a 22 40 GHz radar sensor using nonlinear phase m odulation, IEEE MTT S Int. Microw. Symp. Digest, pp. 579 582, June 2007. 5. C. Li, and J. Lin, "Complex signal demodulation and random body movement cancellation techniques for non contact vital sign d e tection," IEEE MTT S International Microwave Symposium, pp. 567 570, Atlanta, June, 2008. 6. S. Nicolson, P. Chevalier, A. Chanter B. Sautreuil and S. P. Voinigescu "A 77 79 GHz Doppler radar transceiver in silicon", IEEE Compound Semiconduct. Integr. Circu its Symp., pp.1 2007 7. S. Kim and C. Nguyen "On the development of a multifunction millimeter wave sensor for displacement sensing and low velocity measurement", IEEE Trans. Micro w. Theory Tech., vol. 52, no. 52, pp.2503 2512 Nov. 2004 8. Y. Yan, L. Catt afest etho ds and realization of a real vol. 59, no. 12, pp. 3556 3566, Dec 2011. 9. Keyence LK GSeries Laser Displacement Sensor User Manual, Keyence Corporation, Woodcliff Lake, NJ. 10. K. Abe, K. Otsuka, and J. mixing laser Doppler vibrometry with high optical sensitivity: Application to real time sound reproduction, 8.1 8.9, 2003. 11. K. Otsuka, K. Abe, J. Y. Ko and T. time nanometer vibration measurement with a self mixing microchi p solid vol. 27, no. 15, pp. 1339 1341, 2002. 12. Y. Yan, C. Li, ibro MTT S Int. Microw. Symp. Baltimore, MD, June 5 11, 2011.

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109 BIOGRAPHICAL SKETCH Mr. Te Yu Jason Kao receive d the B.S. degree in electrical and control engineering from National Chiao Tung University, Hsin Chu, Taiwan R.O.C. in 2004 and M.S. degree in electrical engineering from the University of Washington, Seattle WA in 2008. He received his Ph.D. degree in e lectrical and computer engineering at the Uni ver sity of Florida in the spring of 2013 From January to August 2011, he worked as a graduate intern technical at Intel Corporation in Chandler, AZ. He worked on signal integrity for hi gh speed IO circuits incl ud ing PCI Express Gen2 interface. In the Ph.D. study, h is research interests include RF and millimeter wave CMOS cir cuit and system passive component design and EM modeling, millimeter wave packaging, Doppler radar sensors, antennas, and bio medical appli cations of RF systems. Mr. Kao is currently a student member of IEEE and is the Student Paper Competition finalist in IEEE 2012 Radio Frequency Integrated Circuits (RFIC) Symposium, Montreal, Canada