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1 ADVANCED METAMATERIAL CIRCUITS FOR MICROWAVE AND MILLIMETER WAVE APPLICATIONS By DAVID ELIECER SENIOR A DISSERTATION PRESENTED TO THE GRADUATE SCHOOL OF THE UNIVERSITY OF FLORIDA IN PARTIAL FULFILLMENT OF THE REQUIREMENTS FOR THE DEG REE OF DOCTOR OF PHILOSOPHY UNIVERSITY OF FLORIDA 2012
2 2012 David Eliecer Senior
3 To my wife and daughter Chen Chen and Annabeth, my parents Lercy and Eliecer, and my sisters Aminta y Lercy
4 ACKNOWLEDG E ME NTS This work was supported in part by the National Science Foundation ( NSF ) ECCS 1132413 The following institutions in Colombia also supported my graduate studies: Universidad Tecnol gica de Bolivar, COLCIENCIAS, De partamento Nacional de Planeaci n (D NP) and Fulbright Commission I would like to star t by thank ing my chairman and advisor Dr. Yong Kyu Yoon, whose continuous support encouragement and guidance provided me with many opportunities during these five years to expand my knowledge, improve my technical and communication skills and have a clear and insightful understanding of my research Dr. Yoon has not only been an excellent mentor, but also a friend for me and my family, first at the University at Buffalo and then, at the University of Flor ida. This work would not have been possible without his guidance, support and patience. I would also like to thank the rest of my committee members Dr. David Arnold Dr. Jenshan Lin and Dr. Peng Jiang for their valuable suggestions and their time to serv e as the reviewers of this research. This work would not be possible without the support of the current and previous members of Multidisciplinary nano and Microsystems Laboratory that is part of the Interdisciplinary Microsystem s Group (IMG). I would espec ially like to thank Dr. Jungkwun Kim, Xiaoyu Cheng, Pitfee Jao, Cheolbok Kim, Melroy Machado and Dr Kyoung Tae Kim for their continuous assistance on the simulation, fabrication and characterization of my devices, as well as the valuable technical discuss ions. I am thankful to Jessica Meloy who gave me the first training on the milling machine and has been very helpful with the organization and maintenance of IMG labs.
5 This work could not be possible without the assistance of the Nanoscale Research Facili ty members Al Ogden, Bill Lewis, David Hayes and Brent Gilla. I am grat e ful to all of them for their training on microfabrication and the continuous technical support I would also like to thank my colleagues in Colombia, Professors Jorge Duque, Eduardo G o mez, Enrique Vanegas, Oscar Acua, Gonzalo L pez, Jose Luis Villa and Ricardo Arjona for their academic support during this time. In the same way, I would like to thank the president of Universidad Tecn logica de Bolivar, Dr. Patricia Martinez Barrios for her great vision of the future of the university and for giving me the opportunity and the financial support to study abroad. During the last year of my research, I have been especially supported and inspired by the love of my wife Chen Chen and my beauti ful daughter Annabeth Nicole They have brought the happiness and joy I need in the most difficult moments. Finally, I am grateful to my parents Lercy and Eliecer, my sisters Lercy and Aminta and my parents in law, Xia n gliang and Guihua for their continuou s love and support.
6 TABLE OF CONTENTS Page ACKNOWLEDGEMENTS ................................ ................................ ................................ .............. 4 LIST OF TABLES ................................ ................................ ................................ .......................... 11 LIST OF FIGURES ................................ ................................ ................................ ........................ 12 LIST OF ABBREVIATIONS ................................ ................................ ................................ ......... 18 ABSTRACT ................................ ................................ ................................ ................................ .... 19 CHAPTER 1 INTRODUCTION ................................ ................................ ................................ ....... 21 1.1 Basic Concepts on Electromagnetic Metamaterials ................................ ........ 22 1.2 Towards Compact Practical Metamateria l Applications ................................ 25 1.3 Motivation ................................ ................................ ................................ ....... 27 1.4 Research Objectives and Contributions ................................ .......................... 29 1.5 Dissertation Organization ................................ ................................ ................ 31 2 ELECTROMAGNETIC METAMATERIALS ................................ ........................... 34 2.1 Theoretical Prediction of Metamaterials ................................ ......................... 35 2.2 Negative Refraction ................................ ................................ ......................... 38 2.3 Phase Velocity and Group Velocity ................................ ................................ 40 2.4 Experimental Demonstration of Metamaterials ................................ .............. 40 2.4.1 Negative Permittivity Medium ................................ .......................... 41 2.4.2 Negative Permeability Medi um ................................ ......................... 42 2.4.3 Left Handed Material Demonstration ................................ ................ 45 2.5 Transmission Line Approach of Metamaterials ................................ .............. 48 2.5.1 The Left handed Transmission Lines ................................ ................ 49 2.5.2 The Composite Right/Left handed Transmission Lines .................... 51 2.5.3 Periodic Structure Implementation ................................ .................... 54 2.6 Review of CRLH Transmission Line Applications ................................ ........ 57 2.6.1 Metamaterial Couplers and Filters ................................ ..................... 57 2.6.2 Multiband Components ................................ ................................ ...... 57
7 2.6.3 Metamaterial Resonators ................................ ................................ ... 58 2.6.4 Compact Multilayer Components ................................ ...................... 59 2.6.5 CRLH Substrate Integrated Waveguide Components ....................... 60 2.6.6 Micr omachined CRLH Applications ................................ ................. 60 3 MICROMACHINED TRANSMISSION LINES ................................ ....................... 62 3.1 Bulk Micromachined Transmission Lines ................................ ...................... 65 3.2 Surface Micromachined Transmission Lines ................................ .................. 68 3.3 Multilayer and Microfabricated CRLH Transmission Lines Applications ..... 71 3.4 Proposed Multilayer Unit Cell for the Implementation of CRLH Metamaterial Applications ................................ ................................ .............. 76 3.5 Proposed Multilayer Embedded Substrate Integrated Waveguide Filter Architecture ................................ ................................ ................................ ..... 78 3.6 Proposed Multilayer Architecture for Micromachined Wideband Bandpass Filters ................................ ................................ .............................. 80 3.7 SU8 and BCB as Dielectric Materials for RF Circuits ................................ .... 81 4 COMPOSITE RIGHT/LEFT HANDED (CRLH) METAMATERIAL APPLICATIONS ................................ ................................ ................................ ......... 83 4.1 Compact Dua l Band Three Way Bagley Polygon Power Divider Using CRLH Transmission Lines ................................ ................................ .............. 83 4.1.1 Compact Bagley Polygon Power Divider ................................ .......... 84 4.1 .2 Dual Band CRLH Transmission Line Theory Review ...................... 85 4.1.3 Design of the CRLH and Conventional Dual Band Quarter Wavelength Transmission Lines. ................................ ....................... 88 4.1.4 Implementation. ................................ ................................ ................. 90 4.1.5 Summary ................................ ................................ ............................ 92 4.2 Surface Micromachined CRLH Unit Cell on SU8 for Microwave Appli cations ................................ ................................ ................................ ..... 93 4.2.1 Unit Cell Structure and Modelling ................................ ..................... 94 4.2.2 Implementation of CRLH Unit Cells. ................................ .............. 102 4.2.3 Fabrication Process ................................ ................................ .......... 105 4.2.4 Measurement Results ................................ ................................ ....... 108 4.2.5 Summary ................................ ................................ .......................... 110 4.3 Bridged Composite Right/Left Handed Unit Cell with All Pass Behavior .. 111 4.3.1 Proposed Bridged CRLH ................................ ................................ 112 4.3.2 Analysis ................................ ................................ ............................ 113
8 4.3.3 Physical Implementation ................................ ................................ .. 115 4.3.4 Summary ................................ ................................ .......................... 118 5 MICROMACHINED METAMATERIAL UNIT CELLS ON BCB ........................ 119 5.1 Analysis of Finite Ground Plane Coplanar Waveguide Transmission Lines on BCB ................................ ................................ ................................ 120 5.2 Design and Implementation of the Dual Band CRLH Unit Cell and Transmission Line ................................ ................................ ......................... 122 5.2.1 Loss Analysis ................................ ................................ ................... 125 5.3 Fabrication ................................ ................................ ................................ ..... 126 5.4 Measurement Results ................................ ................................ .................... 129 5.5 Summary ................................ ................................ ................................ ....... 131 6 METAMATERIAL LOADED THE HALF MODE SUBSTRATE INTEGRATED WAVEGUIDE ................................ ................................ ................ 132 6.1 Single and Dual Band Bandpass Filters Using CSRR Loaded Half Mode Substrate Integrated Wav eguide ................................ ................................ .... 133 6.1.1 Theoretical Backround ................................ ................................ ..... 133 6.1.2 Proposed CSRR Loaded Half Mode Substrate Integrated Waveguide Evanescent Mode R esonators ................................ ....... 141 6.1.3 Two Pole Filter Implementation and Measurement Results ............ 148 6.1.4 Summary ................................ ................................ .......................... 150 6.2 Electrically Tunable Evanescent Mode Half Mode Substrate Integrated Waveguide Resonators ................................ ................................ .................. 151 6.2.1 The Tunable CSRR loaded HMSIW Resonator .............................. 151 6.2.2 Implementation and Measurement Results ................................ ...... 155 6.2.3 Summary ................................ ................................ .......................... 158 6.3 Dual Band Filters Using CSRR and Capacitive Loaded Half Mode Substrate Integrated Waveguide ................................ ................................ ... 158 6.3.1 The Dual Band CSRR and Capacitive Loaded HMSIW Resonator ................................ ................................ ......................... 159 6.3.2 Dual Band Bandpass Filter Design and Measurement Results ........ 161 6.3.3 Summary ................................ ................................ .......................... 163 6.4 Wireless Passive Sensing Application Using a Cavity Loaded Evanescent Mode HMSIW Resonator ................................ .......................... 164 6.4.1 The Evanescent Mode Resonator ................................ ..................... 165 6. 4.2 Proposed Sensor Structures ................................ .............................. 1 66
9 6.4.3 Effect of the Air Gap in the Resonance Frequency ......................... 167 6.4.4 Mechanical Simulation of the Deflection ................................ ........ 168 6.4.5 Wireless Interrogation ................................ ................................ ...... 170 6.4.6 Measurement Results ................................ ................................ ....... 170 6.4.7 Summary ................................ ................................ .......................... 171 7 MICROMACHINED EMBEDDED EVANESCENT MODE HMSIW BANDPASS FILTER ................................ ................................ ................................ 173 7.1 3D SU8 Embedded Resonator Design ................................ .......................... 174 7.2 Two Pole Embedded Filter Design ................................ ............................... 179 7.3 Fabrication Process ................................ ................................ ........................ 180 7.4 Measurement Results ................................ ................................ .................... 183 7.5 Summary ................................ ................................ ................................ ....... 184 8 EVANESCENT MODE BROADBAND BANDPASS FILTERS ........................... 186 8.1 Broadband Bandpass Filters using CSRR Loaded Eighth Mode Substrate Integrated Waveguide Cavities ................................ ...................... 187 8.1.1 General Requirements for Broadband Ba ndpass Filter Design ....... 187 8.1.2 The Eighth Mode SIW Cavity ................................ ......................... 188 8.1.3 The CSRR loaded Eighth Mode SIW Cavity ................................ .. 190 8.1.4 Resonator Analysis and Design ................................ ....................... 191 8.1.5 Two Pole Filters Designs ................................ ................................ 192 8.1.6 Results and Discussion ................................ ................................ .... 194 8.1.7 Summary ................................ ................................ .......................... 194 8.2. Surface Micromachined Broadband Millimeter Wave Bandpass Filters Using CSRR Lo aded Quarter Mode Substrate Integrated Waveguide Cavities ................................ ................................ ................................ .......... 195 8.2.1 The CSRR loaded Quarter Mode Substrate Integrated Waveguide Cavities ................................ ................................ ......... 197 8.2.2 The Design of CSRR loaded QMSIW Bandpass Filters on Flexible LCP ................................ ................................ .................... 197 8.2.3 The Design of CSRR loaded QMSIW Bandpass Filters on BCB ................................ ................................ ................................ 205 8.2.4 Summary ................................ ................................ .......................... 211 9 CONCLUSIONS ................................ ................................ ................................ ....... 212 9.1 Summary of Research Contributions ................................ ............................ 214
10 9.2 Future Work ................................ ................................ ................................ .. 214 9.3 List of Related Publications ................................ ................................ .......... 216 APPENDIX : MICROMACHINED FABRI CATION PROCEDURES ON BENZOCYCLOBUTENE ................................ ................................ ..................... 218 LIST OF REFERENCES ................................ ................................ ................................ .............. 221 BIOGRAPHICAL SKETCH ................................ ................................ ................................ ........ 230
11 LIS T OF TABLES Table Page 3 1 Properties of the dielectric materials ................................ ................................ ............... 82 4 1 Parameters of the unit cell. ................................ ................................ .............................. 89 4 2 Summary of power divider measurements. ................................ ................................ ..... 92 4 3 Design parameters of the MIM capacitors ................................ ................................ ...... 99 4 4 Design parameters of the inductor ................................ ................................ ................. 100 4 5 Parameters of the unit cells ................................ ................................ ............................ 104 4 6 Di mensions of the unit cells in m ................................ ................................ ................ 105 5 1 Dimensions in m of the CRLH unit cell on BCB. ................................ ....................... 122 6 1 Dimensions of the propos ed HMSIW CSRR resonators ................................ .............. 145 6 2 Calculated parameters and dimensions of the two pole HMSIW CSRR filters ............ 148 6 3 Paramet ers of the varactor diode ................................ ................................ ................... 153 6 4 Dimensions in mm of the resonator. ................................ ................................ .............. 166 6 5 Summary of measurements results ................................ ................................ ................ 171 7 1 Dimensions of the resonator ................................ ................................ .......................... 178 7 2 Parameters of the embedded filter ................................ ................................ ................. 179 7 3 Comparison of resonator size in different technologies ................................ ................. 184 8 1 Specification and calculated parameters of the filters ................................ .................... 193 8 2 Design specification for the filters on LCP ................................ ................................ .... 198 8 3 Design parameters of the filters on LCP ................................ ................................ ........ 202 8 4 Design specifications and calculated parameters of the filters on BCB ......................... 206 A 1 Curing of BCB ................................ ................................ ................................ ................ 220
12 LIST OF FIGURES Figure Page 1 1 W ave propagation. ................................ ................................ ................................ ........... 22 1 2 D emonstration of left handed materials in free space... ................................ .................. 23 1 3 Classification of electromagnetic materials.. ................................ ................................ .. 24 1 4 Planar composite right/left handed transmission line. ................................ ................... 26 1 5 Ove rview of the dissertation structure ................................ ................................ ............. 33 2 1 Reflection and refraction of EM waves in the RH and LH media. ................................ 39 2 2 Negative permittivit y medium. ................................ ................................ ....................... 41 2 3 Negative permeability medium. ................................ ................................ ..................... 43 2 4 Variation of the permeability of the SRR with the frequency.. ................................ ....... 44 2 5 Split ring resonator. ................................ ................................ ................................ ......... 45 2 6 Left handed materials in free space.. ................................ ................................ ............... 47 2 7 Mod eling of a uniform right handed transmission line. ................................ .................. 49 2 8 Modeling of a uniform left handed transmission line ................................ ..................... 49 2 9 Planar composit e right/left handed transmission line. ................................ ................... 51 2 10 CRLH transmission line. ................................ ................................ ................................ 52 2 11 CRLH dual band components. ................................ ................................ ....................... 58 2 12 CRLH SIW slot antennas. ................................ ................................ ............................. 60 3 1 Cross section of some transmission lines used in microwave and millimeter wave circuits ................................ ................................ ................................ ........................... 63 3 2 Bulk micromachined transmission lines ................................ ................................ ......... 66 3 3 S urface micromachined transmission lines.. ................................ ................................ ... 69 3 4 Unit cells of multilayer CRLH transmission lines.. ................................ ........................ 72 3 5 Implemented CRLH transmission line on SU8. ................................ ............................. 75
13 3 6 M ultilayer CRLH unit cell on SU8 or BCB ................................ ................................ ... 77 3 7 Cross section of the proposed dielectric embedded resonators and filters. ..................... 79 3 8 Cross section of t he proposed micromachined cavity resonators and filters. ................. 82 4 1 Conventional Bagley p olygon three way power divider. ................................ ................ 85 4 2 CRLH pha se response.. ................................ ................................ ................................ ... 87 4 3 Dual band quarter wavelength transmission line with shunt connections of open and short stubs.. ................................ ................................ ................................ ...................... 89 4 4 Layout o f the 57.74 dual band CRLH /4 transformer.. ................................ ............. 90 4 5 I mplemented dual band Bagley p olygon power dividers.. ................................ .............. 91 4 6 Return loss (S11) and insertion loss (S21 or S 41) for two implemented Bagley p olygon power dividers. ................................ ................................ ................................ ................ 91 4 7 Insertion loss at each port for the two implemented Bagley p olygon power dividers. .... 92 4 8 The CRLH unit cell. ................................ ................................ ................................ ........ 95 4 9 MIM capacitors.. ................................ ................................ ................................ ............. 98 4 10 Equivalent electrical circuit m odel of the two cascaded MIM capacitors. ...................... 98 4 11 Simulation of extracted parameters for the MIM capacitors.. ................................ ......... 99 4 12 Embedded meande r line inductor.. ................................ ................................ ................ 100 4 13 Inductor modeling.. ................................ ................................ ................................ ....... 101 4 14 Extracted parameters for the inductor.. ................................ ................................ ......... 102 4 15 Complete equivalent electrical circuit of the CRLH unit cell including parasitic contributions. ................................ ................................ ................................ ................. 103 4 16 General representation of the u nit cell structure. ................................ .......................... 103 4 17 Comparison of the electromagnetic and circuital simulation for the broadband CRLH unit cell with a 90 phase at 2.4 GHz.. ................................ ................................ .......... 106 4 18 Fabrication process for the multilayer CRLH devices. ................................ ................. 107 4 19 Photographs of the microfabricated broadband CRLH unit cell ................................ 109
14 4 20 Measurement setup consisting of a Cascade Microtech probe station and an Agilent E8361A VNA. ................................ ................................ ................................ ............... 109 4 21 I nsertion and return loss for a balanced microfabricated CRLH unit cell .................... 110 4 22 Dispersion relation for the microfabricated CRLH unit cell.. ................................ ....... 110 4 23 Proposed topology.. ................................ ................................ ................................ ....... 112 4 24 Physical configuration of the B CRLH unit cell.. ................................ ......................... 115 4 25 Simulated and measured results of the B CRLH unit cell. ................................ ........... 117 4 26 Triband two B CRLH unit cells /4 open stub.. ................................ ........................... 117 5 1 Cross section view of the CPW structures.. ................................ ................................ .. 121 5 2 Insertion l oss of a CPW line on a low resistivity silicon substrate with a BCB interface layer ................................ ................................ ................................ .............................. 121 5 3 Electric field in the cross section for a CPW line with W = 55 m, G = 20 m. .......... 122 5 4 Multilayer CRLH unit cell on BCB. ................................ ................................ ............. 123 5 5 Extracted parameters for the MIM capacitors on BCB.. ................................ ............... 124 5 6 Simulated performance of the unit cell.. ................................ ................................ ....... 124 5 7 Simulated loss factor of the unit cell for ideal dielectric, ideal conductor, lossless silicon, radiation loss and total loss. ................................ ................................ .......................... 125 5 8 Current distribution in the CRLH unit cell.. ................................ ................................ .. 125 5 9 Fabrication process of the unit cell.. ................................ ................................ .............. 127 5 10 S EM Image of the CRLH unit cell. ................................ ................................ ............... 128 5 11 Photomicrographs of the microfabricated devices. ................................ ....................... 128 5 12 Measured performance. ................................ ................................ ................................ 129 5 13 Measured phase constant of the two unit cells transmission line showing a dual band behavior around 12.8 GHz and 36 GHz. ................................ ................................ ....... 130 6 1 Geometry of the rectangular waveguide. ................................ ................................ ...... 135 6 2 Simulated electric field distribution for the TE 10 mode in a rectangular waveguide. ... 136 6 3 The substrate integrated waveguide.. ................................ ................................ ............ 137
15 6 4 The half mode substrate integrated waveguide (HMSIW).. ................................ .......... 139 6 5 Electric field distribution of the dominant TE mode ................................ ................... 140 6 6 A typical frequency response for the SIW and HMSIW structures.. ............................ 140 6 7 Proposed HMSIW resonator with a series of vias for electric walls and complementary split ring resonator (CSRR) on the top surface.. ................................ ........................... 141 6 8 Simulated results for a convention al split ring resonator.. ................................ ............ 142 6 9 Simulated results for a CSRR loaded HMSIW resonator.. ................................ ........... 144 6 10 External quality factor Q e of the C SRR loaded HMSIW resonator. ............................. 146 6 11 Internal coupling coefficient.. ................................ ................................ ....................... 147 6 12 Extracted internal coupling coefficients.. ................................ ................................ ...... 148 6 13 T wo pole bandpass filters.. ................................ ................................ ............................ 148 6 14 Measurement and simulation results. ................................ ................................ ............ 149 6 15 Fabricated resonators and filters.. ................................ ................................ .................. 150 6 16 The electrically tunable resonator.. ................................ ................................ ............... 152 6 17 Simulated results.. ................................ ................................ ................................ ......... 154 6 18 Photograph of the fabricated tunable resonators.. ................................ ......................... 155 6 19 Measured results with applied DC voltage.. ................................ ................................ .. 156 6 20 Measured results with applied DC voltage for the resonator C. ................................ ... 157 6 21 Dual band resonator.. ................................ ................................ ................................ ..... 160 6 22 Simulated performance of the dual band resonator.. ................................ ..................... 161 6 23 Proposed dual band filter. ................................ ................................ .............................. 162 6 24 Simulated and measured perf ormance of the implemented devices.. ........................... 162 6 25 Photographs of the fabricated dual band devices.. ................................ ........................ 163 6 26 Simulated and measured resu lts for the resonator used in our study. ........................... 165 6 27 Proposed sensor configurations. ................................ ................................ ................... 167 6 28 Variation of resonance frequency as a fu nction of the air gap ................................ ...... 167
16 6 29 Mechanical simulation for the deflection as a function of an applied pressure. ........... 168 6 30 Broadband ant enna. ................................ ................................ ................................ ....... 169 6 31 Measured results. ................................ ................................ ................................ ........... 171 6 32 Fabrication and test.. ................................ ................................ ................................ ..... 172 7 1 Electrical equivalent circuit of the implemented CSRR loaded HMSIW resonator ..... 175 7 2 Cross section of the proposed SU8 embedded resonator. ................................ ............. 176 7 3 3D view of the embedded resonator. ................................ ................................ ............. 177 7 4 Simulated performance of the resonator.. ................................ ................................ ..... 178 7 5 External Q factor vari ation with the input distance l ................................ .................... 179 7 6 Top view of the two pole embedded filter. ................................ ................................ ... 180 7 7 Electromagnetic simulation results of the two pole filter.. ................................ ............ 180 7 8 Fabrication process. ................................ ................................ ................................ ....... 181 7 9 Scanning electron microscopy (SEM) images of the embedded resonator and filt er.. 183 7 10 Measurement results. ................................ ................................ ................................ ..... 185 8 1 The eighth mode substrate integrated waveguide (EMSIW). ................................ ....... 189 8 2 Parameters of the EMSIW cavities.. ................................ ................................ ............. 191 8 3 Characterization of the EMSIW cavities.. ................................ ................................ ..... 192 8 4 Two pole filters.. ................................ ................................ ................................ ........... 193 8 5 Measured and simulated results.. ................................ ................................ .................. 195 8 6 The quarter mode substrate integrated waveguide (QMSIW) cavities.. ....................... 198 8 7 Extracted external quality factor ( Q e ) of the resonator. ................................ ................. 200 8 8 Extracted internal coupling coefficient k .. ................................ ................................ ..... 200 8 9 Simulated frequency response of a single ended CSRR loaded QMSIW cavity.. ........ 201 8 10 Physical layout of the bandpass filters on LCP.. ................................ ........................... 202 8 11 S imulated results for the two pole bandpass filter on LCP.. ................................ ......... 203
17 8 12 Simulated results for the three pole bandpass filter on LCP.. ................................ ....... 204 8 13 Proposed fabrication process of the LCP filters. ................................ ........................... 205 8 14 Extracted external quality factor of a CSRR loaded QMSIW cavity on BCB. ............. 207 8 15 Extracted internal coupling coefficient for magnetically coupled CSRR loaded QMSIW cavities on BCB. ................................ ................................ ................................ ............ 208 8 16 Layout of the proposed filters.. ................................ ................................ ..................... 208 8 17 Simulated frequency response of the two pole 60GHz bandpass filter on BCB. .......... 209 8 18 Simulated fr equency response of the four pole 60 GHz bandpass filter on BCB. ........ 210
18 LIST OF ABBREVIATIONS BCB Benzocyclobutene B CRLH Bridged Composite right/ left handed CRLH Composite right/ left handed CSR R Complementary split ring resonator DNG Double negative EMSIW Eighth mode substrate integrated waveguide HMSIW Half mode substrate integrated waveguide LCP Liquid crystal polymer LH Left handed PCB Printed circuit board QMSIW Quarter mode substrat e integrated waveguide RH Right handed SIW Substrate integrated w aveguide SRR Split ring resonator TL Transmission line
19 Abstract of Dissertation Presented to the Graduate School of the University of Florida in Partial Fulfillment of the Req uirements for the Degree of Doctor of Philosophy ADVANCED METAMATERIAL CIRCUITS FOR MICROWAVE AND MILLIMETER WAVE APPLICATIONS By David Eliecer Senior December 2012 Chair : Yong Kyu Yoon Major : Electrical and Computer Engineering The exploration of va rious new architectures and fabrication techniques for implementing miniaturized metamaterial circuits for radio frequency (RF) applications is presented T he possibility of using low resistivity silicon glass, liquid crystal polymer (LCP) and conventiona l printed circuit board (PCB) organic substrates for compact advanced metamaterial applications is addressed The design, modeling, simulation and fabrication process es of the compact metamaterial devices are discussed in detail As a first step the c omp osite r igh t / left handed (CRLH) approach for implementing metamaterial s RF circuits is applied to the design of dual band application s using commercial lumped elements on a conventional PCB Next a multilayer surface micromachined fabrication process that utilizes the negative tone photopatternable epoxy SU8 and the negative tone photopatternable resin Benzocyclobutene (BCB) as dielectric interface layers on low cost organic carrier substrates is employed for implementing highly compact CRLH transmission l ines for broad band and dual band operation up to 40 GHz. Our study shows that SU8 and BCB are good candidates for implementing compact metamaterial applica tions. The study continues with the implementation of compact resonators that make use of reduced m ode versions of the substrate integrated waveguide (SIW) for narrow band, wideband
20 and dual band bandpass filter s Th e h alf m ode SIW (HMSIW) quarter mode SIW (QMSIW) and eight mode SIW (EM SIW) loaded with a metamaterial particle, the c omplementary s plit r ing r esonator (CSRR), are proposed to implement compact bandpass filters working below the original waveguide cutoff frequency. Theoretical analysis and experimental demonstration are provided for b andpass filters working at S and X frequency bands on a co nventional PCB substrate. Additional experimental implementations include a surface micromachined SU8 embedded CSRR loaded HMSIW b andpass filter working at 12 GHz At the end, the proposed cavities are also applied for the design of a set of bandpass filt ers for operation at 25 GHz and 60 GHz using the flexible substrate LCP and the BCB resin as dielectrics. Finall y, since the conventional printed circuit board (PCB), low resistivity silicon and glass are selected as the supporting substrates for the mic romachined CRLH devices and filters; the compatibility with conventional microwave PCB implementations and CMOS integrated circuits is maintained.
21 CHAPTER 1 INTRODUCTION In the last decade, the scien tific and engineering communities developed a great interest in a new area of study: Metamaterials. By definition, metamaterials are artificially created materials exhibiting unusual electromagnetic properties not readily found in nature. Mostly based on arrays of periodically organized structures, namely unit cells, their unusual electromagnetic properties represented by the electric permittivity the magnetic permeability and the refractive index n make them interesting for devising new and creative applications for physics and engineering Wirele ss communication s great ly benefit from this new area, not only with the development of bulky metamaterials created to behave as conventional macroscopic materials, but also with planar and non planar implementations of new applications working at microwav e, millimeter wave and terahertz frequencies. Moreover, modern wireless communication technologies demanding new applications, miniaturized devices with enhanced performance, integration with CMOS/MEMS/MMIC processes and the integration with digital circui try, seem to be the motivation for the continuous exploration of new metamaterials circuits, systems and concepts in engineering. In this chapter basic concepts on electromagnetic metamaterials are visited. Section 1.1 i ntroduces the electromagnetic metamaterials Section 1.2 presents a review of the pioneer work on metamaterials for RF applications, emphasizing on the Composite Right / Left handed (CRLH) Transmission Line (TL) a pproach for implementing planar and multilayer metamaterials. Section 1.3 discusses th e motivation of this research Section 1.4 provides the research objectives and finally, sectio n 1.5 summarizes the dissertation organization.
22 1.1. Basic Concepts on Electromagnetic Metamaterials stence of substances with simultaneously negative and of substances are called left handed (LH) ones to express the fact they propagate electromagnetic waves in which the electric field, the magnetic fie ld and the phase constant ve ctors build a left handed triad; in comparison with the conventional materials, also called right handed (RH) ones where the triad is right handed as illustrated in Figure 1 1 H owever it was after three decades that Smith et al  succeeded in the first demonstration of engineered left handed materials by using negative permittivity thin wires (TW) and the negative permeability split ring resonators (SRR) previously proposed by Pend ry et al [3 ] but with limited number of unit cells and measurements in a waveguide environment. Later, Osbay et al.  performed the first experimental demonstration of left handed materials in free space by using intercalated planar arrays of split rin g resonators and thin wires, as illustrated in Figure 1 2 Although this resonant approach is lossy and narrow band ed to be of practical interest for engineering applications, it opened a new research area for mult iple and unique applications on microwave, millimeter wave, terahertz and optics, placing the concept as a good candidate for the next generation devices. Figure 1 1 Wave propagation. A ) Right han ded medium. B ) Left handed medium A B
23 Figure 1 2 Demonstration of left handed materials in free space A ) The left handed struct ure c onsisting of planar implementations on printed circuit board (PCB) of i ntercalated arrays of SRRs and TWs. B) The transmission and reflection response s of the left ha nded or double negative (DNG) material ( Adaptation and reprint of Figures 1 and 4 from E. Ozbay, K. Aydin, E. Cubukcu and M. Bayindir, Transmission and reflect ion properties of composite double negative metamaterials in free space, IEEE Tran s. Antennas and Propagat vol .51, no.10, pp. 2592 2595, Copyright 2003 with permission from IEEE). To further understand the concept of metamaterials, Figure 1 3 illustrates the classification of a medium according to the possible sign combinations in the pair ( ). It is important to understand that the response of a system to the electromagnetic field is mainly determined by the properties of t he constitutive material [5, 6 ]. These properties are described by the macroscopic parameters of the material: The electric permittivity and the magnetic permeability These properties are related to the refractive index n by (1 1) w here r and r are the relative permittivity and permeability related to the free space permittivity and permeability by 0 = / r = 8.854 10 12 F/m and 0 = / r = 4 10 7 H.m 1 respectively. For conventional right handed materials, the refractive index n is positive, while for the left handed materials it is negative. A B x y z
24 Figure 1 3 Classification of electromagnetic materials. ( Adaptation of Figure 1 from C. Caloz and T. Itoh Metamaterials for High Frequency Electronics, Proceedings of the IEEE v ol.93, no.10, pp.1744 1752, Copyright 2005 with permission from IEEE). A medium with both positive permittivity and permeability ( > 0, > 0) is known as the Double Positiv e material (DPS) or R ight H anded material (RH), which is well known naturally occurring conventional isotropic dielectrics [ 5 ]. A medium with negative permittivity and positive permeability ( < 0, > 0) is known as the Epsilon Negative medium (ENG). Some plasmas and metals at optical frequencies behave in this manner. Conventional ferromagnetic and gyrotropic materials behave as a medium with positive permittivity and negative permeability ( > 0, < 0), which is called the Mu Negative medium (MNG) [5 ]. Some DPS, ENG and MNG artificial materials exist in nature and were also demonstrated. A medium with Conventional Dielectric materials Refracted wave RH/forward wave propagation > 0, > 0 I Air Incident wave Reflected wave Plasmas below cutoff, Metals at high frequencies Wire structu re Evanescent wave propagation < 0, > 0 Air Incident Reflected Ferrites, Ferrimagnetic materials, Split ring resonators Evanescent wave propagation > 0, < 0 Air Incident Reflected I I I V Left handed (LH) Materials < 0, < 0 Refracted LH/Backward wave propagation I II Air Incident Reflected
25 both negative permit tivity and permeability is called as a Double Negative medium (DNG) or Left Handed medium (LH), which is the material theorized by Vese lago  There is no clear evidence of this kind of material occurring naturally, but it can be constructed artificially, which can be categorized as a new class of materials. Such Left Handed materials are classified as Metamaterials from the electromag netic material viewpoint where they are artificially fabricated in order to be effectively homogeneous ( p < g /4) and exhibit unusual electromagnetic properties, such as negative permittivity, permeability and refractive index. In Chapter 2 a more detail ed study of metamaterials is provided. 1.2. Towards Compact Practical Metamaterial Applications In the search for practical implementation of metamaterial s for engineering applications, i n 2002, the transmission line approach for implementing planar metamaterials on microstrip technology was proposed almost simultaneously, by t hree different groups [ 7 10 ]. The approach le d to the conception of planar negative refractive index transmission lines or composite right / left handed (CRLH) transmission lines as shown in Figure 1 4 Due to their controllable unusual properties, such as non linear dispersion, broad bandwidth, infinite wavelength regime and positive phase shift, among others, this kind of transmission lines can be carefully designed to satisfy specific requirements depending on the particular application. After their introduction, many new different applications have been implemented by using either surface mounted devices ( SMD ) or integrated planar components, confirming the advantages of using negative properties in microwave engineering such as size reduction, backward propagation, bandwidth enhancemen t zeroth and negative order resonance s and multiband operation [ 11 34 ]. The nee d for additional size reduction and the integration with 3D and multilayer devices extended the transmission line approach to the implementation of super compact multilayer CRLH
26 transmission lines in the frame of conventional printed circuit board processe s and ceramic LTCC process es, where MIM capacitors and meander lines are mostly used [ 26,27 ] These well established PCB and LTCC technolog ies are greatly benefitted from the new CRLH architectures for miniaturized microwave device and system implementatio n Figure 1 4 Planar composite right/left handed transmission line A) Microstrip implementation of a 5 unit cells CRLH Transmission line ( Adaptation of Figure 8 from A. Lai, T. Itoh and C. Caloz Composite right/left handed transmission line metamaterials," IEEE Microwave Mag vol.5, no.3, pp. 34 50, Copyright 2004, with permission from IEEE ). B) Unit cell ( Reprint of Figure 5 from C. Caloz and T. Itoh, hand ed (LH) structures and microstrip realization of a low IEEE Trans. Antennas Propagat ., vol. 52, no. 5, Copyright 2004, with permission from IEEE ). At the same time, t he demand for miniaturized devices, higher operating frequencie s and CMOS MEMS integrable devices seems to be the motivation for using microfabrication techniques in the implementation of left handed t ransmission lines. Qin et al [32 ] successfully implemented left handed transmission lines operating at the V band by using the negative epoxy photoresist S U8 as a dielectric. Tong et al [33,34 ] successfully implemented left handed metamaterial coplanar waveguide components and circuits based on the GaAs and high resistivity silicon MMIC technology. However, the use of G aAs and high resistivity silicon substrates, combined with oxygen plasma etching during the fabrication process, increase A B
27 fabrication costs. On the other hand, p revious work ha s demonstrated that the use of thin and thick polymer dielectric materials such as SU8 BCB and Polyimide, allows the implementation of micromachined transmission lines and passive devices on CMOS grade low resistivity silicon substrates. This approach enables significant size reduction decreas es the losses associated with the low re sistivity s ilicon substrate avoid s the use of expensive high resistivity s ilicon wafers and creat es a way to implement CMOS compatible devices in the microw ave and millimeter wa ve range [3 5 40 ] Since the printed circuit board (PCB) and LTCC processes ar e well established technologies while there are still many challenge s in 3 D organic multilayer fabrication approaches the microfabrication approach based on low res i stivity silicon glass or other low cost substrates with photopatternable low curing temp erature dielectric interface layers have not been broadly explored and used for implementing advanced metamaterial applications for microwave range Meantime, the advantages of this approach includ e the low substrate cost the compact device size, the pro cess compatibility with CMOS MEMS processes, and the integrability with other components. In addition, the process can be easily scaled to the millimeter wave /THz range. 1.3. M otivation The end of the previous section suggests an opportunity to explore the imp lementation of compact metamaterial applications in the microwave and millimeter wave range. Metamaterial co ncepts are used for implementing of a broad variety of applications by using conventional PCB and LTCC processes. However, not so many applications have been demonstrated by using micromachining techniques on low cost substrates such as CMOS grade low resistivity silicon glass and the flexible liquid crystal polymer (LCP) Th e motivation of our study is to explore the use of low resistivity silicon c onventional printed circuit board flexible dielectric materials and organic substrates for achieving compact metamaterial applications that make use of advanced
28 new concepts architectures, and fabrication processes The design, modeling, simulation and fabrication process of hi ghly compact metamaterial devices is addressed in our work In addition to the conventional fabrication techniques based on the printed circu it board technology, we want to implement single layer and multilayer broadband and multib a nd metamaterial applications operating at microwave and millimeter wave frequencies by combining the original metamaterial concepts, namely the CRLH approach and the c omplementary s plit r ing r esonators (CSRR) with a surface micromachined fabrication pro cess that utilizes different dielectric materials such as the negative tone photopatternable resin Benzoclyclobutene (BCB) the negative tone photosensitive epoxy SU8 and the flexible liquid crystal polymer LCP On the other hand, we want to extend our stu dy of metamaterial concepts to devising new compact resonators for wideband filter applications that make use of reduced mode versions of the su bstrate integrated waveguide ( SIW) transmission line structure s [ 41,43 ] loaded with complementary sp lit ring r esonator (CSRR) C onventional cavity resonators, implemented with either the metallic waveguide or the substrate integrated waveguide, are useful for narrow band filter applications due to their high achievable external quality factors. However, when wide band filters are required, i.e. filters with more than 5% fractional bandwidth, these cavities do not offer the required specifications for broadband operation s such as a low external quality factor and a high internal coupling coefficient between coupled cavities To overcome th ese limitations, we propose in our work new in substrate cavity resonators that are useful for wideband filter designs. The working principle, design, simulation and implementation of s ingle, dual band and tunable resonators and bandpass filters, operating under the principle of evanescent wave amplification, are demonstrated. Moreover, by using the same multilayer surface micromachined fabrication process in combination with the in su bstrate waveguide concept micromachined evane scent
29 mode filters are presented on SU8, LCP and BCB substrates It is believed that the in substrate nature of th e resonators and filters can be extended to embedded implementations that allow conventional handling and packaging of 3D passive microstructu res without additional mechanical consideration which otherwise would require a very delicate and expensive vacuum packaging process . Finally, photopatternable SU8 epoxy and BCB have been selected as the dielectrics for implementing embedded passive devices due to their optical, electrical and mechanical properties that offer great optical transparency, low curing temperature, the capability for high aspect ratio vertical interconnection useful for multilayer implementations, the compatibility and in tegrability with CMOS/MEMS processes, and the batch processability for multiple devices. It is expected that since the conventional printed circuit board (PCB) glass, low resistivity silicon and flexible LCP substrates have been selected as the supporting materials for the fabricated CRLH devices and filters, the compatibil ity with conventional PCB implementations and CMOS integrated circuits is maintained. 1.4. Research Objectives and Contributions T he main objective of our study is the design and implementa tion of highly compact advanced metamaterial engineering applications for microwave and millimeter wave applications which can be easily scaled in frequency To achieve compact devices, t he CRLH architecture and the complementary split ring resonator load ed on in substrate waveguide structures are combined with a surface micromachined fabrication process that make s use of different organic substrates as dielectric layers. M ultilayer Grounded Coplanar Waveguide (G CPW) and Finite Ground Coplanar Waveguide (F GC) composite right/ left handed balanced transmission lines are demonstrated for the design of multiban d RF applications up to 4 0GHz. Compact p hotolithographically defined multilayer metal insulator metal ( MIM ) capacitors and polymer
30 embedded meander indu ctors are used to implement the left handed transmission line contribution. The behavior of the CRLH transmission lines is evaluated by both 3D full structure electromagnetic simulations and experimental measurements. The electrical equivalent circuits are provided and the comparison with electromagnetic simulation and measurements is discussed. Further, the use of microfabrication techniques eliminates the necessity of using surface mounting device (SMD) based lumped components and makes the CRLH structure s compatible and integrable with CMOS/MEMS processes. The optimized fabrication process is able to provide an easy way to use micromachining techniques for the implementation of miniaturized microwave and millimeter wave components containing metamaterial transmission lines and concepts. On the other hand, the study of devices that use metamaterial concepts is extended to the implementation of novel 3D integrable compact CSRR loaded reduced mod e substrate integrated waveguides structures for single, dual b and and wideband bandpass filters by using the same dielectric materials used for the CRLH implementations. The combination of metamaterial concepts with the reduced mode version of the substrate integrated waveguide [ 41 43 ], allows the implementation of r esonators with a resonance frequency below the characteristic waveguide cutoff frequency or the original cavity resonance frequency due to evanescent wave amplification [ 4 5 46 ], which offers a great size reduction since the resonator can be smaller than th e quarter wavelength at the resonance frequency Important contributions of our study include i mplementation of g rounded c oplanar w aveguide (G CPW) and f inite g round c oplanar w aveguide (FGC) balanced CRLH transmission lines on organic polymer substrates for broadband and multiband applications ; t he use of integrated and embedded passive components such as metal insulator metal (MIM) capacitors, meander line inductors and complementary split ring resonators (CSRR) for implementing multilayer metam aterial app lications up to 4 0GHz;
31 t he modeling of the CRLH transmission lines and metamaterial filters including dielectric and conductor losses, as well as the electrical equivale nt circuit extraction procedure; t he development of an optimized micromachin ing procedu re based on SU8, BCB and LCP for implementing t he proposed devices; t he introduction of new in substrate waveguide cavity metamaterial resonators for the design of wideband bandpass filters in the micr owave and millimeter wave range; t he introduction of a new CRLH architecture with all pass behavior for future projects 1.5. Dissertation Organization The dissertation is organized in six chapters as fo llows: Chapter 1 introduces the general literature review of the application of e lectromagnetic metamaterials to microwave and millimeter wave engineering. It also presents the motivation and research objectives. Chapter 2 provides the theory background on electromagnetic metamaterials t he t ransmission line approach for implementing p lanar metamaterial and the review of some pioneer work on CRLH transmission lines. Chapter 3 discusses previous work focusing on the technical challenges and the implementation of micromachined RF devices by using both conventional and CRLH transmission lines. Also the proposed general multilayer architectures for CRLH transmission lines and substrate integrated waveguide devices are presented. Chapter 4 presents single band and multiband metamaterial applications that use conventional printed circuit board and micromachining techniques for the i mplementation. A dual band three way Bagley polygon power div ider is implemented with CRLH transmission lines. Also, SU8 is used for implementing multilayer micromachined CRLH transmission lines working at microwave frequencies. At the end, a new CRLH architecture useful for all pass operation is introduced. The new structure is studied both theoretically and experimentally in order to be used as a future work in our study. Chapter 5 explores the implementation of CRLH transmission lines by using BCB as a dielectric interface layer on a low resistivity silicon substrate The design procedure
32 and the electrical simulations and measurement results for a dual band micromachined CRLH unit cell are presented. Chapter 6 introduces the CSRR loaded Half Mode Substrate Integrated Waveguide (HMSIW) resonator for the implementing single and dual band evanescent mod e coupled resonator filters. A tunable resonator, a wireless sensing application and a dual band filter that uses the evanes cent mode resonator are also presented. Chapter 7 demonstrates a 3D SU8 embedded evanescent mode half mode substrate integrated waveguide resonators and filters fabricated by using the same micromachined process. Chapter 8 presents the novel proposed CSRR loaded reduced mode substrate integrated waveguide cavities for wideband filter applications, namely the Quarter Mode and the Eighth Mode SIW. A wideband f ilter is implemented on a conventional PCB by using the CSRR loaded Eighth Mode Substrate Integrated Waveguide (EMSIW). In addition, the design and simulation of wideband bandpass filters on LCP and BCB are presented for microwave and millimeter wave appli cations. Chapter 9 concludes the dissertation with a summary of the major results and c ontributions. Figure 1 5 summarizes the organization of this document.
33 Figure 1 5 Overview of the dissertation structure Introduction Fabricati on Technology Applications Conclusions Metamaterials concepts Compact metamaterials Motivation Objectives Organization Electromagnetic Metamaterials The transmission line approach Metamaterial Applications Micromachined Transmission Lines Micromachined Metamaterial Transmission Lines Proposed metamaterial architectures. Dielectrics choice Dual band CRLH applications Multilayer CRLH Transmission Lines The Bridged CRLH concept. Printed Circuit Board Metal Insulator Metal capacitors Embedded Inductors Multiple dielectric layers Electroplated vertical interconnections Flexible substrates Evanescent mo de HMSIW filters. 3D embedded HMSIW filter. EMSIW and QMSIW micromachined filters. Summary Micromachined Bridged CRLH 3D embedded single and dual band evanescent mode filters. Future Work
34 CHAPTER 2 ELECTROMAGNETIC METAMATERIALS Electromagnetic metamaterials are artificial effective homogeneous structures, not found in nature, with unusual and interesting electromagnetic responses  Artificial complex materials with unusual properties have been an active research area since the first experiments were performed in the 19 th century [ 5 ] It is believed in 1898 Jagadis Chunder Bose con ducted the first microwave experiment with twisted structures that create d what are called today artificial chiral elements [ 47 ]. Further in 1914 Lindel et al. worked on an artificial chiral media composed of randomly oriented smal l wire h elices embedded in a host medium [4 8]. Kock in 1948 [49 ] performed experiments on compact microwave lenses that use periodically embedded metallic strips, wires and disks in order to control the refractive index of the artificial media Modern ex periments, based on new concepts, novel fabrication techniques and the periodic inclusion of novel miniaturized geometric shapes and forms in a host media, have developed new artificial electromagnetic structures and composite materials with similar proper ties to those of their known bulk counterpart, or in some cases, with new properties not readily available in nature. New concepts such as double negative (DNG) materials, chiral metamaterials, split ring resonators and complementary split ring resonators, omega media, bianisotropic media, among others, have been the subject of research of many leading groups around the world. In a bulk composite host medium with inclusions, the e lectromagnetic waves induce electric and magnetic moments, which affect the transmission capabilities of the material and its constitutive parameters such as permeability and permittivity [5 ] The artificial medium shou l d be homogeneous in order to be characterized by the electrical permittivity ( ) and the magnetic permeability ( ). In order to ensure that the medium is electromagnetically uniform along the direction of propagation of an electromagnetic wave, and thus, allowing refractive phenomena to
35 dominate over scattering/diffraction phenomena, the structural average cell siz e p which is the size of the unit cell containing one inclusion, should be much smaller than a quarter of the incident guided wavelength g p < g /4 [6 ] This condition is called as the effective homogeneity limit or the effective homogeneity condition and defines the medium as an effective homogeneous medium that behaves as a real material, in which the electromagnetic waves are essentially unaware of the lattice structure and the macroscopic parameters of the medium depend on the nature of the unit cel l This homogeneity relation is a rule of thumb effectiveness condition. The relation is often used in microwave engineering to distinguish lumped components ( p < g /4) from quasi lumped components, ( g /4 < p < g /2 ) and distributed components (p > g /2 ). 2.1. Theoretical P rediction of Metamaterials The first theoretical work speculating the existence of substances with negative values of permittivity, and permeability, ( left handed m aterials), was introduced by Veselago [ 1 ] in 1967. In his paper, Vesela go predicted a medium which would allow the propagation of electromagnetic waves with the electric fi eld, E the magnetic fi eld, H, and the wave vector, k building a left handed orthogonal set or triad, in contrast with the well known right handed triad i n conventional materials Left handed materials (LHMs), also known as negative index materials (NIMs), or double negative materials (DNG), have simultaneous negative permittivity negative permeability, and negative refractive index, n over a specif ic frequency range This new kind of speculated material would show interesting new electromagnetic properties not readily available in conventional naturally occurring materials, which are currently used for the creation of new applications in physics and engineering. S ome of those properties include
36 f requency dispersion of the constitutive parameters and which means the propagation constant is a nonlinear function of frequency; r eversal refraction ( can be still app lied ) ; a nti paral lel relation of the group and phase velocity; s ubw avelength focusing and imaging. In order to investigate the electromagnetic properties of left handed materials, Veselago first studied how the electromagnetic waves propagate in a medium with both negative pe rmittivity, and permeability, For wave propagation problems, t he source free Maxwell equations are given by ( 2 1 ) ( 2 2 ) ( 2 3 ) and ( 2 4 ) w here E (V/m) is the electric field intensity, H (A/m) is the magnetic field intensity, B (W/m 2 ) is the magnetic flux density and D (C/m 2 ) is the electric field density. In a linear ( and not depending on E or H ) and non d ispersive ( and not depending on frequency ), such as simple homogeneous isotropic dielectric and magnetic materials, the cons titutive relations are as in D= E ( 2 5 ) and B= H ( 2 6 )
37 Now in order to get valuable info rmation on the fundamental response of the medium, a planar wave is considered [11 ] then the electric and magnetic field are given by ( 2 7 ) and ( 2 8 ) w her e = E / H represents the wave impedance. Now replacing previous Equations 2 7 and 2 8 in the Maxwell equations as in E=+ H ( 2 9 ) and H= E ( 2 10 ) the conventional solution for right handed materials is obta ined where the vectors E H and build the well known right handed triad, as shown in Figure 1 1 A. If a left handed medium is considered ( < 0, < 0) then E= H ( 2 1 1 ) H= + E ( 2 1 2 ) w hich creates the left handed triad as previously shown in Figure 1 1 B. It is known that the propagation constant is positive for a right handed medium (outward propagation from the source), which is not true for a left handed medium, in which, according to the previous equations and the previously shown in Figure 1 1 B the propagation constant is negative (inward propagation to the source) and hence, the phase velocity v p is opposite to the phase ve locity o f a right handed medium as in RH medium : >0, v p > 0 ( 2 1 3 ) and L H medium : <0, v p < 0 ( 2 1 4 )
38 The direction of the energy flow is given by the Poynting vector as in S=E H (2 1 5 ) The non zero Poynting vector always forms a right handed coordinate system with E and H independent on the signs of and Therefore, in a left handed material, the wave vector, is in the opposite d irection of the energy density f l ow, S Such a wave is called backward wave I n contrast, in a normal right handed material (RHM), the wave vector, and the energy flow S are in the same direction and the wave is a forward wave 2.2. Negative R efraction The reflect ion and refraction of an incident plane wave on the boundary between two homogeneous media of different dielectric properties are well known classical problem s in electromagnetism As illustrated in Figure 2 1 an incident wave at the interface of two different medi a wi ll gener ate a reflected wave and a refracted (transmitted) The refractive angle is determined by the Snell's Law of refraction : ( 2 1 6 ) In a left handed medium, since the constitutive parameters and are negative, a negative refractive in dex, n < 0, is obtained: n = n (2 1 7 ) When considering an incident wave at the interface between a right handed (RH) medium and a left handed (LH) medium a n egative sign in the refractive index of the LH medium appears which means a negati ve refraction angle is obtained as illustrated in Figure 2 1 A more general form (2 18)
39 w here s is the sign of the refractive index: negative for a LH medium and positive for a RH medium. If the two medi a are left of the two minus signs of the refractive ind ices which means that an incident wave on the boundary between two media with sam e handedness properties has the conventional positive refraction with positive refraction angle. Figure 2 1 Reflection and refraction of EM waves in the RH and LH medi a ( Adaptation of Figure 3 from The electrodynamics of substances with simultaneously negative values of and Sov. Phys. Usp.,vol. 10, Copyright 1968 with permission from IOPSCIENCE ) It is observed in Figure 2 1 tha t the wave vector of the refractive wave in the LH medium heads toward the interface, indicating that the refractive wave is a backward wave traveling towards the source. On the other hand, the Poy nting vecto r, which represents the energy, head s away from the interface which means that the group velocity v g is antiparallel with the phase velocity, v p as it was stated for the propagating waves in the left handed medium. i i t t Incident wave Reflected wave Transmitted wave in LH medium Transmitted wave in RH medium LH medium RH medium
40 2.3. Phase V eloc ity and Group Velocity It was previously mentioned that the phase velocity v p and the group velocity v g are in the opposite direction in left handed materials. The phase velocity is defined by ( 2 1 9 ) w here is the unit vector along the propagation direction Since the frequency is always a positive quantity, the relations in E quations 2 1 3 and 2 14 are valid for the RH and LH medium respectively. T he group velocity, v g which is related t o the Poynting vector, is given by ( 2 20 ) Since the Poynting vector S depends only on the electric field vector E and magnetic field vector H but not on the constitutive parameters and as the propagation constant does it is oriented toward the direction of the energy over time and hence, is parallel to the group velocity Then it is shown that the phase and group velocities are anti parallel in the LH medi um As a summary, the following relations are valid : RH medium : >0, v p > 0, v g > 0 (2 21 ) and LH medium : <0, v p < 0, v g > 0 ( 2 22 ) 2.4. Experimental Demonstration of M etamaterials Although Veselago theoretically proposed the existence of left handed materials in 1967, it was only recently a fter more than 30 years that such materials were implemented and d emonstrated experimentally The first experimental demonstration of left handedness was based on an artificial effectively homogeneous material, instead of a natural ly occurring substance, a s expected by Veselago. The pioneering work of Pendry and his colleagues at the Imperial College,
41 London UK  on microwave plasmonic structures showing negative /positive (metal thin wires) and positive /negative (split ring resonators) w as the f oundation for implementing artificial left handed materials. Both of t hese plasmonic structures are considered effectively homogenous structures since they feature a unit cell size p much smaller than the guided wavelength ( p << g ). Figure 2 2 Negative permittivity medium A) Th e metallic thin wire (TW) array. B) The real part of the permittivity, as a function of the frequency ratio / pe (Adaptation of Figure 2 from J. B. Pendry, A. J. Hol den, D. J. Robbins and W. J. Stewart, IEEE Trans. Microw. Theory Tech. vol. 47, no. 11, pp. 2075 2084, Copyright 1999 with permission from IEEE) 2.4.1. Negative P ermittivity M edium The metal thin w ire (TW) periodic array shown in Figure 2 2 is known as a negative /positive plasmonic structure  Thi s structure can be designed to have its plasma frequency in the microwave range. Although negative permittivity materials are found in nature such as gas and metal plasmas, their plasma frequency is far above the microwave range and their A B p p z
42 negative permittivity values on the order of 1 can be achieved at microwave frequencies. When the wires are excited with an electric field E parallel to their axis, a current is induced along the wires and equivalent electrical dipoles moments are generated. The permittivity of the excited metal wires is a fun ction of the frequency given by ( 2 23 ) w here pe is the electric plasma frequency, and is the damp ing factor due to metal losses. These parameters are defined in terms of the geometry of the periodic wire a rray as in ( 2 24 ) and ( 2 25 ) w here c is the speed of the light, a is the radius of the wires, p is the lattice constant or separation between wires, and is the conductivity of the me tal. The negative permittivity is achieved if Re( r )<0, for ( 2 26 ) which is reduced to r < 0, for < pe ( 2 27 ) w hen losses are not considered ( =0). Then the array shows negative permittivity below the plas ma frequency. 2.4.2. Negative P ermeability M edium In order to achieve negative permittivity over a frequency range, Pendry proposed the positive /negative metal split ring resonator (SRR) d esign [3 ], illustrated in Figure 2 3 When a n incident magnetic field H is perpen dicular to the plane of the rings, resonating currents will be induced on both the inner and outer rings, charges will accumulate between the gaps of both
43 rings, and equivalent magnetic dipole moments are created. Due t o the artificial magnetic dipoles, the SRR has a magnetic response despite not being constructed with any magnetic material. Figure 2 3 Negative permeability medium A ) Split ring resonator (SRRs ) structure with lattice size, p B ) Single SRR configuration. (Adapt ation of Figures 12(a) and 13 from J. B. Pend Magnetism from conductors IEEE Trans. Microw. Theory Te ch. vol. 47, no. 11, pp. 2075 2084, Copyright 1999 with permission from IEEE ). The split ring resonator exhibits a frequency depend e nt permeability, given by [3 ] ( 2 28 ) The factor F is given by ( 2 29 ) w here a is the inner radius of the smaller ring and p is the lattice constant or the center to center distance between the periodically arranged split rings resonators. The magnetic resonance frequency, 0m is given by ( 2 3 0 ) w here is the width of the rings, is the space between the inner and outer rings, and c is the speed of the light. The damping factor due to metal losses, is given by (a) y p A B a
44 ( 2 31 ) w al resistance per unit length. Figure 2 4 Variation of the permeability of the SRR with the frequency. The real part in blue, the imaginary part in red. The magnetic resonance frequency is 0m From E quation 2 28 and as a general case when losses are not considered ( =0), it is observed that negative permeability, Re( r ) < 0, ex ists within the frequency range defined by ( 2 32 ) w ith pm as the magnetic plasm a frequen cy. Due to the resonant behavior of the split ring res onator, the nature of the expression for the permeability is resonant ( = 0m = ), which means that the negative permeability is provided within a narrow band of frequencies. Figure 2 4 illustrates the frequency depend e nt permeability of the SRR. In the SRR, each individual ring behaves as a series LRC circuit with a resonant frequency where the inductance of t he ring is modeled by L C represents the capacitance due to the gap and R models the conductor losses  Figure 2 5 shows the
45 electrical equivalent circuit for the double ring configuration, where the capacitance C m models the capacitive coupling between the rings and the transformer with ratio n models the magnetic coupling. If the mutual coupling between rings is weak, then the circuit parameters of each ring are very close in value because the dimensions of both rings are si milar, then L 1 L 2 L, C 1 C 2 C, which gives a combined resonance frequency close to that of a single SRR with the same dimensions. However, due to the higher current density in the double ring SRR, the magnetic moment is larger  Figure 2 5 Split ring resonator A ) D ouble ring split ring resonator. B ) Electrical Equivalent Circuit. 2.4.3. Left H anded Material Demonstration In naturally occurring bulk materials, the permittivity, and permeability, are used to present a homogeneous view of the electromagnetic properties, such as polarization and magnetization. The macroscopic electr ic and magnetic prop erties of a medium ca n be considered as the average behaviors of the electrons and atoms when inte racting with the electric and magnetic fi elds of electromagnetic waves. Therefore, conventional materials are considered composites, where the atoms and molecules are the i ndividual unit cells with sizes much smaller than the wavelength of the EM waves in the medium. In this sense, artificial composite periodic structures, defi ned by a unit cell with dimension a can be considered as an electromagnetic L 1 L 2 R 1 n C m C 1 C 2 R 2 A B
46 homogenous medium if the size of the unit cell is much smaller than the wavelength of the EM waves ( a << ) Then, t he macroscopic electromagnetic properties of the unit cell are considered as the effective EM properties of the medium, allowing refraction phenomena to dominate over scattering and diffraction when EM waves propagate inside the medium. Followi ng the previous reasoning, in [2 ] Smith et al. create d a composite structure by combining the thin metal wire (TW) and the metal split ring resonator (SRR) structures develope d by Pendry et al.  with the expectation to demonstrate the first artificial l eft handed (LH) material. In order to reduce the coupling interactions between the two structures and to cancel mutual interference, the wires were located in the symmetry center of the rings which creates currents with opposite signs in the SRRs and the TWs [ 2 ]. Then, since there is no mutual interference, it is supposed the composite structure preserve s both the negative permittivity from the thin metal wires and the negative permeability from the split ring resonators. The TWs and SRRs are designed wit h overlapping frequency ranges of negative permittivity and permeability respectively. The experiment is performed by exciting the three structure s, the SRRs array, the TWs array, and the composite structure with an electromagnetic wave e j r in a wavegui de environment As shown in Figure 2 4 a frequency band with negative permeability is obtained for the SRRs structure, which causes a stopband when excited due to the magnetic resonance  When the combined structure is excited the previously observed stopband disappears due to the insertion of the thin wires with negative permittivity. Instead, a passband is observed in the frequency range of interest, which proves that both the permittivity and permeability of the combined st ructure are negative, therefore a propagation constant with a real value is observed A similar experiment was later demonstrated by Osbay et al.  in a free space environment, in
47 which intercalated planar arrays of SRRs and TWs were used to implement a resonant bulk LH material, as previously illustrated in Figure 1 2 and recalled here i n Figure 2 6 Figure 2 6 L eft handed materials in free sp ace. (Recalled of Figure 1 2 ). A ) The left handed struct ure. B) The transmission and reflection response s. (Adaptation and reprint of Figures 1 and 4 from E. Ozbay, K. Aydin, E. Cubukcu and M. Bayindir, M, Transmission and reflec tion properties of composite double negative metamaterials in free space, IEEE Trans. Antennas and Propagat vol .51, no.10, pp. 2592 2595, Copyright 2003 with permission from IEEE). The two experiment s were the beginning of the demonstration of left h anded materials and led to a large number of both theoretical and experimental work confirming the existence and electromagnetic properties of the left handed materials. In [ 50 ], Shelby et al. implemented a two dimensional bulk TW SRR structure with the sh ape of a prism. An electromagnetic wave is launched into the structure and the transmission coefficient of the wave refracted by the structure is measured at different angles. The left handed behavior of the structure is confirmed by observing that the ma ximum of the transmission coefficient of the refracted wave is obtained in to conventional materials). When the structure is replaced by a conventional material ( Teflon) with the same shape, the maximum of the transmission coefficient is obtained in the positive angle A B x y z
48 with respect to the interface of the prism. The measured refractive angle was consistent with the n is negati ve. Further theoretical and experimental work w as per formed by different groups [14 51 56 ], which confirmed the left handed properties of the composite metamaterial designs and proposed practical engineering applications. 2.5. Transmission Line Approach of Met amaterials The first experimental demonstration of the composite left handed (LH) material constituted by the combination of thin wires (TW) and split ring resonators (SRR), opened a new exciting research area in physics and engineering, namely the metamat erials (MTMs). Although the TW SRR structure shows the unusual properties described by the theory of left handed materials, it is not practical for engineering applications due to its resonant behavior, narrow bandwidth and high loss. A resonant structure is not a good propagation medium for a modulated signal because the quality factor of the resonator will affect the propagation of the signals, so these cannot be transmitted without distortion. In contrast, nonresonant medi a can offer a broad bandwidth an d low losses, so the modulated signal can be transmitted efficiently. In order to offer practical implementations of metamaterials for engineering applications, the nonresonant transmission line approach for the design of planar metamaterials was proposed, almost simultaneously, by three different groups [ 7 10 ] in 2002, leading to the conception of the generalized planar negative refractive index transmission lines or the composite right / left handed (CRLH) transmission lines. In contrast with the resonant a pproach, planar metamaterial transmission lines exhibit broad bandwidth, low loss and can be easily integrated with other microwave components and systems. The CRLH transmission lines (TL) are a mix of both right handed (RH) and left handed (LH), which cre ates unique properties widely used for practical engineering applications.
49 2.5.1. The Left handed Transmission Lines The incremental circuit model of a uniform right handed transmission line consists of a series inductor and a shun t capacitor, as shown in Figure 2 7 [ 57 ]. The transmission line theory is derived from this model, which offers the well known telegrapher equations that define the most important transmission characteristics of the transmission line, such as the characteristic impedance, propagation constant, phase and group velocities [5 7 ]. Figure 2 7 Modeling of a uniform right handed transmission line A) Transmission line section of size p B) Incremental ci rcuit mod el Figure 2 8 Modeling of a uniform left handed transmission line A) Transmission line section of size p B) Incremental circuit model. (Adaptation of Figure 1(b) from Lai, A.; Itoh, T.; Caloz, C. ; "Composite right/left handed transmission line metamaterials," Microwave Mag IEEE vol.5, no.3, pp. 34 50, Copyright 2004, with permission from IEEE ). A B Z 0 z = p Uniform L H TL z = p L (F.m) L L (H .m) R (F.m) L R (H .m ) Z c z = p Uniform RH TL z = p A B
50 A left handed (LH) transmission line is considered as the dual of a right h and ed (RH) transmissi on line, with its incremental model consisting of a series capacitance per unit length (C L ) and a shunt inductance per unit length (L L ), as shown in Figure 2 8 [6 ]. By using the telegrapher equations [11 ], and considering the lo ssless case, the transmission characteristics of this kind of transmission lines can be defined as in ( 2 33 ) ( 2 3 4 ) ( 2 35 ) ( 2 36 ) an d ( 2 37 ) w here is the propagation constant, Z c is the characteristic impedance, v p and v g are the phase and group velocities respectively. It is found that the propagation constant is negative, which means backward wav es can be propagate d in the structu re. Also, Equations 2 36 and 2 37 show that the phase and group velocities are antiparalell. In conclusion, this structure shows the previously described left handed behavior, and hence can be used as the constitutive uni t cell for engineering periodic structures that behave as a left handed transmission medium if the effective homogeneous medium condition that the size p of the unit cell would be much smaller than the guided wavelength g ( p<< g ) is accomplished
51 2.5.2. The Com posite Right/ Left handed Transmission Lines A planar microstrip implementation of a left handed transmission lin e is shown in Figure 2 9 A (recalled from Figure 1 4 ) [6 ]. Figure 2 9 B shows the unit cell. T he series capacitance and shunt inductance of the unit cell are implemented with the interdigital capacitor and the shorted stub inductor respectively. Although this implementation can show a left handed behavior, it also ha s r ight handed contribution due to the parasitic series inductance provided by the fingers of the interdigital capacitors and the parasitic trace to ground capacitance of the shorted stub inductor Then, a practical realization of a left handed tran smission l ine is in reality a combination of right and left handed contributions, which constitutes the composite right/ left handed (CRLH) transmissi on line, as introduced in [6 ]. Figure 2 9 Planar compos ite right/left handed transmission line A) Microstrip implementation of a 5 unit cells CRLH Transmission line (Adaptation of Figure 8 from A. Lai, T. Itoh and C. Caloz, Composite right/left handed transmission line metamaterials, IEEE Microwave Mag vo l.5, no.3, pp. 34 50, Copyright 2004, with permission from IEEE). B) Unit cell (Reprint of Figure 5 from C. Caloz and T. handed (LH) structures and microstrip realization of a low IEEE T rans. Antennas Propagat ., vol. 52, no. 5, Copyright 2004, with permission from IEEE ). The equivalent incremental circuit model of a CRLH trans mission line is shown in Figure 2 10 A. At lower frequencies ( 0), the model shows a le ft handed behavior, since only the left A B
52 handed contributions L and L are taken into account because R and R behave as short and open circuits respectively, as illustrated in the dispersion diagram in Figure 2 10 B. At highe r frequencies ( ), the model shows a right handed behavior because L and L tend to be open and short circuit respectively, as illustrated in the dispersion diagram in Figure 2 10 B It is observed that the CRLH unit cell pr esents a non linear dispersion diagram, in contrast to the linear phase presented by con ventional RH transmission lines. The non linear dispersion makes the CRLH approach useful for implementing planar multiband applications [16, 21] Figure 2 10 CRLH transmission line. A) Incremental unit cell model. B ) Dispersion diagram showing the three different kind of lines: Pure RH (green), pure LH (blue) and balanced and unbalanced cases for CRLH line (red) ( Adaptation of Figure 4 from Caloz, C., Itoh, T. "Metamaterials for High Frequency Electronics," Proceedings of the IEEE v ol.93, no.10, pp.1744 1752, Copyright 2005 with permission from IEEE). T he transmission characteristi cs of the CRLH TL are obt ained by using a similar analysis to that used for RH transmission lines The series impedance an d shunt admittance are given by (2 38 ) and ( 2 39 ) A B z = p L L L L R R PRH Line PLH Line CRLH Unbalanced Bandgap LH high pass band w 0 w se w sh p
53 Then, the complex propagation constant is given by ( 2 40 ) The series and shunt resonances are given by and (2 41) Then two frequency ranges for the L H and RH behaviors are observed. Due to the different valu es of the series and shunt resonances, a gap exists between the LH and RH ranges, where the group velocity is zero ( v g =0) and the maximum attenuation occurs. It is found that the resonance frequency of the CRLH unit cell, which is the frequency of m aximum attenuation, is given by ( 2 42 ) In addition, the characteristic impedance is given by: ( 2 43 ) w here Z L and Z R are the characteristic impedances of the left and right handed contributions respectively. The case when the series and shunt resonance are equal ( se = sh ) is called balanced where there is no gap and the three resonance frequency, 0 se and sh are the same. Therefore, the propagation constant and the characteristic impedance f or the balanced case are given by ( 2 44 ) and ( 2 45 )
54 It is observed that the characteristic impedance is frequency independent, which means broadband matching is possible. On th e other hand, it is observed in the diagram for the balanced case that the propagation constant, is zero at the transition frequency 0 which means that an infinite wavelength regime with non zero group velocity is achieved at a non zero frequency and the phase of the propagating wave is uniform along the CRLH transmission line. This property is very unique and useful on the design of zeroth order resonators. Finally, the effective constitutive para meters can be obtained as in ( 2 46 ) and ( 2 47 ) while the refraction index is given by which shows negative values for frequencies below the transition frequency o  2.5.3. P eriodic Structure Implementation It was demon strated that the incremental equivalent circuit of the CRLH transmission line has both right handed and left handed properties. As seen in Figure 2 9 a generalized implementation of CRLH metamaterial transmission lines for microw ave applications is realized by periodically arranged series capacitors and shorted shunt inductors. A variety of conventional substrates and planar technologies can be used for practical e ngineering implementations As previously discussed, if the size, p of the unit cell is much smaller than the guided wavelength ( p << g ) then the periodic structure is considered an effective homogeneous medium. Periodic structures can be described by using the propagation constant ( = + j ) and the Bloch impedance ( Z B ) which are obtained by applying periodic boundary condition s to the unit cell represented by its ABCD matrix [5 7 ] Practical CRLH transmission lines are realized by
55 using real LC components ( C L L L ,C R ,L R ) that implement a determined amount, N of un it cell s with small electrical length ( < /2) or small size ( p < g /4). Then the idealized CRLH transmission line can be considered as a practical transmission line if the following relations are applied: ( 2 48 ) ( 2 49 ) ( 2 50 ) After applying periodic boundary conditions based on the ABCD matrix, the propagation constant, and the Bloch impedance, Z B are obtained [11 ]: ( 2 51 ) or from the ABCD parameters ( 2 52 ) Moreover, since the transmission line is a microwave network, t he S parameters can be used for characterizing the unit cell. As shown in the dispersion diagram, a CRLH transmission line has a passband behavior where the Bloch impedance is real and the structure ca n support traveling waves [11 ]. The lower and hig her cutoff frequencies a re given by : ( 2 53 ) and ( 2 54 )
56 w ith the variables w R and w L introduced for convenience as in ( 2 55 ) and ( 2 56 ) In addition, since the Bloch impedance is frequency depend e nt and not constant in the passband, CRLH transmission line based designs do not offer optimum solutions in terms of the passband insertion loss. However, the unique properties they offer make the m attractive for a great variety of microwave engineering applications. On the other hand, because of the non linear behavior of the dispersion diagram ( Figure 2 10 B ), the group velocity, a nd the phase velocity, of a propagating wave along the structure can be parallel (RH range ) or antiparallel (LH range). During the LH range ( cL < < 0 and < 0) the group velocity is positive while the phase velocity is negat ive, which means that a backward wave is propagated where the energy flows from source to load and the phase a dvances toward the source [11 ]. During the RH range ( 0 < < cR and > 0) both the group and phase velocities are positive, which means tha t a conventional forward wave propagation is obtained where the energy flows from source to load and the phase advances toward the load [11 ]. A special and interesting case is presented when the frequency of the propagating signal is equal to the transitio n frequency ( = 0 ). At this point, the propagation constant is zero ( = 0), which means that the phase velocity is infinite ( v p = ) and the group velocity is positive. Therefore, an infinite wavelength propagation regime is presented where the energy flo ws toward the load but the phase is constant along the CRLH transmission line (as a DC behavior).
57 2.6. Review of CRLH Transmission Line Applications After the introduction of the transmission line approach for implementing planar metamaterials, many microwave and millimeter wave guiding and radiating applications have been demonstrated. The unusual properties of the proposed CRLH transmission lines such as non linear dispersion, bandpass behavior, infinite wavelength regime, backward wave propagation and l eaky wave radiation, are exploited for implementing creative new applications In this session, s ome examples of the CRLH transmission lines are discussed and their operating principle is explained. 2.6.1. Metamaterial Couplers and Filters The c onventional broadband microstrip directional couplers have weak coupling levels of 10 dB or less [ 57 ] Tight coupling levels around 3 dB are normally achieved over a narrow bandwidth (<10%) by using non coupled line couplers such as the branch line and ring couplers [ 57 ]. However, if tight coupling over a broad band width is desired, the use of complicated coupler configurations, such as the Lange coupler, is necessary. As an alternative, PCB based symmetric (two CRLH TLs) and asymmetric (only one line is CRLH TL) broadband (50%) arbitrary coupling level (e ven 0 dB) d irectional couplers [12,13 ] were demonstrated by using CRLH transmission lines. A rbitrary t igh t coupling achieved in the CRLH TL couplers is the result of imaginary even and odd mode impedances within the coupling range which offers a broad matching frequ ency range. On the other hand, the broadband passband behavior of CRLH transmission lines make them useful for the implementation of low loss broadband filter applications [1 4, 15 ]. 2.6.2. Multiband C omponents The unique non linear dispersion characteristic of t he CRLH transmission lines makes them useful for the design of dual band components, since the specific values of phase shifts can
58 be obtained at arbitrary frequencies [1 6 ]. Moreover, the combination of conventional CRLH transmission lines with the dual CR LH transmission line concept [ 17 ] is used for the design of triple band and quad band miniaturized applications [18, 19 ]. Branch line and rat race ring couplers [16 ] quadrature mixers [ 20 ] Bagley polygon power divider [21 ] (part of our study ) and dual ba nd antennas [ 22 ] were demonstrated by using dispersion engineering of the CRLH and dual CRLH transmission lines. Figu re 2 11 shows one of the dual band applications of the CRLH transmission lines. Figu re 2 11 CRLH dual band components A ) Dual band CRLH branch line coupler B) Frequency response of the dual band coupler. ( Reprint of Figures 10 and 11 from I.H. Lin, M. De Vincentis, C. Caloz and T. Itoh. band componentsusing composite right/left IEEE Trans. Microw. Theory Tech ., vol. 52, no. 4, pp. 1142 1149, Copyright 2004, with permission from IEEE ). 2.6.3. Metamaterial Resonators The CRLH transmission lines can also be used as short or open ended resonators if their length is a multiple of h alf of the guided wavelength [23 ]. In addition to the conventional positive order resonances (in the RH band of the resonator), CRLH resonators exhibit negative order resonances i n the LH band, and a n unusual zero order resonance at the transition frequency A B
59 o Moreover, since the propagation constant = 0 at o CRLH TL zero th order resonators are independent of the physical length, which allows them to be arbitrarily small. This kind of resonators ha s been used for the implementation of small zeroth order resonant anten nas and tunable applications [24 25 ]. 2.6.4. Compact M ultilayer C omponents Although the CRLH approach is a good way to implement planar metamaterials, some of the demonstrated applications are implemented by using surface mount technology (SMT) discrete chip components (capacitors and inductor s), which are available only in discrete values and have a limited frequency operation range due to the natural self resonance frequency associated with the chips. On the other hand, chip components cannot be easily integrated in flexible, multilayer or 3D structures that may be used for miniaturization purposes or reconfigurable radiated wave applications. The above considerations, which represent the need for additional size reduction operation at higher frequencies and 3D integration, extended the tran smission line approach to the implementation of super compact multilayer vertical CRLH transmission lines on conventional printed circuit board processes and ceramic LTCC process es combined with MIM capacitors and meander lines [26 27 ] This multilayer app roach offer s an alternative to the conventional implementations with printed planar passive components (interdigital capacitors, stub inductors) and allows 3D structure implementation. T he LTCC technology is an example of 3D approaches for miniaturized CR LH devices for the microwave range. Since the CRLH concept does not depend on the fabrication technology used, the multilayer vertical CRLH transmission line has all the unusual properties previously described and thus, can be used for the implementation o f new metamaterial applications
60 2.6.5. CRLH Substrate I ntegrated W aveguide C omponents Recently, the substrate integrated waveguide (SIW) [ 41 ] and the half mode substrate integrated waveguide (HMSIW) [42 ] technologies, have been integrated with interdigital capa citors in order to achieve SIW and HMSIW CRLH transmission lines that can be used for implementing microwave applications , with all the well known advantages offer ed by the SIW and HMSIW, such as high Q factor and low radiation losses. Broadband and d ual band couplers [28,29], leaky wave antennas  and negative order resonator antennas and filters  ( Figure 2 12 ) have been demonstrated by using the concept, which places the recently introduced SIW and H MSIW technologies as good architectures for the implement ing CRLH devices for microwave and millimeter wave applications. Figure 2 12 CRLH SIW slot antennas A ) One and two stages CRLH SIW antennas B ) Frequency response of the two stages open ended antenna ( Reprint of Figures 10 and 11 from Y Dong, T. Itoh, Miniaturized Substrate Integrated Waveguide Slot Antennas Based on Negative Order Resonance, IEEE Trans. Antennas and Propagat vol.58, n o.12, pp.3856 3864, Copyright 2010 with permission from IEEE ) 2.6.6. Micromachined CRLH A pplications As the frequency increases, the demand for miniaturized devices is stronger. This seems to be the motivation for using microfabrication techniques CMOS and MMI C process es in the implementation of left handed t ransmission lines. Qin et al [32 ] successfully demonstrated left A B
61 handed transmission lines operating at the V band by using the negative epoxy photoresist SU8 as a dielectric. Also, CRLH transmission lines have been implemented in MMIC Ga As and Si processes [33, 34 ]. Previous work ha s demonstrated that the use of thin and thick polymer dielectric materials wit low loss such as SU8, BCB and Polyimide, allows the implementation of micromachined transmission l ines and passive devices on a CMOS grade low resistivity silicon substrate The dielectric on top of the silicon serves as a passivation layer that decreases the losses due to the silicon and avoids the use of the expensive high resistivity s ilicon wafers offering at the same time a way to implement CMOS compatible devices in the microwave and millimeter wa ve range [35 40 ]. However, due to the availability of low cost printed circuit board (PCB) and LTCC processes, and the technical challenges in multilaye r fabrication, organic based microfabrication processes have not been broadly used for implementing multilayer metamaterial devices in the microwave range and millimeter wave range
62 CHAPTER 3 MICROMACHINED TRANSMISSION LINES During the past decade a d vanced 3D integration microelectromechanical systems (MEMS) fabrication technolog ies ha ve been used to build RF components and devices with similar or superior performance to that of conventional counterparts, while simultaneously achieving more compact an d less expensive products. MEMS fabrication techniques for 3D structures, such as surface micromachining, bulk micromachining, LIGA processes, wafer bonding, polymer based processes and lamination, have revolutionized the development of RF and wireless com munication systems. The new applications and fabrication technologies have created what is commonly known today as RF MEMS, which include s not only RF devices with movable microelectromechanical par ts, but also passive guiding and radiating wave RF compon ents fabricated by using MEMS technolog ies in order to reduce size, enhance performance parameters and achieve 3D integrability These microfabrication technologies have been beneficial for the development of new devices for micr owave and millimeter wave a pplications such as RF micro switc hes [5 8 ] miniaturized micromachined filters [59 ] and antennas [60,61 ] miniaturized embedded high Q passive components [4 4 ] and micromachined transmission lines [35 40]. It is well known that a t high frequencies the sig nal to be transmitted is considered to be an electromagnetic wave, wh ose wavelength is normally comparable with the size of the passive and active components in the system. That is the reason to use different kind s of distributed wave guiding structures, s uch as transmission lines, which can effectively propagate the signal with lo w loss and distortion. The c haracteristic impedance of the transmission line the propagation constant and the wave propagation mode are important parameters taken into account du ring the design process. According to the wave polarization properties, different modes of propagation
63 are possible in a wave guiding structure, such as Transversal Electro Magnetic (TEM), Transversal Electric (TE) and Transversal Magnatic (TM). Most pract ical applications work with a s ingle mode of propagation which is defined by the geometry of the structure and the wave polarization properties. Figure 3 1 shows some of the planar transmission lines that are wide ly used for microwave and millimeter wave applications. Figure 3 1 Cross section of some transmission lines used in microwa ve and millimeter wave circuits. A) Micro strip. B) Coplanar Waveguide ( CPW). C) Grounded Coplanar Waveguide ( G CPW). D) Finite Ground CPW (FGC). E) Coplanar Strip Line (CPS). B) Substrate Integrated Waveguide (SIW). One of the most popular planar guiding structures is the microstrip line in Figure 3 1 A which has a metallic signal line on top of a ground plane The fundamental propagation mode in the micros trip line is known as quasi TEM, because of its similar ity to the pure TEM mode. However, because of the needs for vias to the gro und, some dispersion in the parameters of the line (frequency dependence) high radiation loss and cross talk between adjacent lines, the microstrip line is often rejected to be used in some microwave and millimeter wave applications. As alternatives, the coplanar waveguide (CPW) in Figure 3 1 B and its variants such as the S H Ground plane Substrate W S Ground W S W g W g W S S S Ground Vias A B C D E F W S Ground Ground
64 Grounded Coplan ar Waveguide (GCPW) in Figure 3 1 C and the Finite Ground Copla nar Waveguide (FGC) in Figure 3 1 D are used in a variety of applications The uniplanar implementation of the CPW allows only CPW modes to be transmitted when all the ground terminal s are kept at the same potential. However, in the majori ty of the applications air bridges are necessary to connect the ground terminals and avoid slotline modes. One of the most interesting variants of the CPW is the Finite Coplanar Waveguide (FGC) [6 2 ], which has finite size narrow ground planes in order to a void parallel plate modes and spurious resonances normally presented in the CPW or GCPW configurations, while at the same time reducing mutual coupling between adjacent lines and radiation losses. Another popular waveguiding structure is the Coplanar Stri p s Line (CPS), shown in Figure 3 1 E which has a balanced structure that is useful for RFIC applications with reduced cross talk, high common rejection ratio and high quality signal integrity . Recently, the Su bstrate Integrated Waveguide (SIW) [ 41 ], shown in Figure 3 1 F has been widely used for microwave and millimeter wave applications The SIW and their different variants, such as the Half Mode SIW (HMSIW), offer a planar in substrate implementation of a rectangular waveguide like transmission line. A row of metalized vias at each side connecting top and bottom conductors behave as a metallic wall, and so a waveguide mode is propagated. This in substrate implementati on offer s all the advantages of a conventional cavity waveguide, such as high Q factor, low radiation losses and low cross talk between adjacent lines, but in a planar fashion that is compatible with conventional microstrip and CPW implementations. However the necessity of an array of via holes makes its implementation more complex than that of CPW or Microstrip.
65 It is observed that the availability of MEMS micromachining techniques has made it possible to fabricate miniaturized transmission lines for mic rowave, millimeter wave and terahertz applications. It is the purpose of this chapter to present a literature review of the most important and interesting works on micromachined conventional and left handed transmission lines 3.1. Bulk Micromachined Transmis sion Lines In selected microwave and millimeter wave applications, d ifferent mic romachining techniques are used for fabricating transmission lines with reduced dielectric and radiation losses in part due to the use of low resistivity substrates This is t he reason why micromachining techniques have been used more on both conventional low resistivity and high resistivity silicon than on GaAs substrates. By removing part of the lossy dielectric substrate, where the electromagnetic field is confined, the line is basically implemented on air dielectric and thus the dielectric losses and the dispersive characteristics of the parameters of the line are significa n tly reduced. In fact, having air rather than the lossy substrate as the dielectric, the attenuation of the CPW can be reduced considerably On the other hand, the use of microfabricated shields can reduce the radiation losses. Bulk micromachining is the selected technique to remove the substrate between metalized signal lines or ground planes. One of the p ioneer works is presented in [63 ] by Herrick et al ., in which trenched finite ground coplanar waveguide (FGC) lines are implemented with a removed substrate between terminal s by using ethylenediamene pyrocatechol (EDP) wet etching. Figure 3 2 A shows the final struct ure. T he measured attenuation of the micromachined FGC line is 0.115 dB/mm at 60 GHz The fabricated line has lower dielectric, ohmic and radiation losses in addition to lower parasitic and coupling effects between adj acent lines.
66 Figure 3 2 Bulk mi cromachined transmission lines. A ) T renched finite ground coplanar (FGC) micromachined transmission line. B ) Dielectric membrane supported coplanar wavegu ide transmi ssion line (CPW) (Adaptation of Figures 1(c) and 2(c) from K. J. Si micromachined coplanar waveguides for use in high frequency circuits IEEE Trans. Microw Theory Tech., vol. 46, pp. 762 768, Copyright 1998 with permission from IEEE ). C ) Partially membrane supported CPW line with SiO 2 membrane (Adaptation of Figure 2 from V. IEEE Trans. Mic row Theory Tech vol. 45, pp. 630 635, Copyright 1997 with permission from IEEE ) D ) High aspect ratio CPW transmission lines (hicoplanar) (Adapt ation of Figure 5 from S. T. Todd, X. T. Huang, J. E. Bowers and N. C. MacDonald, Fabrication, modeling, and characterization of high aspect ratio coplanar waveguide," IEEE Trans. Microw Theory Tech vol. 58, no. 12, part 1, pp. 3790 3800, Copyright 2010 with permission from IEEE) E ) Membrane supported cavity backed grounded CPW transmission line (GCPW) (Ada pt ation of Figure 1(b) from Y. Yoshida T. Nishino, J Jiao S Lee Y. Suehiro, K. Miyaguchi T. Fukami, M. Kimata, O. Ishida, A novel grounded coplanar waveguide with cavity structure," in Proc. Micro Electro Mechanical Systems Conf. Kyoto vol., no., pp. 140 143 Copyright 2003 with permission from IEEE ). A different approach reduces the dielectric losses by partially or completely removing the substrate underneath the signal trace, which creates a suspended thin dielectric membrane backed by an air cavity. In [6 4 ] anisotropic wet etching is used to remove part of the silicon substrate in order to create a thin membrane. Then, CPW transmission lines are fabricated on a thin dielectric membrane made of SiO 2 /Si 3 N 4 /SiO 2 which is grown on the surface o f the substrate The lines Ground Signal Ground Si Si Dielectric membrane Signal Ground Ground Si Si Air SiO 2 Open areas Ground Ground Signal Si Air M embrane A B C D E Ground Ground Signal Si SiO 2 Ground Ground Signal Air cavity Si Si
67 are characterized up to 40 GHz and their performance is compared with th at of CPW lines fabricated on bulk high resistivity silicon wafers, featuring lower dielectric loss. Similar work ha s used different dielectric materials for implementing the thin membrane, such as SiO 2 [65 ] a nd polyimide [66 ], with similar results in term of attenuation, ranging from 0.06dB/mm at 6 GHz to 0.4 dB/mm at 40 GHz. Some membrane based structures are shown in Figure 3 2 B a nd Figure 3 2 C High aspect ratio coplanar waveguide transmission lines (hicoplanar) have been implemented by using a combination of m icromachining techniques. In [67 ] Todd et al. have used deep reactive ion etching (DRIE), therm al oxidation, an i sotr o pic oxide etching, electroplating and chemical mechanical planarization for the fabrication of a thick membrane (38 m) that implements a high performance cavity backed hicoplanar waveguide transmission line with low conductor loss. An a ttenuation constant of 2.4 dB/cm at 30 GHz has been obtained. The fabricated structure is shown in Figure 3 2 D Membrane supported grounded coplanar waveguide (GCPW) transmission lines have been also implemented by u sing bulk micromachining. In [68 ] Yoshida et al. present a GCPW transmission line fabricated by using a combination of alkaline etching, thick photoresist patterning and chemical mechanical planarization (CMP). Silicon nitride is used as the dielectric material for the membrane. The ground plane is metalized on the cavity after the alkaline etching is performed. A maximum attenuation of 0.08 dB/mm is obtained at 20 GHz. The fabricated structure is shown in Figure 3 2 E The work mentioned ab ove has offered new ways to implement miniaturized micromachined transmission lines for MMIC applications and open ed a new area for the development of wireless communication devices and RF MEMS. However, bulk micromachining i s not the only technique that h as been used for these new implementations.
68 3.2. Surface Micromachined Transmission Lines A different group of transmission lines for RFIC applications has been fabricated by using surface micromachining techniques, which instead of removing part of the substra te incorporate the use of different non photopatternable and photopatternable organic materials on top of the substrate as dielectric membranes or dielectric interface layers in /on which the transmission lines are implemented. Further, this approach allows the fabrication of multilayer structures with vertical interconnections for microwave and millimeter wave applications. In  Ponchack et al. characterized thin film microstrip lines (TFMS) on polyimide. A thin polyimide layer is deposited onto a previo usly metal coated silicon wafer used as a carrier substrate. Thicknesses from 2.45 m to 7.45 m are used. Standard metal evaporation, polymer spinning, lithography and via hole formation by using deep reactive ion etching (DRIE) are used for the fabricati on of the transmission lines. Characterization is performed from 1 to 110 GHz, featuring a maximum attenuation constant around 5 dB/cm at 60 GHz for a TFMS line implemented on a 7.45 m thick polyimide. The cross section of the fabricated structure is show n in Figure 3 3 A Following his own work, in  Ponchack et al. implemented low loss CPW transmission lines on low resistivity silicon substrates with a micromachined polyimide interface layer. A 20 m thick poly imide is deposited as an interface layer on top of a low resistivity silicon substrate. The thickness of the polyimide is optimized in order to minimize the interaction of the electromagnetic field with the lossy silicon substrate. Further steps of lithogr aphy, metallization and DRIE are followed to create a trenched CPW line in which the polyimide between conductors has been removed in order to reduce the attenuation constant. The cross section of the structure is shown in Figure 3 3 B The etching of the polyimide lowers the
69 effective dielectric constant, line capacity and current density, thus, decreasing conductor and dielectric losses. A maximum attenuation of 2.75 dB/cm is obtained at 40 GHz. Figure 3 3 Surface micro machined transmission lines. A) Thin film microstri p l ine (TFMS) on polyimide (Adaptation of Figure 1 from G. E. Ponchak and A. N. Downey, Characterization of thin film microstrip lines on p olyimide, IEEE Trans. Comp., Packag., Manufact. Technol. B vol. 21, pp.171 176 Copyright 1998 with permission from IEEE ) B) Trenched CPW line on a micromachined dielectric layer ( Adaptation of Figure 1from G. E. Ponchak, A. Margomenos and L. P. B. Katehi, "Low loss CPW on low resistivity Si substrates with a micromachined polyimide interface layer for RFIC interconnects," IEEE Trans. Microw Theory Tech. vol. 49, pp. 866 870 Copyright 2001 with permission from IEEE ). C) G rounded CPW lin e on SU8 ( Adaptation of Figure 1from F. D. Mbairi and H. Hesselbom, High frecuency design and characterization of SU 8 based conductor backed coplanar waveguide transmission lines, in IEEE Int. Symposium on Advanced Packaging Materials Copyright 2005). D) Coplanar s trips line on SU8 ( Adaptation of Figure 1from M. S. Arif and D. Peroulis, Loss optimization of in Proc. Asia Pacific Microw Conf. Copyright 2009 with permission from IEEE ) Similar work is presented in , in which a thick SU8 layer is used as a dielectric interface on top of a low resistivity silicon wafer. The use of SU8, which is a negative tone photoresist widely used in MEMS fabrication, offers various thicknesses, high aspect ratio imaging, low c uring temperature, high transparency and easy fabrication of multiple layers, in contrast with the use of polyimide. The SU8 is patterned by using standard UV lithography, S Si H SU8 W S A B C D W Si H Polyimide t Ground S Si H Dielectric L ayer t W W S Carrier Substrate H W W SU8
70 followed by metallization and lift off techniques to define the trenched CPW lines. Maximum attenuation of 0.6 dB/mm at 60 GHz is achieved. Although the attenuation is higher than in previous work with polyimide, the use of SU8 offers advantages in the fabrication process. In a similar way, Benzocyclobutene (BCB) has also been used as an interface layer for implementing low loss CPW transmission lines on low resistivity silicon, although with increased fabrication costs . Grounded coplanar waveguide (GCPW) and coplanar strip (CPS) transmission lines have been also implemented by using SU8 as a dielectric. In  GCPW transmission lines are fabricated and characterized up to 50 GHz. A 35 m thick SU8 layer deposited on top of a metalized substrate, in this case glass, is used as the dielectric material. Maximum attenuation of around 3. 5 dB/cm is obtained at 50 GHz. The work also provided the loss tangent and dielectric constant of the SU8. The cross section of the structure is shown in Figure 3 3 C On the other hand, in  loss optimization of CPS lines on low resistivity silicon is presented as shown in Figure 3 3 D SU8 thicknesses of 10 m and 15 m are s elected for th e implementation. Conventional U V lithography, metallization and lift off technique s are used for fabrication. A maximum attenuation constant of 0.9 dB/mm at 40 GHz is obtained, which is comparable with th at obtained in previous work reporting CPW transmission lines. Further, the use of a less lossy substrate such as polyimi de or BCB can decrease the loss, at the expense of higher fabrication complexity and costs. Finally, a complicated dielectric post elevated structure with air gap as dielectric, which basically eliminates dielectric losses, has been demonstrated for the implementation of microstrip lines for millimeter wave applications by using surface micromachining techniques [ 6 9 ]. By elevating the microstrip line, the loss can be reduced to
71 1.1 dB/cm at 50 GHz, in comparison with the 10dB/cm achieved by a conventional microstrip line implemented for comparison. It is clear that micromachining techniques have been used for the implementation of transmission lines and passive components at microwave and millimeter wave frequencies. These techniques offer great flexibility during th e fabrication and allow the implementation of vertical 3D interconnected multilayer devices compatible with the CMOS MEMS technology. On the other hand, it is worth to mention that multilayer lamination technologies, such as the low temperature cofire cera mic (LTCC) [70 72 ] and the liquid crystal polymer (LCP) [73 75 ] have been used, in combination with MEMS fabrication techniques, for the implementation of compact 3D integrable passive microwave and millimeter wave devices In the same way, some work ha s explored the microfabrication of metamaterial transmission lines. In the next session, a review of left handed transmission l ines fabricated by using MEMS or lamination processes is provided 3.3. Multilayer and Microfabricated CRLH Transmission Lines Applica tions Multilayer metamaterial transmission lines and passive devices are implemented in a variety of technologies. The need for size reduction, high frequency operation and the 3D vertical integration with multilayer devices extended the transmission line approach to the implementation of compact multilayer CRLH transmission lines on conventional printed circuit board processes an d ceramic LTCC processes. In [26 ] Horii et al. introduced a n unbalanced s uper compact multi layered left handed transmission line with a narrow left handed passband The implementation is done based on the conventional printed circuit board technology with multiple layers containing parallel plate capacitors and grounded meander line inductors for providing the left handed contribut ions Since the direction of propagation is on the vertical direction, perpendicular to the ground plane, large electrical length can be achieved over a short
72 footprint of the transmission line. A diplexer for 1 GHz / 2 GHz operation is implemented by usin g the compact multilayer tr ansmission line. Further, in [76 ] the same author proposed the balanced version of the transmission line with a wide bandwidth from 1.8 GHz to 8.7 GHz, featuring at the same time less inner coupling between capacitors and lack of in band transmission zeros. Following the same trend in the implementation of miniaturized CRLH transmission lines, in [77 ] Nguyen et al. presented a 6 GHz 3dB coupled line coupler implemented with CRLH transmission lines that incorporate metal insulator metal (MIM) capacitors for providing the left handed contributions. Conventional printed circuit board was used for the implementation. A 30% bandwidth around the design frequency is obtained. The coupler features compact size, tight coupling and spurious free response. The schematic of the proposed structured is shown in Figure 3 4 A Figure 3 4 Unit cells of multilayer CRLH transmission lines A ) MIM CRLH unit cell us ed in [ 77 ] (Reprint of Figure 1(a) from H.V. Nguyen and C. Caloz, Simple Design and Compact MIM CRLH Microstrip 3 dB Coupled Line Coupler, in IEEE MTT S Int. Microw. Symp. Dig ., C opyright 2006, with permission from IEEE). B) A similar MIM CRLH unit cell implemented on LTCC [ 27] (Reprint of Figure 2 from A Rennings, T. Liebig, C. Caloz and P. Waldow, CRLH series mode zeroth order resonant antenna (ZORA) in Proc Asia Pacific Microw Conf. Bangkok, Thailand, Copyright 20 07, with permission from IEEE). A B
73 Although the previous mentioned works used the conventional printed circuit board technology, the ar ch itecture is useful for implementation on advanced lamination processes such as low temperature cofire ceramic (LTCC) or li quid crystal polymer (LCP). In  Piatnitsa et al. demonstrated fully integrated multilayer LTCC 3 dB and 10 dB directional, as well as dual band rat race couplers, u sing a combination of right handed and left handed transmission lines. The structure fe atures an area four times smaller than conventional couplers implemented for comparison. Further, in [2 7 ] a CRLH zeroth order resonant antenna (ZORA) is implemented in the LTCC technology. The schematic of the unit cell is presented in Figure 3 4 B Metal insulator metal (MIM) capacitors and grounded stub inductors are used for the left handed contributions. The antenna exhibit s an excellent efficiency of 71% and a high gain of 10 dB around 11.5 GHz. The high gain is provided by the large electrical size of the antenna (2 0 ), which is a result of the series implementation. Leaky wave antennas and dual band couplers designs by using multilayer CRLH transmission lines on the LTCC technology have been also implemented [71 72], demonstra ting high performance and a reduced size. Although f lexible substrates, such as liquid crystal polymer (LCP), have been explored for the implementation of transmission lines and passive devices at microwave and millimeter wave frequencies, no significant work has been presented so far on metamaterial devices in the LCP technology [73 75 ] The implementation of devices on liquid crystal polymer have used MEMS fabrication techniques, such as conventional UV lithography, metallization, lamination and via form ation tro ugh laser or mechanical machining and can be a good alternative for implementing metamaterial devices on low cost organic substrates. On the other hand, t he demand for 3D CMOS MEMS integrable miniaturized devices, MMIC applications and higher o perating frequencies s eems to be the motivation for using
74 microfabrication techniques in the implementation of left handed transmission lines and applications Qin et al  successfully implemented left handed transmission lines operating at the V band by using the negative epoxy photoresist SU8 as a dielectric. The mushroom structure [ 11 ] is selected as the unit cell for 1 D and 2 D CRLH transmission line s. W ide left handed passband between 42 GHz and 73 GHz is achieved for 5, 7 and 9 unit cells CRLH tr ansmission lines The structure of the implemented trasnsmission lines and the measurement results are shown in Figure 3 5 D. Despite of the high insertion loss, the structure demonstrated to be useful for implementing miniaturize d devices for millimeter wave applications. A similar approach is used by Tong et al  for implementing left handed metamaterial coplanar waveguide components (transmission line, short and open stub) on the GaAs MMIC technology with application to a fi lter and a power divider. A m iniaturized left handed bandpass filter working from 1.02 GHz to 1.42 GHz and a power divider working from 2.8 GHz to 3.72 GHz were demonstrated. In similar work, the same authors presented a 3D multilayered left handed b andpass filter on high resistivity silicon, featuring a total area of 2.6 mm 2 . Wide band operation from 2 GHz to 7.5 GHz is achieved. However, the use of GaAs and high resistivity silicon substrates, combined with oxygen plasma to etch thin polyimide layers during the fabrication process, increases fabrication costs and complexity. In addition, full planar implementation of CRLH transmission lines on high resistivity silicon has also been presented. In [ 78 ] interdigital capacitors and grounded stub i nductors are used for the implementation of a CRLH based directional coupler working from 10 GHz to 14 GHz. Standard UV lithography and metallization are used for the fabrication of the coupler. However, it should be noticed that interdigital capacit ors present high parasitic inductance, which causes low self resonance frequency resulting in limited operating bands
75 Figure 3 5 Implemented CRLH transmission line on SU8 A) 5 c ell tran smission line structure. B) Side view of the CRLH TL. C) Measured insertion loss. D) Measured phase highlighting the left handed region. (Reprint of Figures 1 and 6 f r om C. Qin, A. B. Kozyrev, A. Karbassi v band le ft handed Metamaterials vol. 2, Copyright 2007, with permission from Elsevier ). Recent work ha s integrated MEMS devices with microfabricated CRLH transmission lines in order to c reate switchable devices. In [79 ] MEMS switches are int egrated with a CPW CRLH transmission line implemented on the Silicon on Glass (SOG) technology. Left handed behavior from 7.5 GHz to 10.4 GHz with an insertion loss of 3.4 dB 0.9 dB is achieved. Moreover, switchable negative to positive phase response or vice versa is obtained by activating the MEMS switches that connect or disconnect MIM capacitors in the structure. Using a similar approach, in [80 ] Ouagague et al. implement reconfigurable CRLH cells with RF MEMS switches on the high resistivity (HR) si licon technology. MEMS switches and shunt stub capacitor s are used to modif y the frequency range of the left handed and right handed behavior s. Measurement results from 2 GHz to 30 GHz are provided, showing 15 % of frequency tunability by adding the shunt stub capacitor and up to 150 % by changing t he MEMS capacitor state B D A C
76 Recently, a new work demonstrating a different configuration of a CRLH unit cell with an all pass behavior up to 35 GHz has used high resistivity silicon and air bridges . The previo usly mentioned research works ha ve demonstrated the feasibility of the integration of CRLH transmission lines and RF MEMS devices and fabrication technologies to achiev e flexible and reconfigurable applications. As a summary, it is observed that a great variety of fabrication technologies are used for implementing compact multilayer metamaterial applications in the microwave and millimeter wave range. However, most of the work use s a high resistivity silicon or GaAs substrate, which increases fabrication costs and is not compatible with the conventional CMOS technology As previously discussed, some work ha s demonstrated the possibility of implementing miniaturized conventional transmission lines on low resistivity silicon by using dielectric interface la yers. However, due to the availability of printed circuit board (PCB) and LTCC processes, and the technical challenges in multilayer fabrication, organic based microfabrication processes based on low resistivity silicon have not been broadly used for imple menting multilayer metamaterial devices for microwave range. It is the aim of our study to develop single and multiband multilayer micromachined metamaterial applications for microwave range up to 4 0GHz. In the next session, the proposed unit cell for the implementation of CRLH metama terial applications in our study is introduced. 3.4. Proposed Multilayer Unit C ell for the Implementation of CRLH M etamaterial Applications T he main focus of our research is the design and implementation of hig hly compact metama terial engineer ed components and devices for microwave and millimeter wave applications At first, t he previously discussed CRLH architecture is combined with a multilayer surface micromachined fabrication process based on the negative tone photopatternabl e epoxy
77 SU8 and the negative tone photopatternable resin Benzoclyclobutene (BCB) as dielectric interface layers implemente d on low cost substrates such as low resistivity silicon, glass and conventional printed circuit board substrates. M ultilayer grounde d coplanar waveguide (G CPW) and finite g round c oplanar w aveguide (FGC) composite right/ left handed bal anced transmission lines are demonstrated for the design of multiband microwave applications up to 4 0 GHz. Moreover, with minor modifications, the unit c ell can be used for implementing a different version of the CRLH architectures such as the dual CRLH [ 17 ] or the new CRLH unit cell proposed in our study The schematic and simplified equivalent circuit of the proposed structure for a multilayer microma chined composite right / left handed unit cell is presented in Figure 3 6 Note the structure is patterned in a dielectric embedded fashion, which can be SU8 or BCB, while the dielectric and upper ground plane is not drawn for structural clarity in Figure 3 6 A The via connects the inductor to the ground plane. The structure allows coplanar waveguide (CPW), grounded coplanar wa v eguide (GCPW), finite ground coplanar waveguide ( FGC) or microstrip (MS) implementations. In this work, FGC and GCPW implementations are explored. Figure 3 6 M ultilayer CRLH unit cell on SU8 or BCB A ) Schematic. B ) S implified equivalent circuit. B MIM capacitor w i g i h To carrier substrate Via Hole SU 8 or BCB embedded inductor A d l i W cap L cap W line L line L L L R /2 L R /2 2C L 2C L C R
78 The left handed inductor L L is implemented using a dielectric embedded meander line inductor. The metal patches on the lower and upper layers create MIM capacitors, which make the left handed capacitance con tribution C L The right handed contribution is provided by the feeding transmission lines at both sides, modeled by the right handed inductor and capacitor L R and C R respectively. The left handed inductance value can be controlled by adjusting the width w i of the inductor trace, the number of turns and the total length L i Although the inductance of an embedded single layer meander line inductor is smaller than that of the conventional solenoid or spiral inductor, it facilitates the fabrication process. N eglecting fringing effects, the left handed capacitance value is controlled by the area of the metallic patches and the separation distance d between the upper and lower layers. The total height of the structure is represented by h The inter capacitor gap g i is selected to be large enough to be negligible in the left handed capacitor design. From the previous description it is observed that the combination of the right handed and left handed components L R C R L L and C L create s a c omposite right/ left hand ed artificial transmission line which presents a frequency range with backward wave propagation and positive phase The design, simulation, fabrication and testing of this unit cell for single and dual band applications at microwave and millimeter wave fr equencies vs presented in our study 3.5. Proposed Multilayer Embedded Substrate Integrated Waveguide Filter Architecture Th e study of devices that use metamaterial concepts is not limited to the composite right/ left handed approach for planar metamaterial tr ansmission lines. A d if ferent metamaterial concepts is used for the implementation of 3D integrable, compact SU8 embedded evanescent mode r esonators and bandpass filters by using the same multilayer surface micromachined fabrication process in combination with the in substrate waveguide. The complementary split
79 ring resonator (CSRR) [ 82 ] which is considered a metamaterial particle, is loaded in a transmission line implemented with the half mode substrate integrated waveguide (HMSIW ) which produce s a reson ance frequency below the characteristic waveguide cutoff frequency due to evanescent wave amplification [ 45 46 ] The evanescent wave amplification concept offers a great size reduction since the resonator can be smaller than the quarter wavelength at the r esonance frequency. Photopatternable SU8 epoxy is selected as the dielectric for implementing the embedded passive devices on low cost carrier substrates. C onventional printed circuit board (PCB) or glass are selected as the supporting substrates for the micromachined filters, which keeps the compatibility with conventional microwave PCB implementations and CMOS integrated circuits Although the complete implementation of these filters require s further study, the cross section of the multilayer embedded f ilters is shown in Figure 3 7 [83 ] Figure 3 7 Cross section of the proposed dielectric embedded resonators and filters. In Figure 3 7 it is observed that multilayer fabrication is possible by adding multiple dielectric layers and vertical interconnections. BCB can also be used as a dielectric for the filters. A long metalized via wall is used for the substrate int egrated waveguide architecture. Since the SU8 or BCB Metal Metal
80 process uses surface micromachining, the supporting substrate can be either silicon, glass, or organic materials such as ones for printed circuit board. The resonator is implemented on top of the first dielectric l ayer. The ground plane is implemented on the second dielectric layer, and the electroplated vertical interconnects are used for the signal line and the vias to ground as proposed in [4 4 ] The upper layer of dielectric allows the implementation of CPW or m icrostrip lines for feeding as well as different components such as wideband antennas. More detail on the implementation of the multilayer filters is further provided in Chapter 5 3.6. Proposed Multilayer Ar chitecture for Micromachined Wideband Bandpass Filters In addition to the previously described embedded filter architecture, we also study the design and implementation of surface micromachined wideband cavity filters. New in s o ubstrate waveguide cavity r esonators are proposed by using reduced mode versions of the original substrate integrated waveguide cavity. Conventional printed circuit board, liquid crystal polymer (LCP) and BCB are selected as dielectric materials for the proposed filters. At first, a wideband bandpass filter is demonstrated on a printed circuit board and fabricated by using a conventional CNC (computerized numeric control) milling machine. Then, LCP is used for the implementation of two pole and three pole filters for 25 GHz. The fabr ication process of the filters on LCP is proposed and uses a combination of mechanical drilling of the via holes and surface micromachining techniques for metal patterning. Finally, millimeter wave filters for 60GHz are also designed. Benzocyclobutene (BCB ) is selected as the dielectric substrate for the implementation. The same micromachining process used for implementing the CRLH devices is used for this purpose. Two a nd four pole wideband filters are designed and simulated The fabrication of the filters is left as a future work of our research hence, only simulated results and analysis are presented. Figure 3 8 illustrates the 3D
81 structure used for the filter implementation on BCB A single coating of 21 m BCB is used. However, multiple coatings are possible in order to increase the substrate thickness. 3.7. SU8 and BCB as Dielectric Materials for RF Circuits Although specific details of the f abrication process is f urther provided, it is the purpose of this section to discuss the pro perties of SU8 and BCB (Benzocyclobutene) as dielectric materials for RF applications. For micromachined RF devices the dielectric material s to be used have two basic requirements: to have appropriate RF electromagnetic properties (dielectric constant an d loss tangent) and to be easily micromachinable. SU8 and BCB can be easily spin coated onto any carrier substrate, such as silicon, glass or printed circuit board organic material. The two dielectric materials are p hot patternable, with good optical, elect rical and mechanical properties that offer great optical transparency, low curing temperature, the capability for high aspect ratio vertical interconnection useful for multilayer implementations, the compatibility and integrability with CMOS/MEMS processes and the batch processability for multiple devices. Moreover, the negative tone SU8 resist and the BCB resin are permanent, which is a requirement since the RF devices are to be fabricated on top of the dielectric. Since they offer a permanent dielectric, photolithography, metallization, electroplating and deposition of multiple dielectric layers are possible. SU8 2000 TM series from Microchem TM and Cyclotene 4026 resin from Dow TM are selected in our study Table 3 1 summarizes the electrical and mechanical properties of SU8 and BCB
82 Figure 3 8 Cross section of the proposed micromachined cavity resonators and filters. Table 3 1 Properties of the dielectric materials Property SU8 2000 series BCB Cyclotene 4026 Dielectric Constant 3.4 00 @ 10GHz 2.65 0 (2 20GHz) Loss tangent 0.027 @ 30GHz 0.002 @ 10GHz Thermal stability 5% wt. loss @ 315 C 1.7% wt. loss @ 350 C T hermal conductivity 0.3 W/mK 0.29 W/mK at 24C Coeff. of thermal expansion CTE (ppm) 52 42 ppm at 25C Tensile Strength (MPa) 60 87 9 2 2.9 0.2GPa Curing temperature 150 C to 250 C 250 C UV processing Near UV 350 400nm Nea r UV 350 400nm
83 CHAPTER 4 COMPOSITE RIGHT/ LEFT HANDED (CRLH) METAMATERIAL APPLICATIONS This chapter introduces some work on single band and multiband composite right/left handed ( CRLH ) metamaterial applications. A dual band three way Bagley po lygon power divider is implemented with CRLH transmission lines  Lumped elements are used to realize the left handed (LH) contri bution of the unit cell, while conventional microstrip transmission lines provide the right handed (RH) contribution. The d ual band behavior of the CRLH transmission lines is discussed theoretically. The procedure for a dual band design using the CRLH TL is also provided. Measur e ment and simulations are compared In the same way, as an advanced implementation of the CRLH appr oach, the second part discuss es the design, fabrication and test of surface micromachined compact CRLH unit cells for broadband and dual band applications using the photosensitive SU8 as dielectric Grounded Coplanar Waveguide (G CPW) CRLH unit cells worki ng up to 8 GHz are implemented The fabrication process on SU8 serves as the initial stage for implementing CRLH unit cells using BCB as a dielectric, which is studied in Chapter 5 Measurement and simulati on results are compared for both cases. At the end, a new CRLH structure with an all pass behavior and triband response is fully demonstrated and proposed as a unit cell for future work in this area. 4.1. Compact Dual Band Three Way Bagley Polygon Power Div ider Using CRLH Transmission Lines A compact dual band three way power divider based on the Bagley p olygon is implemented using composite right/ left handed (CRLH) transmission lines consisting of microstrip lines and lumped elements for the GSM frequencies of 860 MHz and 1.92 GHz [ 21 ] Also, a dual band power divider consisting of conventional quarter wavelength ( /4) transmission lines with shunt connections of open and short stubs has been implemented for comparison. An advantage of using the Bagley p olygon for three way power divider is that it
84 allows an arbitrary phase selection at the third port by using singl e or dual band, CRLH or conventional transmission lines. The CRLH based power d ivider shows an area of 7.95 cm 2 and a fractional 3 dB fractional bandwidth of approximately 8% at both bands while the comparison structure shows an area of 51.98 cm 2 a bandwi dth of 5.8% at 860 MHz, and 2.6% at 1.92 GHz. Less than 5.5 dB insertion loss is achieved in both cases. Also, full wave structure simulations are performed and the results agree well with those of measurement. 4.1.1. Compact Bagley Polygon Power Divider The odd N way Bagley p transmission lines. Figure 4 1 shows the original three way Bagley p olyg on and its equivalent circuit [5 7 ]. The characteristic impedance Z q Z h and Z o respectively. Due to the symmetry of the circuit, the impedance connected at port 2 is Z o / 2 and the input impedance at port 2 is Z o / 3 The impedance seen at the matched port 1 should be Z o therefore the characteristic impedance for the quarter wavelength input transformer is given as Z q = 2 Z o /3 The value of Z h does not affect the matching condition, but usually is taken as the same value of Z q In [8 6 ine is replaced by an arbitrary flexible electrical length line, L a providing phase control at the third port. The matching condition is achieved by specifying the characteristic impedances Z a for the new line and Z q for the quarter wavelength transformer at the input. According to the analysis performed in [8 6 ] for a 3 way divider, the characteristic impedance of the lines connecting port 3 to ports 2 and 4, Z a is given by 2 Z o nge. In this work dual band CRLH and conventional powers dividers of this type are implemented. Results from measurements and full wave simulation are compared and analyzed. The dual band CRLH theory is first studied in the next section, and then the desi gn, simulation and fabrication of the power dividers are addressed.
85 Figure 4 1 Conventional Bagley p olygon three way power divider A ) L ayout schematic B ) E quivalent circuit 4.1.2. Dual B and CRLH T r ansmission L ine T heory R eview Some work describing dual band quarter wavelength transformers use s a combination of conventional transmission lines with open stubs and short stubs [ 87 ]. By using these structures dual band operation is achieved over a modera te range of frequency ratios that depend on the limitation of the fabrication process. The use of very low and very high characteristic impedance for implementing the lines and stubs, in combination with the electrical length depending on the frequency rat io, does not allow significant size reduction and frequency ratios greater than 4.9 and smaller than 2.19 By using the non linear dispersion property of the CRLH transmission lines [16 ] the compact Bagley Polygon power divider can be implemented for dua l band operation at two arbitrary frequencies f 1 and f 2 with a broader frequency ratio. The quarter wavelength transmission lines connecting ports 1 and 2, and 1 and 4 are replaced by dual band CRLH transmission lines w 1 and 90 at f 2 The lines connecting ports 3 and 2 or 3 and 4 can be designed by using single or dual band, conventional or CRLH transmission lines depending on the application. One single band conventional transmission line is used in this work for compar ison purpose s The theoretical analysis of dual A B
86 band CRLH transmission lines is presented below. The CRLH unit cell consists of a right handed (RH) series inductance L R and a shunt capacitance C R which create a conventional transmission line, and a left ha nded (LH) series capacitance C L and a shunt inductance L L which create the artificial transmission line with left handed properties. The characteristic impedance for the right and left handed contributions respectively are given by and (4 1 ) Series and shunt resonant frequencies, se and sh respectively, can be defined for the unit cell by and (4 2 ) When the unit cell is balanced the resonant frequencies are equal which means that L R C L =L L C R and Z oR = Z o L then the characteristic impedance Z o of the CRLH unit cell is given by ( 4 3) By periodically cascading N balanced unit cells a CRLH transmiss ion line can be implemented with a positive phase in the left handed region. The phase of a balanced CRLH transmission line with N unit cells can be approximated b y using the following equation [1 6 ]: (4 4 ) Figure 4 2 shows the phase response of the LH, RH and CRLH transmission lines. The non linear behavior of the LH TL in combination with the linear phase of the R H TL creates a controllable non linear phase for the CRLH TL. A positive phase + and a negative phase can be achieved at two different arbitrary frequencies 1 = 2 f 1 and 2 = 2 f 2 harmonically not related.
8 7 Therefore, /4 transmission lines for dual band operation can be implemented and size reduction is achieved. Figure 4 2 CRLH phase response. A ) P hase responses of LH, RH a nd CRLH unit cell. B ) S chematic of a CRLH unit cell. For the design, the two targeting frequencies are selected and the phases + and required at each freque ncy are specified. The equation system created by Equations 4 3 and 4 4 evaluated at each frequency, are solved for L R C R L L and C L E quations 4 5 to 4 8 show the expressions for calculating the lumped element values of the CR LH transmission line. These equations are in the function of the frequency ratio 2 / 1 ( 4 5 ) (4 6 ) (4 7 ) (4 8 ) B A
88 The number of unit cells N, is selected according to the requirements. The negative phase contributions, which are the electrical length of the RH TL needed to achieve the phase specifications at f 1 and f 2 respectively, are given by and ( 4 9 ) 4.1.3. Design of the CRLH and C onventional Dual Band Q uarter W avelength T ransmission L ines The quarter wavelength transmission lines used in the Bagley p olygon have a characteristic impedance Z q = 57.74 with Z o = 50 The lines L a h ave a characteristic impedance Z a = 2Z o = 100 By following the procedure outlined in [ 87 ], conventional /4 dual band TLs are designed for operating at GSM frequencies of 860 MHz and 1.92 GHz. Figure 4 3 shows t he design schematic. The frequency ratio is 2.23, which gives an electrical length for the lines and stubs of = 55.68 The characteristic impedance of the lines is Z line = 39.41 and the characteristic impedances for the open and short stubs are selec ted to be Z open =102.62 and Z short = 20 respectively. Meander lines allow an additional size reduction in microstrip. An Arlon Diclad 880 (Arlon Materials for Electronics) substrate with r = 2.2 and a thickness h =31 mil is used. The design is optimized in order to achieve the specifications. By following the CRLH dual band design procedure explained in section 4 .1.2, quarter wavelength transmission lines are designed. Phases 1 = +90 at f 1 =860 MHz and 2 = 90 at f 2 = 1.92 GHz are desir ed for dual band operation. Frequency ratios higher than 5 and smaller than 2 are also achievable, limited by the value of the lumped elements and their self resonance frequency (SRF), especially L L which is kept below 20 nH with SRF = 6 GHz for GSM frequ ency designs. Three unit cells with a phase of each cell 1 of 30 at f 1 are selected. Table 4 1 su mmarizes the calculated and selected commercial values for the lumped elements and the
89 phases of the LH, RH and CRL H lines. Surface mounting device inductors and capacitors (Taiyo Yuden, Inc.) are employed for the implementation, and the Arlon Diclad 880 substrate is used. Due to parasitic effects, the lumped elements are selected based on their manufacturer provided f requency response to have effective component values and behavior close to those of calculation at the operating frequencies. Figure 4 4 shows the layout of the dual band CRLH /4 transmission line. Figure 4 3 Dual band quarter wavelength transmission line with shunt connections of open and short stubs A ) S chematic (Adaptation of Figure 2(a) from H. Zhang and H. Xin, in IEEE MTT S Int. Microw. Symp. Dig pp. 1223 1226, Copyright 2008 with permission from IEEE) B ) I mplementation in microstrip. W 3 =3.6 mm, W 1 =0.66 mm, W 2 = 8.04 mm, L 3 =47 mm, L 1 =41 mm, L 2 = 42 mm. Table 4 1 Parameters of the unit cell. Parameter Calculated Real Implementation C L 3.379pF Taiyo Yuden 2.7pF EVK105CH2R7 L L 11.266nH Taiyo Yuden 15nH HK1005_15N C R 1.362pF Zo=57.74 Transmission Line L R 4.538nH LH 54.34 0 at f 1 24.34 0 at f 2 54.64 at f 1 24.94 at f 2 RH 24.34 0 at f 1 54.34 0 at f 2 24.34 RH TL at f 1 54.34 RH TL at f 2 CRLH 30 at f 1 30 at f 2 30.3 0 at f 1 29.4 0 at f 2 A B
90 Figure 4 4 L ayout of the 57.74 dual band CRLH /4 transformer. 2 C L is implemented with 2 2.4 pF. 4.1.4. Implementation. Two Bagley p olygons are implemented in microstrip using the design procedures described in the previous section. Full wave structure simulations and circuital co si mulations using High Frequency Structure Simulator (HFSS v. 10, Ansoft Inc.) and Ansoft Designer v. 2.0 (An soft Inc.) are performed in order to evaluate the performance of the power dividers. Figure 4 5 shows the implemented dual band Bagley p olygon power dividers T he fabrication is carried out by using a CNC milling machine For the convention al dual band architecture in Figure 4 5 A the total lengths and wid ths of the /4 TL are L 3 = 47 mm, L 1 = 41 mm, L 2 = 42 mm, W 2 = 3.6 mm, W 1 = 0.66 mm, W 2 = 8.04 mm, as previously shown in Figure 4 3 Conventional 50 lines are used for the ports of the power divider with W= 2.4 0mm. The total area is 51.98cm 2 For the CRLH dual band divider, L Li = 27 mm and W Li = 1.74mm as previously shown in Figure 4 4. The meander lines allow additional size reduction. The total area is 7.95cm 2 Figure 4 6 and Figure 4 7 show the simulation and measurement results of the return loss (S11) and the insertion loss (S21 or S41) for the two architectures. Dual band operation with center frequencies of 860 MHz and 1.92 GHz is clearl y observed. The simulation and measurement results show good agreement. A frequency shift of 25 MHz around 1.92 GHz associated with the tolerance of
91 the lumped elements and the fabrication process is presented in the CRLH device. Table 4 2 summarizes the test results for both dividers. A maximum insertion loss of 5.5 dB within the bandwidth has been observed in Figure 4 7 Figure 4 5 Implemented dual band Bag ley p olygon power dividers. A) C onventional transmission lines with shunt conn ections of open and short stubs. B ) CRLH transmission line approach. Figure 4 6 Return loss (S11) and insertion loss (S21 or S41) for two implemented Ba gley p olygon power dividers. A) CRLH TL power divider. B) C omparison of CRLH and conventional dividers A B B A
92 Figure 4 7 Insertion loss a t each port for the two implemented Bagley p olygon power dividers. Table 4 2 Summary of power divider measurements. CRLH Conventional Frequency (GHz) 0.86 1.92 0.86 1.92 Insertion Loss (S 21 S 41 ) 5.34 5.04 4.95 4.96 Insertion Loss (S 31 ) 5.32 4.98 5.05 4.75 f o measured 0.87 1.94 0.85 1.94 20dB Bandwidth 7.93% 8.1 0 % 5.8 0 % 2.83 % Phase (S 21 S 41 ) 91.5 0 89.2 0 89.7 0 91.2 0 Magnitude imbalance (S 21 to S 31 ) 0.02 dB 0.06 dB 0.1 0 dB 0.21 dB Area occupied 7.95 cm 2 51.98 cm 2 4.1.5. Summary This section presented a compact d ual band three way CRLH Bagley p olygon power divider for 860 MHz and 1.92 GHz, and a fully right h anded dual band power divider with shunt connections of open and short stubs for comparison. The power divider offers a degree of freedom for the phase at the third port since the half wavelength transmission lines used in the original Bagley polygon desi gn are not used here. The dual band properties of artificial transmission lines allow the selection of two arbitrary frequencies with achievable frequency
93 ratio higher than 5 and smaller than 2, limited only by the value of the lumped elements and their re sonant frequencies instead of the length and impedance of the transmission lines, which is an advantage compared to the conventional dual band /4 transformer approach. CRLH TL based implementation shows an 84% size reduction and an enhanced bandwidth of 8 % for both bands compared to the conventional approach. Full wave simulations confirming the dual band operation are in good agreement with measured characteristics. The insertion loss in both cases is kept lower than 5.5dB, which is acceptable for three w ay power dividers. This work is developed as a preliminary step towards the implementation of multilayer micromachined dual band metamaterial applications. 4.2. Surface Micromachined CRLH Unit Cell on SU8 for Microwave Applications In this section a highly co mpact composite right/ left handed (CRLH) unit cell is designed, simulated and implemented at microwave frequencies using a multilayer surface micromachined fabrication process with the negative tone photopatternable ep oxy SU8 as a dielectric layer. G rounde d coplanar w aveguide (GCPW) CRLH unit cells for broadband operation up to 8 GHz are implemented as a first step towards the implementation of transmission lines working at higher frequencies. Metal insulator m etal capacitors and an SU8 embedded meander in ductor are used for a left handed unit cell The microfabrication process and measurement test results are discussed in detail. The dispersion diagram of a unit cell designed for 2.4 GHz operation is extracted from the S parameters, showing broad left hand ed behavior from 2 to 5.5GHz. Full wave structure and circuital simulation s a re compared with measurement results. For higher frequencies of operation, BCB is used in our work to implement f inite g round c oplanar w aveguide (FGC) CRLH transmission lines. The design, simulation, modeling and testing of the fabricated unit cell on SU8 are provided.
94 4.2.1. Unit Cell Structure and Modelling Figure 4 8 presents t he general s tructure, s implified equivalent circuit and cross section view of the proposed microfabricated composite right / left handed un it cell which has been pr eviously introduced in Chapter 3 Note the structure is patterned in a dielectric embedded fashion, while the dielectric la yer and upper ground plane is not drawn fo r structural clarity in Figure 4 8 A. It is worth to mention that the unit cell can be implemented on a variety of dielectric materials, and for our study SU8 and BCB have been selected. The via connects the inductor to either the upper or lower ground plane of a CPW, GCPW or Microstrip implementations. The left handed inductor L L is implemented using a meander line inductor embedded in the dielectric The metal patches on the lower and upper layers create MIM capacitors, which make the left handed capacitance contribution C L The right handed inductive contribution, L R is mainly provided by the combination of small pieces of the feeding transmission lines at both sides and the MIM capac itor pads The right handed capacitive contribution C R is provided by the combination of the parasitic capacitance between the transmission lines and the ground, and the parasitic capacitance between the meander line and the ground. The left handed inducta nce value can be controlled by adjusting the width w i of the inductor trace, the number of turns and the total length L i Although the inductance of an embedded single layer meander line inductor is smaller than that of the conventional solenoid or spiral inductor, it s single layer architecture facilitates the fabrication process. Neglecting fringing effects, the left handed capacitance value is controlled by the area of the metallic patches and the separation distance d between the upper and lower layers. The total height of the structure is represented by h The inter capacitor gap g i is selected to be large enough to be negligible in the left handed capacitor design.
95 Figure 4 8 The CRLH uni t cell A) Structure of the proposed m icromachined CRLH unit cell. B) Simplified electrical equivalent circuit C) Cross section view of the 3D structure. From the previous description it is observed that the combination of the right handed and left ha nded components L R C R L L and C L create s a c omposite right/ left handed artificial transmission line which contains a frequency range with backward wave propagation and positive phase. E quations 4 1 to 4 9 presented in the previous se ction, are used in this section. 22.214.171.124 Implementation and m odeling of MIM c apacitors u sing SU8 as d ielectric l ayer In this section the design, simulation, modeling and parameter extraction of the integrated MIM capacitors on SU8 are discussed. MIM capacitors ranging from 0 .3 pF to 4.2 pF are designed and simulated. Simulations are performed on the 3D full wave structure simulator HFSS TM The electrical equivalent circuit models for the integrated components and the parameter extraction based on the simulated S and ABCD para m eters are discussed in detail. The Low Res Si or Glass h h 1 d BCB or SU8 BCB or SU8 Via interconnection Ground plane only for CBCPW and Microstrip C Copper B MIM capacitor w i g i h To carrier substrate Via Hole SU 8 or BCB embedded inductor A d l i W cap L cap W line L line L L L R /2 L R /2 2C L 2C L C R
96 design of the CRLH unit cel ls on SU8 uses grounded copla nar waveguide implementation (G CPW), and hence, bottom and top ground planes are used for this case Since the CRLH unit cells to be implemented are symmetrical, th e design and simulation of the MIM capacitor is done by using a configuration of two capacitors in cascade, as shown in Figure 4 9 In Figure 4 9 A the 3D view of the two capacitors details the geometrica l parameters. The cross section of the structure is shown in Figure 4 9 B The top conductor in each side with the dielectric embedded bottom conductor is used to create the MIM parallel plate capacitor, wh ose geometry and basic fo rmula is shown in Figure 4 9 C It is observed that the two MIM capacitors are connected by the small transmission line gap section on the bottom conductor. The fringing effects and parasitic capacitances to the ground planes are not taken into account for the capacitance formula, however, they are considered in the electromagnetic simulation and modeling of the MIM capacitors. The complete electrical equivalent circuit model of the two capacitors in cascade is shown in Figure 4 10 The inductor L S models the parasitic inductance of the MIM capacitor due to the metallic trace. The capacitor C S models the parallel plate capacitance of the MIM capacitor, which is basically due to the capacitor created by t he top and bottom conductors and the fringing effects. The shunt capacitors C P model the overall parasitic capacitance of the m etal traces to the ground plane, which also takes into account fringing effects The inductor L gap models the small inductance du e to the gap and the capacitor C gap models the small capacitance due to the gap. The equivalent circuit is reduced to two capacitors directly connected in cascade i f the gap is selected to be long enough to make the gap capacitance C gap neglectable at the frequency range of interest, so it will behave as an open circuit. In the same way, since the gap is a short piece of transmission line, its inductance is very small so the parasitic inductor L gap can be considered a
97 short circuit at the freque ncy range of interest. With these simplifications, the frequency depend e nt ABCB parameters of the two cascaded paramet ers can be expressed as in (4 10 ) w here [ABCD] 1 and [ABCD] 2 define the ABCD matrix of capacitors 1 and 2 respective ly. In the same way, each capacitor can be define by its frequency depend e nt admittance matrix (Y) as in (4 11 ) w ith Y Cp and Z Cs as the shunt admittance and the series impedance of the capacitor equi valent circuit model, g iven by (4 12 ) and (4 13 ) By using the previous equations and the simulation values of the frequency response of the two capacitors in cascade, the extraction of the parameters of the equivalent circuit model for each capacitor can be performed. In the 3D structure simulator the ABCD matrix is obtained for the two cascaded capacitors, and neglecting the gap capacitance and inductance, the [ABCD] 1,2 matrix for each indiv idual capa citor is obtained as .Then a n extraction procedure, based on the conversion to Y parameters and the solution of linear system equations is performed It is not the aim of the section to show the detailed extraction procedure, but the extracted values of capacitance that are useful for our study are presented in Figure 4 11 F ull
98 wave 3D structure simulations are performed from 1 GHz to 8 GHz. Table 4 3 shows the geom etrical parameters for the implementation of the MIM capacitors using SU8 as a dielectric. The width W is selected based on the design of a 50 GCPW transmission line on a 100 m thick SU8 layer, which is further explained. Figure 4 9 MIM capacitors A) Structure of two cascaded MIM capacitors. B) C ross section view. C) Parallel plate capacitor geometry and formula. Figure 4 10 Equiv alent electrical circuit model of the two cascaded MIM capacitors. Capacitor 1 Capacitor 2 d Area Conductive plates Dielectric d Gap W L A B C
99 Figure 4 11 Simulation of extracted p arameters for the MIM capacitors. A ) Extracted series capacitance C S and calculated t heoretical value C th B ) Extracted parasitic capacitance C P C ) Extracted parasitic inductance L S Table 4 3 Design parameters of the MIM capacitors Parameter Value W 200 m d 2 m L 100 m 750 m G ap 100 m It is observed that by varying the length L of the MIM capacitor, capacitance values ranging from 0.38 pF up to 2.7 pF are obtained. The parasitic parameters are kept very low due to the small size of the capacitors and the relatively thick SU8 dielectric layer between the electrodes and the ground plane With these values of capacitors it is possible to implement CRLH transmission lines working in the frequency range of interest. A B C
100 126.96.36.199 Implementation of m eander l ine SU8 e mbedded i nductors Meander li ne inductors are selected for implementing the inductive left handed contribution of the CRLH transmission lines. Due to the multilayer implementation of the unit cell, the inductors are embedded with a distance of 3 m from the surface in the selected diel ectric (SU8 in this case) and connected to the bottom conductor of the capacitors, which facilitates the fabrication procedure and protects the inductor from oxidation. The geometry of the inductor is presented in Figure 4 12 A A t hree turn inductor is used The width of the trace is w i = 20 m with a separation distance of d = 80 m. The inductance is then controlled by the length L i Figure 4 12 B depicts the cross section of the inductor in order to highlight its embedded nature. Table 4 4 summarizes the geometrical parameters. Figure 4 12 Embedded meander line inductor A) Geometry. B ) Cross sect ion. Table 4 4 Design parameters of the inductor Parameter Value w i 20 m D 80 m L i 400 m to 1000 m Via to ground 100 m 100 m Embedded distance 3 m in SU8 Thickness trace 1 .5 m Co p per A B w i d L i Via to ground Bottom Conductor
101 The eq uivalent electrical circuit of t he inductor is shown in Figure 4 13 A The inductor L s models the connection to the bottom conductor, and the inductor L P models the inductance value of the meander line inductor. The parasitic capac itance of the trace to the ground is modeled by the capacitor C P The conductor loss due to the trace is modeled by the series resistor R S The extraction procedure is base d on the simulation of the frequency response of a transmission line loaded with t he inductor, as shown in Figure 4 13 B Neglecting the inductance L s which is used only for simulation purposes, the admittance of the inductor is defined by the Equation 4 14 Finally, the parameters C P L P and R S are extracted from the simulated fre quency response of the inductor (4 14 ) Figure 4 13 Inductor modeling A) Electrical equivalent circuit. B ) S imulation setup in 3D structur e simulator HFSS TM Figure 4 14 shows the extracted values for a variation in the inductor length, L i from 400 m to 1000 m. It is observed that the parasitic capacitance is kept with a low value, mainly due to the thin 20 m inductor trace and the relatively thick SU8 dielectric layer of 100 m. Inductance values from 1.35 nH up to 2.95 nH are obtained with this variation. The resistor A B
102 values, from 1.2 to 2.8 mainly due to the skin depth in Copper, can be reduced if the thickness of the inductor trace is increased. Full wave structure simulations from 1 GHz to 8 GHz are performed. Figure 4 14 Extracted parameters for the inductor A) Inductance L P an d series resistance R S B ) Parasitic capacitance C P 4.2.2. Implementation of CRLH U nit C ells. As previously studied, the physical implementations of MIM capacitors and meander line inductors have additional parasitic capacitances and inductances, which depend o n the dimensions of these components and can change the behavior of the unit cell if they are not compensated. Figure 4 15 shows the full equivalent circuit of the unit cell taking into consideration the parasitic effects. The com plete equivalent circuit s of the MIM capacitors and meander line inductors are included. The small inductance of the gap is split into two in order to be used for a symmetrical unit cell. The right handed contribution due to the conventional transmission lines are modeled by the capacitor C R and the inductor L R which are split into two because a symmetrical representation of the unit cell is used. Conductor and dielectric loss es are modeled by the resistor R C The resistor R S models the intrinsic resista nce of the inductor trace, which takes into account the metal resistivity and the skin depth effect. A B
103 Figure 4 15 Complete equivalent electrical circuit of the CRLH unit cell including parasiti c contributions. Figure 4 16 General representation of the u nit cell structure A ) L ayout of top microstrip or coplanar waveguide line and the embedded inductor. B ) C ross section view Transition s are used for measurement purposes. CPW top ground planes are not shown for clarity. The via can be connected to either upper or lower ground planes in GCPW implementations. Two different unit cells are implemented in this work: one unit cell with a 90 electrical length at 2.4 GHz, and one unit cell for dual band operation with 45 electrical length at 2.4 GHz and 45 at 5.8 GHz. The dual band unit cell is designed to be used in the design of a dual band CRLH quarter wavelength transmission line. The im plementation of the unit cell is done on SU8 2025 from Microchem TM Figure 4 16 presents the layout of the top transmission line, the C R /2 C R /2 C CP C CP C CP C CP L R /2 L R /2 C gap L gap /2 L gap /2 L L C L R S 2C L 2C L R C /2 R C /2 L CP L CP A B
104 SU8 embedded inductor, and the cross section view of the unit cell. For this application, the h eight of SU8 substrate is selected to be 100 m, which is achieved by double coating of SU8 2025 on a carrier substrate. SU8 2000 series are well known to have good thermal stability, good adhesion, single coating thick films and high resistance to s olvents and electrochemical processes. T he inductor is embedded in SU8 3 m below the top layer, with a trace thickness of 1 m, which is close to a skin depth of copper at 2.4 GHz. The width of the implemented inductors is 20 m, which is selected in orde r to decrease any parasitic capacitance. As previously discussed, the inductors are implemented with three turns with 80 m spacing and variable length according to the required inductance value. The MIM capacitors are implemented with the bottom and top m etallic layers. The interlayer distance d is 2 m in order to achieve high values of capacitance while keeping the overall area small The width of the capacitors is selected to be 10 m less than the width of the conventional top transmission line, giving a good tolerance range to the fabrication process. The theoretical capacitance is calculated based on the formula for the parallel plate capacitor. Further optimization processes are performed in order to achieve design specifications. Table 4 5 Parameters of the unit cells Unit Cell L R (nH) C R (pF) L L (nH) C L (pF) Broadband 90 at 2.4GHz 4.2 00 1.68 0 2.6 00 1.04 0 Dual band 45 at 2.4GHz and 5.8GHz 1.866 0.746 2.388 0.955 Grounded coplanar waveguide imple mentations have been selected for this work. The characteristic impedance of the top GCPW line is 50 which allows the calculation of its width. The length of the top transmission line depends on the phase requirements of the unit cell. The design is bas ed on E quations 4 1 to 4 3 and the dual band design proc edure described in section
105 4.1.2 Table 4 5 summarizes the calculated parameters of the two designed unit cells. In the same way, Table 4 6 summarizes the dimensions of the two implemented unit cells in m. Table 4 6 Dimensions of the unit cells in m Unit Cell w cap h h1 d L cap L ind W line w ind Broadband 90 at 2.4 GHz 200 100 97 2 340 1000 210 20 Dual band at 2.4 GHz and 5.8 GHz 200 100 97 2 590 930 210 20 Figure 4 17 compares the electromagnetic and circuital simulation results of a balanced quarter wavelengt h unit cell for 2.4 GHz operation. A loss tangent of 0.02 is used for the SU8. Less than 0.9 dB insertion loss at 2.4 GHz is expected. The circuital simulation is performed with the extracted values. It is observed that the full wave electromagnetic simula tions show the expected results for the designed unit cells. The next section describes the fabrication process and measurement results. 4.2.3. Fabrication P rocess In this section the procedure used for the fabrication of the unit cells is described. This fabrica tion procedure will be used for the micromachining of all the multilayer CRLH devices to be implemented in this research. For demonstration purposes, a printed circuit board (PCB) FR 4 substrate has been selected for the fabrication, although the fabricati on can also be done on silicon and glass wafers as carrier substrates The fabrication procedure is based on micromachining techniques such as metallization using DC sputtering and electroplating, multilayer coating of SU8 2025, conventional UV lithography of SU8 and negative photoresist, and cleaning procedures  Figure 4 18 illustrates the fabrication procedure. First, substrate cleaning is performed with TCE ( trichloroethylene ), followed by a rinse with Isopropanol, DI water and dehydration in a vacuum oven at 120 C for 10 minutes. Further, a seed layer of 30 nm / 300 nm of Titanium/Copper is sputtered to create the ground plane, followed by electroplating
106 5 m Copper. Finally, a thin layer of 30 nm Titanium is sputtered as an adhesion promoter for the SU8 2025 layer to be coated. Figure 4 17 Comparison of the electromagnetic and circuital simulation for the broadband CRLH unit cell with a 90 phase at 2.4 GHz. Ext racted values are C L = 0.65 pF, C CP = 0.028 pF, L CP = 0.1015 nH, L L =2.6 nH, R S = 2.8 C LP = 0.058 pF. The total length of the transmission line is 2.3 mm. Inter capacitor gap is 100 m. Following the formation of the ground plane, SU8 2025 is coated in a two step process in order to achieve better uniformity. Both coating steps are performed at 1500 RPM for an approximate thickness of 55 m. Edge bead is removed. Soft bake is performed on a hot plate with the temperature ramped up to 65 C at a rate o f 250 C/hour and kept for 30 min. Temperature is ramped up to 95 C at the same rate and kept for 20 minutes. At the end of the soft baking process the samples are allowed to cool down to room temperature on the hot plate. The second coating step is then pe rformed and the soft baking time at 95 C is increase d to 30 minutes. Lithography with conventional UV light (365nm) is performed with with a dose of 240 mJ/cm 2 Post bake is realized performed on a hot plate with the temperature ramped up to 65 C a t a rate of 250 C/hour and kept for 5 min utes Temperature is then ramped up to 95 C at
107 the same rate and kept for 1 0 minutes The samples are develop ed in SU8 developer for 60 minutes and finally rinse d with Isopropanol and blow dry with nitrogen gun. Figure 4 18 Fabrication process for the multilayer CRLH devices. Further, a metallic seed layer of Ti/Cu/Ti (30 nm/300 nm/30 nm) is sputtered on top of SU8 followed by conventional negative p hotoresist patterning of the embedded inductor and bottom patch of the capacitor by using negative resist NR9 8000. Electroplating a 1.5 m Copper layer is performed with a prior etching of the top Titanium layer used for protecting Copper from oxidation A fter electroplating, NR 9 8000 is removed and also the seed layer of Ti/Cu/Ti is time etched. This step creates the bottom conductor of the MIM capacitors and the meander line inductor, along with the vias connecting the ground planes. The next step consis t s of the coating of a 4 m think layer of SU8 2002 to create the interlayer dielectric for the MIM capacitor. Soft bakes is performed on a leveled hot plate with the temperature ramped up to 9 5 C at a rate of
108 250 C/hour and kept for 5 min utes. Exposure i s performed with a dose of 150mJ/cm 2 Post bake uses the same temperature profile used for soft bake. A near 50% planarized SU8 2002 layer is obtained with the coating of 4 m o f SU8 on top of a 1.5 m thick and 200 m width metal layer, which gives 2 m S U8 layer needed for the capacitors. Finally, a layer of Ti/Cu/Ti (30nm/300nm/30nm) is again sputtered, followed by 10 m thick negative resist NR 9800 coating. Lithography is performed with a dose of 300 mJ/cm 2 and the top layer pattern is created. T itaniu m etching is performed with diluted hydrofluoric acid (HF) in de ionized water (DI) with a 1:10 ratio. E lectroplating of a 5 um thick Copper layer is performed to create the top line and capacitor top conductors. At the end, the seed layer of Ti/Cu is time etched. Figure 4 19 shows scanning electron microscope (SEM) photographs of the fabricated unit cell. 4.2.4. M easurement R esults Measurements are performed by using a Cascade Microtech Probe Station connected with an A gilent E8361A Vector Network Analyzer. Short Open Load Thru (SOLT) Calibration is done from 1 GHz to 8 GHz. Figure 4 20 shows the measurement setup. Figure 4 21 shows the measured return and insertion loss for the 50 ohms broadband CRLH unit cell. Less than 2 dB insertion loss is achieved in the left handed range with around 3 dB at 2.4 GHz, mainly due to the tolerance in the fabrication process and difference in the dielectric constant and loss tangent of the SU8. F igure 4 22 shows a broad ened left handed behavior rang ing from 2 GHz to 5.5 GHz with 90 phase at around 2.4 GHz. On the other hand, since the area of the unit cell is very compact, around 5.6 mm 2 an area reduction clo se to 90% is achieved when compared with conventional quarter wavelength transmission line based PCB implementation on a substrate with the same dielectric constant.
109 Figure 4 19 Photographs of t he microfabricated broadband CRLH unit cell A) SEM picture of the unit cell, B) C lose look of the embedded inductor C) C ross section view of the unit cell. Total area is around 5.6mm 2 Figure 4 20 Measurement setup consisting of a Cascade Microtech probe station and an Agilent E8361A VNA A B C
110 Figure 4 21 I nsertion and return loss for a balanced microfabricated CRLH unit cell F igure 4 22 Dispersion relation for the microfabricated CRLH unit cell. The electrical length around 2.4GHz is 90 4.2.5. Summary This section has presented the design, simulation and fabrication of a super compa ct multilayer micromachined CRLH unit cell by using SU8 as a dielectric layer. Circuital and
111 electromagnetic simulations show good agreement with measured data. Almost 90% size reduction, in comparison with one fabricated by conventional PCB processes is achieved for the unit cell. Although insertion loss seems to be large in comparison with that of conventional implementations, mainly due to the relatively large loss tangent of SU8 and tolerances in the fabrication process, the surface micromachined proce ss on SU8 still can be used for achieving compact circuits at microwave frequencies This preliminary step has develop e d the design procedure and fabrication process to be used for the fabrication of CRLH devices working at higher frequencies. 4.3. Bridged Comp osite Right/Left Handed Unit Cell with All Pass Behavior This section studies theoretically and experimentally a modified design of the composite right/ left handed (CRLH) unit cell showing all pass behavior and triband response  By using an additional inductance which cross couples the input and output ports of the conventional CRLH, a bridged CRLH unit cell (B CRLH) is created. This allows additional right handed (RH) wave propagation below the cutoff frequency of the conventional CRLH to DC, forming a new low frequency RH band with a non linear dispersion relationship. Meantime, a mid frequency LH band and a high frequency RH band, which are inherited from the conventional CRLH, are still present, resulting in three band configuration. The dispersion relation and Bloch impedance of the B CRLH are investigated by standard periodic analysis. Balanced conditions to close transitions between the three bands, to realize all pass behavior, are derived. The proposed structure is fully investigated w ith analyt ical calculations, circuital and full wave simulations and physical implementation Furthermore, the triband operation is demonstrated with an open stub configuration. For demonstration purposes, this B CRLH is implemented on PCB technology by using a dou ble layer fashion that combines metal insulator metal (MIM) capacitors, meander line and stub inductors, and a patterned ground plane.
112 Figure 4 23 Proposed topology A) Circuit model of the bridged CRLH (B CRLH) B ) Bridged T topology for analysis 4.3.1. Proposed Bridged CRLH Figure 4 23 A sh ows the circui t model of the proposed bridged CRLH unit cell in the case its size p is much smaller than the guided wavelength ( p << g ). The inductor L B cross couples the input and output ports of the B CRLH unit cell. The two series LC resonators defined by and the shunt LC resonator are inherited from the original CRLH structure. In order to f acilitate the analysis, we start with the balanced case where the series and shunt resonators have the same resonance frequency se = sh = o The left handed and right handed characteristic impedances of the CRLH unit cell are given by and respectively [6 ]. In the low frequency range, below the left handed cutoff frequency cL of the original CRLH unit cell ( < cL ), the series LC resonator is mainly capacitive, while the shunt LC resonator is mainly inductiv e. Meantime, the inductance L B with a low impedance value, cross couples the input and output ports and the B CRLH structure is considered a bridged T left handed unit cell with a right handed wave propagation at low frequencies i.e. below the low cutoff frequency. At frequencies higher than the low cutoff frequency and smaller than the resonance frequency ( cL < < o ), the series and shunt resonators keep their capacitive and inductive behavior, respectively, while the inductance L B A p L R /2 2C L C R L L L B L R /2 2C L B Z 1 Z 2 Z 2 Z 3
113 becomes a high imped ance value. Then, the behavior of the B CRLH unit cell is that of an LH transmission line with backward wave propagation. It is noticed that the operation of the unit cell has a transition from RH to LH operation in the low frequency band, as that of the d ual CRLH (D CRLH) structure [ 17 ], with a possibility of a seamless all pass behavior by choosing an appropriate L B value. At higher frequencies, ( 0 < < cR ), where cR is the high cutoff frequency of the CRLH unit cell, the behavior of the B CRLH unit cell is that of an RH transmission line. Therefore, the proposed topology exhibits two RH and one LH frequency bands with non linear dispersion behavior and with the possibility of all pass behavior. The conditions for seamless transitions from alternating RH LH RH bands are analyzed. 4.3.2. Analysis Bloch analysis is used to investigate the operation of the unit cell. The Bloch complex propagation constant and impedance of the unit cell are given by ( 4 15 ) and ( 4 16 ) where p and p are the attenuation and phase shift per unit cell of size p respectively. The bridged T configuration in Figure 4 23 B is used in the analysis. The A parameter of the ABCD mat r ix of the network is given by (4 17 ) where and  Replacing, (4 18 )
114 where are t he series and shunt resonance frequencies, respectively, is related to the cutoff frequency of a pure left handed (PLH) transmission line unit cell, and is the resonance frequency of the LC resonator that would be created by the bridge inductor L B and the series branches of the B CRLH unit cell. It is observed that at the shunt and series resonance frequencies the A parameter is 1, which means that the complex propagation constant is zero, = 0 and = 0, as in the original CRLH. For the balanced case it is assumed se = sh = 0 and Z oR = Z oL = Z o When the frequency is = B the A parameter is reduced to (4 19 ) where after replacing expressions, A = 1, which means that the comp lex propagation constant is + j = 0 + j It is worth to notice that the point = B does not necessarily represent a seamless transition from the right handed (RH) to the left handed (LH) behavior of the B CRLH. From Equations 4 18 and 4 19 it is obse rved that a seamless transition ( = 0) is possible if an appropriate condition is presented. In order to obtain the all pass behavior of the network, where the propagation constant is purely imaginary, one more condition is evaluated at the left handed cu toff frequency cL of the original C RLH unit cell, forcing = cL = B as given in ( 4 20 ) where After some mathematical steps L B is calculated as in (4 21 ) which defines t he optimum value of the inductor L B needed in the B CRLH unit cell to achieve
115 the all pass behavior with = 0, = and a seamless transition from RH to LH operation at = B = cL Meantime, when the frequency L << R the LH cutoff frequency of the original CRLH unit cell is approximately cL L /2 [11 ]. By replacing terms in Equati o n 4 21 the inductor L B can be approximated to L B = 4L L L R which offers a simpler expression and an initial value that can be further optimized. Figure 4 24 Physical configuration of the B CR LH unit cell. A) T op layer view of the geometry. B) B ottom layer showing the stub inductor and patterned ground plane with g u = 2 mm. C) Top lay er of the implemented unit cell. D) Bottom layer. 4.3.3. Physical Implementation To demonstrate the design concept, the B CRLH unit cell is implemented on a double side fashion. The substrate Arlon DiClad 880 with a thickness of 0.508 mm and a dielectric const ant of r = 2.2 is used. Figure 4 24 A and Figure 4 24 B show the geometrical configurations for the top and bottom layers, respectively. Figure 4 24 C and Figure 4 24 D show the top and bottom views of the physical implementation. Square MIM capacitors with a side length of l c = 5.4 mm B A g w b l c l c g w i g u l i Ground D C
116 are used to provide the LH capacitive contribution, C L A st ub inductor with a width w i = 0.25 mm and a le n gth l i = 3.35 mm is patterned on the ground plane to implement the LH inductor L L The RH capacitor, C R and inductor, L R are provided by the parasitic effects of the MIM capacitors and the stub inductor. The bridged inductance L B is implemented on the top layer using a meander line structure with a total length of 17.3 mm and a width of w b = 0.25 mm, while the ground plane underneath is removed in order to reduce its parasitic capacitance. Feeding lines at bo th ends of the unit cell, with a 50 characteristic impedance, are used for testing purposes, which are de embedded from simulations and measurements. The unit cell is designed based on the previous theoretical analysis. Figure 4 25 A shows the dispersion diagrams for the circuital simulation from 300MHz to 8GHz of balanced and unbalanced unit cells, and for the full wave simulation of a balanced unit cell using Ansys Designer V.6.1. It is observed that the circuital and full wave simulations show good agreement, with differences at higher frequencies due to additional parasitic effects not taking into account in the circuit model In Figure 4 25 A seamless transitions from RH LH RH bands are observed, as p reviously predicted, where the B CRLH unit cell shows all balanced conditions. The first RH region is observed from 300 MHz to 1.8 GHz. The LH band is from 1.8 GHz to 2.8 GHz. The second RH region is observed up to 6.5 GHz. It is also observed that the unb alanced case has stop bands between the RH LH RH regions. Simulated and measured frequency responses, presented in Figure 4 25 B show good agreement. In addition, in Figure 4 26 A the imple mentation of a 50 /4 open stub made of two B CRLH unit cells is presented while the simulated and measured results are shown in Figure 4 26 B. The inset of Figure 4 26 B shows the open stub schematics. In Figure 4 26 B the triband behavior of the B CRLH structure is highlighted Due to the continuous operation, phase angles per unit cell of 45 and 135 are achievable, which gives 90 270
117 for a two unit cell TL. Because of this, two additional peaks appear between the frequencies f 1 and f 2 of interest, as observed in Figure 4 26 B. However, the s e peaks can be eliminated if the all pass c ondition is relaxed and an unbalanced implementation is used. Complete details of the multiband beha vior are left as a future work. Figure 4 25 Simulated and measured re sults of the B C RLH unit cell. A ) dispersion diagrams fo r balanced and unbalanced cases. B ) frequency response of the balanced unit cell. Unit cells parameters are L R = 2.23 nH, C R = 0.89 pF, L L = 2.27 nH, C L = 0.91 pF, and L B = 10.95 nH for the balanced case. C R = 0.59 p F and L B = 5 nH for the unbalanced case. Figure 4 26 Triband two B CRLH unit cells /4 open stub. A) Photograph of the fabricated stub. B) Insertion loss. Operation at f 1 = 0.61 GHz, f 2 = 2.4 GHz and f 3 = 3.5 GHz are highlighted. Additional peaks are due to the continuous mode of operation. A B unbal bal p f 1 f 2 f 3 A B Z o /4
118 4.3.4. Summary A bridged CRLH (B CRLH) metamaterial unit cell design with an all pass behavior consisting of two RH bands and one LH band was proposed and an alyzed. The conditions for achieving seamless transitions between RH LH RH bands were provided. The B CRLH topology also enables multiband operation as the CRLH does, while the B CRLH offers the possibility of more band selections due to the additional low frequency RH band. Also, an achievable low frequency band could be much lower than that of the original CRLH, which allows implementing devices with much greater size reduction. The B CRLH working principle was validated both numerically and experimental ly. Also, a triband operation was fully demonstrated.
119 CHAP TER 5 MICROMACHINED METAMATERIAL UNIT CELLS ON BCB This chapter explores the implementation of highly compact three dimensional (3D) integrable metamaterial based transmission li nes on a low resistivity (10 20 .cm) CMOS grade silicon substrate for microwave and millimeter wave applications. The composite right / left handed (CRLH) architecture is able to be integrated with an integrated circuit (IC) using a multilayer surface micro machined fabrication process as a post CMOS process. The fabrication process employs the negative tone photo sensitive Benzocyclobutene (BCB) as a low loss dielectric interlayer material allowing packaging compatible high performance RF circuits. Since the low temperature and multilayer fabrication is compatible with CMOS/MEMS processes, it allows the batch fabrication of multiple devices and the easy implementation of 3D vertical interconnects. Finite ground coplanar waveguide (FGC), widely used in CMOS an d MMIC design [36 37 ], has been selected for the implementation of multilayer CRLH unit cells and transmission lines suitable for broadband and multiband microwave and millimeter wave applications The design, modeling, fabrication and on wafer characteri zation are presented for 50 compact multilayer finite ground coplanar waveguide (FGC) CRLH unit cells and transmission lines for broadband and multiband operation at Ku and Ka frequencies of 14 GHz and 35 GHz, respectively. Also, the comparison between t he simulation and measurement results up to 40 GHz on the aforementioned 3D electromagnetic structures is provided. The left handed capacitance and inductance components of the CRLH structures are implemented with photolithographically defined Metal Insula tor Metal (MIM) capacitors and BCB embedded meander inductors, respectively, which allows the fabrication of very compact CRLH devices. The fabricated dual band unit cell features a size of 0 /30 at 14 GHz and an insertion loss of less than 2 dB within the passband 
120 5.1. Analysis of Finite Ground Plane Coplanar Waveguide Transmission Lines on BCB A CMOS grade silicon wafer with a thickness of 280 m and a resistivity of 10 .cm is used in this work. Prior to t he design of CRLH unit cell, a finite ground coplanar w aveguide transmission line on BCB is designed, implemented and tested in order to analyze the attenuation offered by this type of line on BCB. Figure 5 1 A presents the cross section view of a finite groun d plane CPW line implemented on the low resistivity silicon with a Benzocyclobutene (Cyclotene 4026 46 from Dow Chemical, r = 2.65, tan = 0.002 at 10 GHz ) interface layer. Figure 5 1 B shows the side view of th e unit cell to be implemented. For the implementation, the BCB thicknesses are h 1 = 14 m, h = 21 m and the inter capacitor distance, d to create the MIM capacitors is 5 m. The selected CPW gap G, width W, and ground plane length W G for a 50 feeding line are 20 m, 55 m and 440 m, respectively. Sputtered Ti/Cu /Ti (30 nm / 300 nm / 30 nm ) and subsequent electroplated Cu with thicknesses of 2 m and 5 m are used for the embedded and top metal layers, respectively. Different values o f the BCB thickness h are used for the analysis of loss. v A loss analysis is performed based on simulation results. Figure 5 2 A shows the simulated attenuation of a 2 mm long CPW line with different values of the BCB thickness, h It is observed that the thicker the BCB layer is the smaller the loss is mainly due to the minimization of the interaction of the electromagnetic field with the lossy silicon substrate. Figure 5 2 B shows the measured performanc e when a 21 m thick BCB layer is used, featuring an attenuation smaller than 0.3 dB/mm at 40 GHz. In our study, the first layer of BCB is 14 m, and the second layer is 7 m thick in order to achieve a total thickness of 21 m. A similar procedure can be used to achieved thicker layers of BCB, however, 21 m is selected in our work.
121 Figure 5 1 Cross section view of the CPW structures A) CPW on low resistivity silicon with a Benzocy clobutene (BCB) int erface layer. B ) Multilayer architecture for the CRLH implementation Figure 5 2 Insertion loss of a CPW line on a low resistivity silicon substrate with a BCB interface layer : W = 55 m, G = 20 m. A) Simulated attenuation, B) Measured and simulated return loss and insertion loss for a 2 mm long CPW line with BCB thickness h = 21 m Figure 5 3 shows the electric field in the cross section of the CPW when the BCB thi ckness h is 21 m, which minimizes the interaction of the electromagnetic field with the s ilicon substrate. The interaction of the electromagnetic field with the silicon substrate results in an eddy current loss, which is significant when low resistivity s ilicon is used. For this reason, this interaction needs to be minimized. The BCB with a thickness of 21 m, which offers a modest insertion loss and can be achieved by two coating steps, is selected in our study However, the use of thicker BCB and less lo ssy substrates (such as glass) will decrease the loss even further. Low Resistivity Si (1 0 .cm) h G Benzocyclobutene (BCB) Ground W G t A B Top metal L cap Low Resistivity Si (10 .cm) h h 1 Embedded Metal d BCB BCB B A
122 Figure 5 3 Electric field in the cross section for a CPW line with W = 55 m, G = 20 m. 5.2. D esign and Implementation of th e Dual Band CRLH Unit Cell and Transmission Line By following the dual band CRLH design procedure described in section 4.1.2, a CRLH unit cell is designed to implement a quarter lambda two unit cell transmission line with dual band operation at Ku and Ka b and frequencies of 14 GHz and 35 GHz, respectively. The same unit cell structure previously shown in section 4.2.1, recalled here for convenience in Figure 5 4 is used for the implementation. Calculated parameters are C R = 0.12 pF, C L = 0.173 pF, L R = 0.297 nH and L L = 0.434 nH. Table 5 1 shows the geometrical parameters of the CRLH unit cell implemented by using the structure introduced in Fig ure 5 4A The first and second layers of BCB are 14 m and 7 m thick, respectively. The inter capac itor distance is 5 m which is achieved as the result of a planarization of 71% for a BCB layer over a 2 m thick and 250 m wide copper trace Small pieces of CPW line with a width W line of 55 m and a length L line of 40 m are used at both feeding sides to compensate for the right handed contribution. Table 5 1 Dimensions in m of the CRLH unit cell on BCB. W Cap h h 1 d L cap L ind W line w ind 250 21 14 5 260 440 55 20
123 Figure 5 4 M ultilayer CRLH unit cell on BCB. A ) 3D schematic of conductors: note the ground and BCB layers are not see n for clarity. B ) Cross section view of the via interconnection The meander inductor is embedded in BCB 5 m below the top layer, with a trace thickness of 2 m, which is close to 4 times the skin depth of copper at 14 GHz. The width of the implemented inductors is 20 m, which is selected in order to achieve high values of inductance. In addition, each turn has a turn spacing of 40 m and variable lengths according to the required inductor value. The MIM capacitors are implemented with the bottom and top metallic layers. The interlayer distance d is 5 m. The width of the capacitors is selected to be 250 m and the leng th is variable according to the needed capacitance value. The theoretical capacitance is calculated based on the formula for the parallel plate capacitor. Further optimization processes are performed in order to achieve design specifications. Figure 5 5 shows the extracted parameters of the equivalent circuit of the MIM capacitors with variable length L Ca The equivalent circuit presented in section 188.8.131.52 has been used Losses are n ot taken into account for the simple capacitor m odeling, but they are considered in the complete model of the unit cell. Full wave s tructure simulation of the unit cell is performed in 3D high frequency structure simulator (HFSS, ANSYS Inc.). Figure 5 6 A c ompares the electromag netic and circuital simulation results of the frequency response for a balanced unit cell. Less than 1.27 dB insertion B Low Res Si or Glass h h 1 d BCB BCB Via interconnection MIM capacitor w i g i h To carrier substrate Via Hole BCB embedded inductor A d l i W cap L cap W line L line
124 loss at 35 GHz and a return loss better than 15 dB are expected, which are comparable with the loss values per unit cell obtained in othe r work  [34 ]  Figure 5 6 B shows the simulated results for the propagation constant p It is observed that t here is a dual band operation at 14 GHz and 35 GHz where the phase constant of the unit cell is approxima tely 45 and 45 respectively, as designed. Figure 5 5 Extracted parameters for the MIM capacitors on BCB. Extracted series capacitance C S (2C L ), parasitic capacitance C CP and calculated theoretical value C TH The Extracted parasitic inductance L CP is also shown. Figure 5 6 Simula ted performance of the unit cell. A)Insertion and return loss. B ) Phase constant. The extracted val ues are: 2C L = 0.33 pF, C CP = 0.028 pF, L CP = 0.11 nH, L L = 0.43 nH, C LP = 0.01 pF, L R = 0.0764 nH. A B
125 5.2.1. Loss Analysis In order to analyze the loss factor of different elements in the unit cell, simulations are performed in various conditions that all the parameters are assumed lossless except one parameter, whose loss contribution needs to be identified. Figure 5 7 shows the loss factor as a function of frequency. It is observed that the major loss is caused by the silicon wafer d ue to its low resistivity. Radiation loss is higher in the high frequency band where the size of the unit cell becomes comparable with a quarter guided wavelength Figure 5 8 A and Figure 5 8 B show the cur rent distribution at 14 GHz and 35 GHz respectively, where it is concentrated on the left handed components, especially in the meander inductor. It is speculated that the left handed components have a great contribution to the loss in the unit cell, which is not avoidable Figure 5 7 Simulated loss factor of the unit cell for ideal dielectric, ideal conductor, lossless silicon, radiation loss and total loss Figure 5 8 Current distribution i n the CRLH unit cell A) Current distribution at 14 GHz. B ) Current distribution at 35 GHz. A B Loss Factor=1 |S 11 | 2 |S 21 | 2
126 5.3. Fabrication In our study, a silicon wafer with a thickness of 280 m and a resistivity of 10 .cm is used as the carrier substrate, which maintains compatibility with the standard CMOS/MEMS processes. Cyclotene 4026 46 from Dow Chemical, has been selected as the BCB dielectric interlayer due to its low loss properties (tan =0.002 at 10 GHz). Metallization is perfor med via DC sputtering of Ti/Cu seed layer and Copper electroplating. Fig ure 5 9 illustrates the fabrication procedure. First, substrate cleaning is performed with aggressive Piranha etching (H 2 SO 4 : H 2 0 2 = 7:3 ratio ), followed by rinse in deionized (DI) water and dehydration on a hot plate at 120 C for 15 minutes. Further, a layer of BCB with a thickness of 14 m is coated on the silicon substrate by following the recommended procedure by Dow Chemical [8 4] which can be found in the Ap p endix Soft bake is performed on a hot plate at 100 C for 90 seconds. At the end of the soft baking process the samples are allowed to cool down to room temperature. Lithography with conventional UV light (i line: 365 n m) is performed with the recommended values for exposure, developing and post baking times. The samples are developed for 3 minutes in DS3000 developer at 3 5 C, followed by a second developing step at room temperature to stop the developing process, and fi nally rinsed with DI water and blow dried with a nitrogen gun. Curing is performed in a vacuum oven at 150 C for 15 minutes and 210 C for 40 minutes. Descum is performed for 60 seconds using reactive ion etching (RIE) with a n O 2 :SF 6 gas composition of 90:1 0 at 10mTorr. A metallic layer of Ti/Cu (30 nm/300 nm) is sputtered on top of the BCB layer followed by negative photoresist patterning with NR9 8000 (Futurrex, Inc.) for the embedded inductor and the bottom conductor of the capacitor. Electroplating of a 2 m Cu layer and etching of the
127 Ti/Cu seed layer are performed. A thin 7 m layer of BCB is further coated to create the interlayer dielectric that forms the MIM capacitor. Soft bake, exposure, developing and curing are repeated as the Dow Chemical recom mended procedure Figure 5 9 Fabrication process of the unit cell. BCB is used for achieving the interlayer distance d = 5 m. Finally, a layer of Ti/Cu /Ti (30 nm /300 nm /30 nm ) is sputtered, fol lowed by negative resist NR 9800 coating to a thickness of 10 m, and the lithographical patterning of the top layer. After removing the upper Titanium layer, a 5 m thick Cu layer is electroplated to create the top metal trace and the capacitor top conduc tors. At the end, the seed layer of Ti/Cu is etched away. Figure 5 10 shows the SEM image of the fabricated CRLH unit cell, where the shadow of the embedded inductor is observed. The total area of the unit cell, including just the ground planes but not the launching feeding lines, is 0.975 mm 0.67 mm, which gives 0.0455 0 0.03 0 at 14 GHz and 0.11 0 0.078 0 at 35 GHz, confirming the compact fashion of the implementation. Fig ure 5 11 shows the microphoto graphs in color of the fabricated devices, where the embedded inductor can be appreciated. The unit cell, the two unit cell transmission line and the CPW feeding line used in the initial analysis are presented. DC sputtering of seed layer. Coating of the second layer of BCB and lithographically pattern of the via interconnection. Spin coating of 14 m of BCB layer on Silicon substrate. LR Silicon Pattern of embedded conductor NR9 8000 Cu electroplating of 2 m layer and etching of seed layer Cu LR Silicon LR Silicon LR Silicon LR Silicon LR Silicon Ti Sputtering of seed layer, pattern of top conductor and electroplating of 5 m Cu layer. BCB
128 Figure 5 10 SEM Image of the CRLH unit cell. Figure 5 11 Photomicrographs of the microfabricated devices A ) Unit cell B ) CPW transmission line. C ) D ual band tr ansmission line. A B C
129 5.4. Measurement Results Figure 5 12 shows the measured return and insertion loss of the 50 ohms unit cell. Conventional on wafer SOLT (s hort, o pen, l oad and t hru) calibration is used from 5 GHz to 40 GHz. Measure ments are performed with a vector network analyzer Agilent E8361A and a Cascade Microtech probe station with a probe tip pitch of 150 m. Measurement results show good agreement with simulated results. An insertion loss less than 2 dB and a return loss bett er than 10 dB are achieved within the passband of the CRLH unit cell. As observed, the fabricated unit cell is not completely balanced and the measurement results show additional loss mainly due to the tolerance in the fabrication process, metal roughness variation in the metal thickness, resistivity and loss tangent of the silicon and variation in the loss tangent and dielectric constant of the BCB. The t olerance in the fabrication process and the resolution of the photomask, as well as mis aligning bet ween multiple layers, can cause variation in the width of the inductors and capacitors, which results in the degradation of the performance. However, the results are very promising and a much more optimized fabrication can alleviate the fabrication errors and improve the performance. Figure 5 12 Measured performance A ) Unit cell B ) Two unit cell dual band CRLH transmission line. A B
130 Figure 5 12 B shows the measured performa nce of the two unit cell transmission line, which features a maximum insertion loss of 6 dB at 14 GHz and 5 dB at 35 GHz. The implementation shows higher losses than expected that can be attributed to tolerances in the dimensions of the capacitors and indu ctors, as well as the CPW gap, resulting in a slightly unbalanced frequency response as observed in Figure 5 12 B However, the fabrication process can be optimized in order to reduce loss. Figure 5 13 Measured phase constant of the two unit cells transmission line showing a dual band behavior around 12.8 GHz and 36 GHz. Figure 5 13 shows the measured phase angle of the two unit cell transm ission line. A slight difference in the low frequency is observed, in which the /4 behavior is presented around 12.8 GHz instead of 14 GHz. These results are attributed to the differences in the final dimensions of the capacitors, inductors and the CP W gap during the fabrication process. However, an optimized fabrication process would alleviate these differences. Nevertheless, the measured results are in good agreement with the expected results, demonstrating that an optimized multilayer process by usi ng low resistivity silicon and BCB interlayer is suitable for
131 the implementation of comp act multilayer CRLH devices. Additional steps can be undertaken in order to minimize loss, such as a thicker BCB interlayer that reduces even further the interaction o f the electromagnetic field with the silicon substrate, a thinner inter capacitor BCB layer (which is currently 5 m) to reduce the size of the MIM capacitors, conductor backed coplanar (CBCPW) launching feeding lines (although not preferred), or micromach ined CPW gap by using reactiv e ion etching as featured in [35 ]. 5.5. Summary This section has presented the design, simulation and fabrication of super compact multilayer micromachined CRLH transmission lines by using BCB as a dielectric interface layer on a C MOS grade low resistivity silicon substrate. The selection of BCB is based on its mechanical and electrical properties. Circuital and electromagnetic simulations show good agreement with preliminary measured data. The insertion loss seems to be still high mainly due to cross talk with the lossy silicon substrate as identified from the loss analysis, while the usage of a thicker BCB layer is expected to reduce the loss factor significantly. The surface micromachined process on BCB has demonstrated to be usef ul for achieving compact circuits at microwave and millimeter wave frequencies. A compact dual band unit cell and a two unit cell CRLH transmission line have been designed, fabricated and tested. The fabrication process is being optimized in order to reduc e differences between the simulated and measured results. This process can be extended to the fabrication of a great variety of metamaterial circuits.
132 CHAPTER 6 METAMATERIAL LOAD ED THE HALF MODE SUBSTRATE INTEGRATED WAVEGUIDE High performance m icrowave bandpass filters with low insertion loss, high selectivity, compact size and multiple bands are widely used for wireless and satellite communication systems  Conventionally, high performance filters are implemented with the conventional bulky wavegui de technology, which, however, is not readily compatible with other planar or multilayer integrated circuits technologies such as printed circuit board (PCB), monolithic microwave integrated circuits (MMIC) and complementary metal oxide semiconductor (CMOS ) processes. In order to reduce the size and improve the performance, d uring the years, bandpass waveguide filters have used all kinds of metallic and non metallic insertions . Since the first experimental demonstrations of metamaterial particles exhib iting either negative permeability such as the split ring resonators (SRR), or negative permittivity such as the complementary split ring resonators (CSRR), different implementations that combine the rectangular bulky w aveguide with such structures [92 93] have also been investigated. Such combination has been mainly motivated by their extraordinary property of generating backward wave transmission below the waveguide cutoff frequency. However, m odern technologies demand more and more System on Package (SoP ) or System on Substrate (SoS) approaches to achieve the compactness of the devices and systems. Planar Microstrip and Coplanar Waveguide bandpass filters are good alternatives for hybrid or on chip implementations, while their performance is usually infer ior to that of the bulky waveguide filters in terms of radiation leakage and Q factor [90, 94]. The need for new applications and integration with digital circuitry is the motivation for proposing and implementing planar microwave filters with performance s similar to those provided by the bulky waveguide filters. Recently, the substrate integrated waveguide (SIW) and the half mode substrate integrated waveguide concepts (HMSIW) [41 42] have demonstrated to
133 be key wave guiding structures for the implementat ion of low loss, high quality factor and improved selectivity waveguide bandpass filters on the printed circuit board (PCB) technology. In addition, taking into account the possibility of having forward wave propagation below the waveguide cutoff frequenc y, the substrate integrated waveguide has been combined with complementary split ring resonators (CSRR) for the implementation of compact size and high selectivity bandpass filters . In this chapter, the half mode substrate integrated waveguide (HMSIW) concept and the complementary split ring resonators (CSRR) are used to implement miniaturized bandpass filters working with forward wave propagation below the waveguide cutoff frequency. The half mode substrate integrated waveguide allows additional size reduction compared to that provided by the substrate integrated waveguide (SIW) and more flexibility in the control of the quality factor, since no complicated microstrip to HMSIW transition is needed. At first, a single band bandpass filter working at 5.2 5 GHz and a dual band filter working at 3.4 GHz and 5.85 GHz are presented. The filters make use of CSRR loaded HMSIW resonators working under the principle of evanescent wave amplification. The second part shows how an electrically tunable resonator can b e implemented by using the proposed CSRR loaded HMSIW resonator. The third part presents how the proposed evanescent mode CSRR loaded HMSIW resonator can be used for wireless sensing purposes. Finally, the design of dual band filters by using CSRR loaded H MSIW resonators is presented. 6.1. Single and Dual Band Bandpass Filters Using CSRR Loaded Half Mode Substrate Integrated Waveguide 6.1.1. Theoretical Backround Before studying the proposed resonators, the technology used to implement them is studied. For that p urpose, the conventional rectangular waveguide is first analyzed. The equations that
134 define the propagation of its dominant TE 10 mode are visited. In the same way, in order to illustrate its working principle, the simulation of the electric field distribu tion of the dominant TE 10 mode is provided. In addition, the planar version of the rectangular waveguide, the Substrate Integrated Waveguide (SIW) and its reduced mode version, the Half Mode Substrate Integrated Waveguide (HMSIW) are also studied. The geom etries, design equations and the simulation of the distribution of the electric field for the propagation of the quasi TE 10 mode in both structures are provided. At the end, the proposed evanescent mode HMSIW resonators are introduced. The single band and dual band resonators are analyzed in detail, emphasizing on the working principle, the electrical equivalent circuit and the parameters to be used for bandpass filter design. 184.108.40.206. Rectangular waveguide In Figure 6 1 t he geometry of the conventional rectangular waveguide is presented, where it is assumed a material with an electric permittivity and a magnetic permeability is filling the waveguide. In a conventional waveguide the side a is longer than the side b ( a > b ) . It is not the aim of this section to give a complete mathematical treatment of the rectangular waveguide, but to offer a highlight of the equations that define the propagation of the dominant transverse TE 10 mode. Propagating transverse modes in the waveguide have null longitudinal component of the electric field, E z = 0 . For the case of the TE 10 propagating mode, the field components can be reduced to  (6 1) (6 2)
135 (6 3) and E x = E z = H y = 0 (6 4) where A 10 is the amplitude. Additionally, the TE 10 mode waveguide cutoff frequency, propagation constant and phase constant in the medium are defined by (6 5) (6 6) (6 7) Figure 6 1 Geometry of the rectangular waveguide. T he effective dielectric constant of a rectangular waveguide is o btained as in (6 8) from which is observed the rectangular waveguide presents negative effective electric permittivity below the waveguide cutoff frequency Equation 6 5 shows th e high pass behavior Metallic wall Metallic surface x z y a b
136 of the rectangular waveguide. In order to illustrate the propagation of the TE 10 mode in a rectangular waveguide, Figure 6 2 shows the simulated distribution of the electric field. It is observed that the fie ld is concentrated in the center of the waveguide. Figure 6 2 Simulated electric field distribution for the TE 10 mode in a rectangular waveguide. 220.127.116.11. The substrate integrated waveguide Although th e rectangular waveguide is a useful guiding structure for filter design , due to its bulky configuration the integration with c onventional planar technologies is challenging. To overcome this issue, the substrate integrated w aveguide (SIW) concept was introduced in , which preserves the wave guiding properties of the conventional rectangular waveguide and allows the implementation of high performance bandpass filters and components using the well known standard rectangular waveguide techniques. Figure 6 3 A s hows the SIW basic geometry, where the waveguide is implemented in a substrate of height h and relative dielectric constant r as shown in Figure 6 3 B. Its simp lest configuration consists of fe eding microstrip line s tapered transition s at both ends and the waveguide section of width W and length p A row of metalized via holes replace the metal lic sidewalls of the waveguide. The transitions are necessary to convert the propagating quasi TEM mo de of the microstrip line into a propagating quasi TE 10 mode of the SIW. x y z
137 Figure 6 3 T he substrate integrated waveguide A ) Top v iew with geometrical parameters. B ) Cross section view depicting the metallic via walls. The SIW behaves as a rectangular waveguide if the separation of the metalized via holes is much smaller than the guided wavelength, so the via wall can be considered a metallic wall and the radiation loss is minimized . For design purposes  : (6 9) and (6 10) Equations 6 9 and 6 10 offer good design guidelines, but they are not considered to be completely necessary. On the other hand, the TE 10 cutoff freq uency can be approximated by : (6 11) which is similar to equation (6 5) taking into consideration an effective width W eff_SIW for the SIW, given as in  (6 12) Before illustrating the pro pagation of the quasi TE 10 mode in the SIW, the next section studies its reduced mode version, the Half Mode Substrate Integrated Waveguide (HMSIW) . s D W p Microstrip Line Transition Ground plane Metallic via wall Top metal r h A B
138 18.104.22.168. The half mode substrate integrated waveguide A different version of the in substrate rectangular wav eguide uses only half of the conventional SIW , the so called half mode substrate integrated waveguide (HMSIW). Its configuration is illustrated in Figure 6 4. Its wor king principle is derived from E quations 6 1 to 6 4, where it is observed that the t angential component of the magnetic field along the direction of propagation H z is zero at the center of the waveguide ( x = a/2 ) hence, if the SIW is cut into half along this direction (E plane), the open end will behave as a perfect magnetic wall unde r the condition that the width a is much larger than the thickness of the dielectric h ( a >>h ) [42, 95]. Because of the magnetic wall, the propagating mode in the HMSIW keeps half of the field distribution of the quasi TE 10 mode in the conventional SIW, therefore the TE dominant propagation mode can be denominated a quasi TE 0.5,0 mode . Since the propagation characteristics of the HMSIW are similar to those of the SIW, its design equations can be derived from those of the SIW. The cutoff frequency of the fundamental quasi TE 0.5,0 mo de of the HMSIW is calculated as in  (6 13) where the W eff_HMSIW is the effective width of the HMSIW, calculated by the empirical formulas : (6 14) and (6 15) where
139 Figure 6 4 The half mode substrate integrated waveguide (HMSIW) A ) Top v iew with geometrical parameters. B ) Cross section view depictin g the metallic via wall. The advantages of using the HMSIW is that the size is reduced almost 50% in comparison with the original SIW However, due to its reduced size, the unloaded quality factor is also reduced to almost half of that of the SIW . T o illustrate the wave propagation of both SIW and HMSIW, Figure 6 5 shows the electric field distribution of the dominant quasi TE modes on these two structures. It is observed that the HMSIW propagates a mode that resembles the h alf portion of the TE mode in the SIW, hence, the propagation characteristics of the SIW are preserved by the HMSIW. In addition, Figure 6 6 shows a typical frequency response for an SIW or an HMSIW designed for an 8.2 GHz quasi TE 10 mode cutoff frequency. It is observed that the frequency response is also similar to that of the dominant TE 10 mode in a conventional rectangular waveguide . Therefore, the propagation characteristics of the SIW and HMSIW make them useful for imp lementing microwave and millimeter wave applications with similar performance to those of their rectangular waveguide counterparts [41, 42]. Since the performance and geometry of the SIW and the HMSIW are similar to that of the dielectric filled rectangula r waveguide, the well known rectangular waveguide circuits such as filters, power dividers and couplers can be implemented with similar design procedures [41,42]. The next section discusses HMSIW evanescent mode resonators proposed in our work. Ground plane Metallic via wall Top metal r h s D W HMSIW Microstrip Line Transition p A B
140 Figure 6 5 Electric field distrib ution of the dominant TE mode A) T he substrate integrated waveguide (SIW) B) T he half mode substrate integrated waveguide (HMSIW). Figure 6 6 A typical frequency response for the SIW and HMSIW structures. The cutoff frequency is 8.2 GHz. x y z A B
141 6.1.2. Proposed CSRR Loaded Half Mode Substrate Integrated Waveguide Evanescent Mode Resonators The proposed CSRR loade d HMSIW resonators for single and dual band operation are presented in Figure 6 7  All designs are implemented on the substrate Arlon Diclad 880 with a thickness of 0.508 mm, a dielectric constant of 2.2 and a loss tangent of 0.0009. Each resonator has a row of metalized vias with a diameter d of 0.7 mm and a separation from center to center s of 1.4 mm are used to provide the electric sidewall of the waveguide. The waveguide cutoff frequency is selected to be 8.2 GHz, which i s achieved with an optimized width w of 6 mm for HMSIW implementations. The complementary split ring resonator is etched on the metallic sur face of the waveguide. A 50 microstrip line with a width w 1 of 1.547 mm is connected directly to the waveguide wit h no transition. Figure 6 7 Proposed (HMSIW) resonator with a series of vias for electric walls and complementary split ring reso nator (CSRR) on the top surface. A) Single band. B) Dual band. 6. 1.2.1 Working principle The working principle is based on the epsilon negative behavior of a rectangular waveguide under the cutoff frequency, whose effective dielectric constant, eff for the propagating TE 10 mode in a substrate with relative permittivit y r wa s previously given by Equation 6 8 The complementary split ring resonator is a resonant metamaterial particle that A B
142 provides a negative permittivity at its particular resonance frequency . When the CSRR is loaded on the surface of a waveguide, the interaction with the HMSIW structure causes the resonance frequency to be different to that of the original CSRR and generates a positive effective dielectric constant eff which provides a frequency band with forward wave propagation below the wavegu ide cutoff frequency [ 4 5]. In order to further understand its working principle, at first a conventional complementary split ring resonator is simulated and its electrical permittivity is retrieved by using the method described in . It is not the aim of our study to show the complete retrieval procedure, but to proof the metamaterial behavior The frequency response of the conventional CSRR is shown in Figure 6 8 A where it is observed that the CSRR presents a bandstop behavior at 6.1 GHz. In addition, Figure 6 8 B confirms the negative electrical permittivity of the CSRR at its resonance frequency. Figure 6 8 Simu lated results for a con ventional split ring resonator. A) Frequency response. B ) extracted electrical permittivity and magnetic permeability. Geometrical parameters are a = b = g = 0.25 mm, c 1 =c 2 = 3.85 mm, p = 0.77 mm, l = 1.4mm. Next, the simulated resu lts of the CSRR loaded HMSIW resonator for single band operation at 5.25 GHz are presented in Figu re 6 9 All simulations have been performed by using A B
143 Ansys High Frequency Structure Simulator (HFSS). Figu re 6 9 A shows that the behavior of the HMSIW resonator is bandpass, in contrast to the bandstop behavior of the original CSRR. Moreover, due to the interaction with the waveguide, the resonance frequency the HMSIW is shifted up On the other hand, the effective permittivity of the HMSIW resonator has been retrieved from the S parameters and it is presented in Figu re 6 9 B As observed, the effective permittivity is positive at the reso nance frequency, which indicates that a frequency band with forward wave propagation is created below the original waveguide cutoff frequency. To further confirm the forward wave propagation behavior, the propagation constant p of the HMSIW resonator is shown in Figu re 6 9 C A passband with a positive slope is obtained around 5.25 GHz, which confirms the forward propagation behavior. Moreover, the attenuation constant, presented in the same plot, shows a near zer o value in the passband, indicating a low attenuation to the propagating wave. It is also confirm that the waveguide cutoff frequency is shifted up to a higher value In addition Figu re 6 9 D shows the electric fiel d distribution at resonance, indicating that the propagating mode is mainly concentrated in the CSRR and is completely different from the original quasi TE 0.5,0 dominant mode of the HMSIW. This also is an indication that a tapered microstrip to waveguide transition is not strictly necessary. The same analysis can also be done for the dual band resonator of Figure 6 7 B with similar results. In this case, the inner ring of the CSRR uses a meander line configuration i n order to lower the second intrinsic resonance frequency and achieve a double forward wave propagation below the characteristic waveguide cutoff frequency. Next section studies the resonator characterization for bandpass filter design.
144 Figu re 6 9 Simulated results fo r a CSRR loaded HMSIW resonator. A) Frequency response. B) Extracted electrical permitt ivity and magnetic permeability. C) Simulated dispersion an d attenuation diagram. D) Electri c field distribution at resonance. Geometrical parameters are a = b = g = 0.25 mm, c 1 =c 2 = 3.85 mm, p = 0.77 mm, l = 1.4 mm, w = 6 mm, t = 0 mm. 22.214.171.124 Bandpass filter design The classic methodology for the design of coupled resonator banpdasss filter s is followed . For bandpass filter design, the loaded or external quality factor Q e of the resonator and the internal coupling coefficients M ij between resonators play important roles in the design procedure . According to the filter design specif ications, the circuit for a low pass prototype of the filter is first synthesized After the low pass filter design, the required external quality factor and internal coupling coefficients are determined. The next step is to match the physical C A B D
145 design with the synthesized circuit design. This section briefly discusses the basic design guidelines without giving a complete treatment of the filter theory available in . Resonator characterization : In the proposed resonators the position t of the microstri p feeding line and the offset location l of the complementary split ring resonator in the unit cell control the external quality factor ( Q e ). For simplicity, all designs use t = 0. On the other hand, the distance p from the top of the waveguide also affect s the external quality factor of the resonator, therefore, it is selected to be fixed in the design. Thus, the external quality factor is entirely controlled by the offset distance l Out of four possible CSRR orientations in the waveguide, the proposed co nfigurations in Figure 6 7 show the best transmission responses since the electric field in the HMSIW structure is mainly concentrated at the center. In Figure 6 7 B the in ner ring of the CSRR is modified with a meander line structure in order to reduce the second intrinsic resonance frequency of the CSRR and achieve dual band operation below the waveguide cutoff frequency. Thus, the resonance frequencies can be arbitrarily selected by modifying the dimensions of the CSRR. Selected resonance frequencies and the dimensions of the unit cells are summarized in Table 6 1 Full wave structure simulations are performed using High Frequency Structure Simulator (HFSS, Ansys Inc.). Table 6 1 Dimensions of the proposed HMSIW CSRR resonators Resonator g=a=b=c c 1 c 2 p Single band at 5.25 GHz 0.25mm 3.85mm 3.85mm 0.77mm Dual band at 3.5 and 5.85 GH z 0.25mm 6mm 4.5mm 0.57mm Initially, the external quality factors (Q e ) of the single and dual band resonators are obtained. For that purpose, a double loaded resonator is used in order to get the frequency response in terms of the S parameters. Then, the external quality factor is obtained as in (6 16)
146 where f o is the resonance frequency and BW 3dB is the 3dB bandwidth for the response of S 21 as illustrated in Figure 6 10 A. Figure 6 10 B shows the range of Q e for the single band resonator when varying the offset distance l Figure 6 10 C shows the external quality factor of the dual band resonator at both frequency bands. It is observed that the Q e of the first band achieve higher values than those of the second band, mainly due to weaker second order resonance of the CSRR Figure 6 10 External quality factor Q e of the CSRR loaded HM SIW resonator. A) Illustration of the method used to obtain Q e for a doubly loaded resonator B) Extracted Q e from simulations for the sing le band resonator. C) Extracted quality factors for the dual band resonator at both bands. C A f o f H f L BW 3dB S 21 B f
147 Internal coupling coeffi cient : When two resonators or stages are close each other, the electromagnetic coupling between them causes the original resonance frequency of the resonator to be split into two different resonance modes, as illustrated in Figure 6 11 Conventionally the inter resonator coupling coefficient between the stage i and the stage j M ij is extracted from those two new modes as in (6 17) where f 1 and f 2 are the resonance frequencies of the low and hi gh new generated modes, respectively. Generally, the sign of the coupling coefficient does not affect the design procedure, while it is only important when designing cross coupled ban dpass filters . Different methods can be use to get the internal coup ling coefficient [45, 90]. Figure 6 11 shows one way to extract the internal coupling coefficient, in which the resonators are fed with a weak external coupling which causes a high external quality factor . For this case, a we ak capacitive external coupling is used for that purpose, while the internal coupling is controlled by the inter resonator distance l r Figure 6 12 A and Figure 6 12 B show t he extracted internal coupling coefficient of the two coupled resonator for single and dual band cases. Figure 6 11 Internal coupling coefficient. A) Split resonance frequency. B) Extraction of the coupling coefficient. A B f 1 f 2 S 21 f L I R Weak external coupling
148 Figure 6 12 Extracted internal coupling coefficients. A) Single band case. B) Dual band case. 6.1.3. Two Pole Filter Implementation and Measurement Results In order to demons trate the design concept of HMSIW filters, a set of two pole Chevishev filters have been designed by using the coupled resonator design procedure . A two pole single band Chebyshev filter for 5.25 GHz with a fractional bandwidth of 3.8% and a 0.02dB pa ssband ripple, and a two pole dual band Chebyshev filter for 3.5GHz and 5.85 GHz with a 20dB return loss and a fractional bandwidth of 5.7% and 7.5%, respectively, are designed and implemented. Table 6 2 summarizes the calculated parameters and optimized dimensions. Figure 6 13 T wo pole bandpass filters. A) Single band. B) Dual band. Table 6 2 Calculated paramet ers and dimensions of the two pole HMSIW CSRR filters Filter Q M l l r Size Single band at 5.25GHz 14.16 0.0756 1.4mm 9 mm 0.174 g 0.406 g Dual band at 3.5GHz 11.6 0 0.094 0 0.3mm 8.6 mm 0.117 g 0.263 g Dual band at 5.85GHz 8.6 0 0.127 0 A B A B
149 Figure 6 14 shows the measurement and simulation results of the fabricated filters. The fabricated HMSIW CSRR unit cells and filters are shown in Figure 6 15 The conventional printed circuit board (PCB) fabrication process with a CNC milling machine has been used. Measurements are performed using an HP8510C vector network analyzer after standard short open load through (SOLT) calibration in the frequency range of 2 GHz to 10GHz. Measurement results agree well with those of simulations. A maximum insertion loss of 2dB is obtained for the single band filter. The effect of the connectors and feeding lines has not been extracted, which indicates the total loss of the filter might be less. Minute differe nces can be attributed to the tolerances in the fabrication process. Figure 6 14 Measurement and simulation results. A) Single band resonator. B) Single band two pole filter. C) Dual band re sonator. D) Dual band two pole filter. Simulation Measurement Simulation Measurement Sim. Meas. Simulation Measurement S 21 S 11 S 21 S 11 S 21 S 11 S 21 S 11 A B C D
150 Figure 6 15 Fabricated resonators and filters A) Single band. B) Dual band. 6.1.4. Summary Single and dual band resonators are implemented using the half mode subst rate integrated waveguides (HMSIWs) loaded with complementary split ring resonators (CSRR). Forward wave propagation is achieved below the characteristic cutoff frequency of the waveguide due to the evanescent wave transmission properties of the CSRR. Sin ce no transition is needed, a very compact size is achieved. Single and dual band two pole filters are implemented by using the theory of coupled resonator filters. Full wave simulations are in good agreement with measurements. Minute differences are obser ved and mainly due to fabrication tolerance and additional loss due to the connectors. For comparison, the available external quality factors for the CSRR loaded HMSIW resonators in this work are lower than those offered by the CSRR loaded SIW resonators of reference , mainly due to the reduced size of the HMSIW structure. However, some advantages of using the HMSIW are that the size is reduced and the external quality factor can be easily controlled by three different geometrical parameters. In additi on, since the HMSIW has an open side, tunable applications by loading varactor diodes are easy to implement. A B
151 6.2. Electrically Tunable Evanescent Mode Half Mode Substrate Integrated Waveguide Resonators This section explain s the implementation of electrically tunable evanescent mode half mode substrate integrated waveguide (HMSIW) resonators for S band applications  The previously studied CSRR loaded HMSIW resonator, which achieves forward electromagnetic wave transmission below the characteristic wavegui de cutoff frequency due to evanescent wave amplification, is additionally loaded with a variable capacitor connected to one of the conductors of the CSRR. The capacitive loading changes the effective capacitance to ground, resulting in frequency tuning of the resonator. Three different configurations are investigated with a varactor diode connected between the ground and three different contact points of the CSRR. The external Q factor is slightly affected by the frequency tuning. Two electrical equivalent circuits, representing the three different cases, are used to model the behavior of the tunable resonators. More than 15% tunability is achieved around 3.4 GHz. Full wave structure simulation results are in good agreement with those of measurement. 6.2.1. The Tun able CSRR loaded HMSIW Resonator The single band evanescent mode HMSIW resonator introduced in section 6.1.2 is recalled in this section and shown in Figure 6 16 A The width of the HMSIW, w controls the waveguide cutoff frequency, while the resonance freq uency is mainly controlled by the dime nsions of the CSRR. Figure 6 17B shows the electric field distribution at resonance, which is mainly concentrated on the conductors of the CSRR with the higher concentration in the inner metal patch, suggesting three d ifferent tuning configurations by connecting an external variable capacitor to three different locations of the CSRR.
152 Figure 6 16 The electrically tunable resonator. A) Previously p roposed CSRR loaded HMSIW r esonator. B) Electric f ield distribution at resonance. C) Simulated and measured frequency response of the CSRR loaded HMSIW resonator and a conventional HMSIW structure. D) Layout of the proposed electrically tunable resonator, where connection points A, B and C offer three different resonators (A, B and C). The geometrical parameters are h = 0.5mm, g = a = b = 0.25mm, w = 6.4mm, l x = l y = 4.8mm, l =1.2mm, w L = 1.55mm. All resonators are implemented on a substra te Arlon DiClad 880 with a thickness of 508 m and a dielectric constant of 2.2. The via diameter, d, is 0.75 mm with a pitch, s, of 1.4 mm. The width of the HMSIW, w is 6.4 mm for a waveguide cutoff frequency of 7.1 GHz, as shown in Figure 6 16 C. A 50 microstrip line is di rectly connected. The external quality factor is controlled by the offset distance l Figure 6 16 C shows the measured and simulated results of the original CSRR loaded HMSIW resonato r. The resonance frequency below the waveguide cutoff Min Max A B C D
153 frequency is observed at 3.6 GHz. The simulated and measured external Q factors are 19 and 23, respectively. As shown Figure 6 16 B, the electric field at resonance is not unifo rmly distributed on the CSRR, therefore the electric coupling with the variable capacitance will depend on the location of the connectio n. In Figure 6 16 D t hree different tunable configurations are possible. The variable capacitan ce, implemented by a varactor diode in series with a decoupling capacitance (C DC ), can be connected to three different points: The open sidewall of the waveguide in point A (the outer conductor of the CSRR, which has the weakest electric field at resonance frequency), the inner strip of the CSRR in point B (the conductor between the split rings, on which the electric field is slightly stronger), or the inner conductor of the CSRR in point C (with the strongest electric field, and therefore it is supposed to offer the largest shift in the resonance frequency with a g iven capacitance value). Table 6 3 shows the parameters of the varactor diode (SMV1231, Skyworks TM Inc.). Table 6 3 Parameters of the varactor diode Model SMV 1231 Voltage (V) Capacitance (pF) V R = 15 V R s = 2.5 L s = 0.7 nH 15 0.466 5 0.683 2.5 1.09 0 Figure 6 17 C shows the simulated results of the shift of the resonance frequency for the three reso nators and the electrical equ ivalent circuit in Figure 6 17 A and Figure 6 17 B when the capacitance of the diode is 0.466 pF. The bias circuit and the diode model are considered in the simulations. The re sonator C presents the largest frequency shift. Resonators A and B have slightly different frequency shifts and a dual band frequency behavior, on which the upper resonance is produced by the connection of the variable capacitance to the waveguide structur e, implying a different electrical equivalent circuit compared to that of the resonator C, as depicted in Figure 6 17 A The electrical equivalent circuit of the resonator C, in Figure 6 17 B is similar
154 to that used in , but for this case both the coupling capacitance C C and the CSRR capacitance C R are affected by the variable capacitance. In each case, the inductor L V represents the inductance of the metalized via row, the inductor L c and capacitor C c represent the inductive and capacitive coupling of the HMSIW with the CSRR, and the CSRR is represented by the inductor L r and the capacitor C r Figure 6 17 S imulated results. A) Electrical equivalent circuit for resonators A and B. B) Electrical equivalent circuit for resonator C. Extracted parameters are C c = 0.6 pF, L c =1.07 nH, L v = 0.62 nH, C r = 3 pF, L r = 2 nH, L d = 1 nH. C T models the variable capacitance. C) Simulation of t he resonators with different tuning configurations and the same capacitance value. C T = 0.466 pF. C A B
155 6.2.2. Implementation and Measurement Results The three tunable configurations are implemented and tested. Additional capacitors and RF choke inductors (Taiyo Yu den Inc.) are used to implement the bias circuit for the varactor diode. Three non zero voltages of 2.5 V, 5 V and 15 V are applied to the diode and the frequency shifts measured. Measurements have been performed with a vector network analyzer HP 8719D Figure 6 18 shows the photograph of the fabricated tunable resonators A and C. Figure 6 18 Photograph of the fabricated tunable resonators. Resonator A (lef t) and Resonator C (right). Figure 6 19 shows the measurement results for resonators A and B. For the selected capacitance values, tunable ranges of 17% from 3.36 GHz to 2.78 GHz and 22% from 3.21 GHz to 2.48 GHz are achieved for the resonators A, B, respectively. The external quality factor Q e is modified to lower values between 15 and 17 during the tuning process. Less than 2.18 dB and 2.6 dB insertion loss is observed within the tunable range for the resonators A and B, respect ively, which can be attributed to the loss in the additional lumped elements and the connectors. As observed, the tunable resonators A and B present a degraded dual band behavior,
156 mainly due to the additional capacitive loading of the HMSIW provided by the varactor diode. Therefore, the electrical equivalent circuit shown in Figure 6 17 A is a representation of the dual band behavior of these two resonators, in which the varactor diode is connected as an independent branch represen ted by the capacitance C T and the inductor L d Figure 6 19 Measured results with applied DC voltage. A) and B) Return and insertion losses for r esonators A (solid lines with markers) and B (dashed lines). A B
157 Meanwhile, Figure 6 20 shows the measured results for the resonator C, featuring the largest resonance frequency shift from the primary resonance frequency of the original resonator, deep inside t he evanescent region of the waveguide, which causes lower return losses at resonance and a less than 4 dB insertion loss within the tunable range. As observed, the resonator C keeps the single band frequency response as the original CSRR loaded HMSIW reson ator, which features the forward wave propagation band below the waveguide cutoff frequency and the degraded upper waveguide passband of the original HMSIW structure. Then, the electrical equivalent circuit previously introduced in Figure 6 17 B is a representation of the single band behavior of the resonator C, in which the tuning effect of the diode affects not only the original resonance frequency, but also the out off band transmission zero and the degraded waveguide passband. A tunable range of approximately 14% has been obtained from around 2.7 GHz down to 2.3 GHz. Table 6 4 summarizes the measured results for the three implemented resonators. The size of the resonators, including the bias circuit, is also given. Figure 6 20 Measured results with applied DC voltage for the resonator C.
158 Table 6 4. Summary of the measured results for the three implemented resonators 6.2.3. Summary Three configurations of electrically tunable evanescent mode resonators implemented on the CSRR loaded HMSIW have been studied. The CSRR etched on the top surface of the HMSIW allows forward transmission below the waveguide cutoff frequency. Moreover, by connecting a varactor diode to the CSRR, the resonance frequency can be tuned when different DC voltages are applied. Measurements results are provided around 3.4GHz. A tunability of more than 14% for a variation of 12.5 V DC has been achieved. Additional insertion loss can be minimized by selecting low loss lumped components. The resonators can provide an easy way to implement planar tunable bandpass filters. 6.3. Dual Band Filters Using C SRR and Capacitive Loaded Half Mode Substrate Integrated Waveguide In the previous section, the CSRR loaded HMSIW resonator working below the waveguide cutoff freq uency was used to implement a set of single and dual band two pole filters and an electrically tunable resonator. The approach demonstrated to be useful for the design of miniaturized, low cost bandpass filters. However, for dual band filters the indep ende nt control of the external quality factors and inter resonator mutual coupling coefficients at both bands is challenging with the previous work. In this section, a different approach for implementing compact, pl anar dual band HMSIW filters is investigated. By loading the HMSIW with both a CSRR on the top surface and a capacitive metal patch, two independent frequency bands are generated. Moreover, the external Q factor and mutual coupling coefficient for both bands Resonator Original f o (GHz) f o at 15 VDC f o at 5 VDC f o at 2.5 VDC Tunability Size at f o = 3.6 GHz A 3.63 3.36 3.13 2.78 17.2% 0.256 g 0.254 g B 3.21 2.91 2.48 22.7% C 2.69 2.51 2.32 14 .0 %
159 present relatively independent control. Th e dual band resonator offers a compact size, low loss, and good selectivity, which makes it useful for filter applications A dual band resonator and a two pole miniaturized bandpass filter wit h a size of 0.221 0 0.106 0 are demonstrated on the conventio nal printed circuit board (PCB) technology The external and internal mutual coupling variations are fully investigated. Full wave structure simulation and measurement results are provided 6.3.1. The Dual Band CSRR and Capacitive Loaded HMSIW Resonator Figure 6 21 A illustrates the previously introduced single band CSRR loaded HMSIW resonator. The proposed dual band resonator is introduced in Figure 6 21 B Two loading structures, a CSRR and a capacitive metal patch, are integrated in the waveguide for providing two relatively independent frequency bands below the waveguide cutoff frequency. A row of metalized vias with a diameter d of 0.75 mm and a pitch s of 1.4 mm are used to reali ze the electrical sidewall of the HMSIW. The external Q factor of the lower frequency band is controlled by adjusting the input distance l while the combination of the distance l and the width w s in the metal patch offers the controllability of the second band. The substrate Arlon Diclad 880 ( r =2.2 and a thickness of 0.508 mm) is used for the implementation. The symmetric electrical equivalent circuit of the dual band resonator is introduced in Figure 6 21 C The resonant tank formed by L R and C R models the CSRR, while the capacitor C 2 models the capacitive coupling of the HMSIW with the CSRR . The effect of the capacitive patch is modeled as a series reactance formed by L p and C p The inductances L 1 and L 2 mo del the inductive contribution of the HMSIW, while the inductance L v models the inductive effect of the via wall. In order to validate the electrical equivalent circuit, Figure 6 22 A shows the comparison of the ele ctromagnetic and circuit simulation results. It is clearly observed that two independent
160 resonances are generated at 3.5GHz with an external Q factor of 20.5 and at 5.8GHz with an external Q factor of 22.3. Figure 6 21 Dual band resonator A) Previously proposed single band resonator. B) Proposed dual band resonator. C) Electrical equivalent circuit of the dual band resonator. To further understand the working principle, Figure 6 22 B and Figure 6 22 C show the electric field distribution at 3.5 GHz and 5.8 GHz, respectively, in which it is observed that the CSRR has its major contribution to the first resonance frequency with the field concentrated around the ring, while its contribution to the second resonance frequency is mainly due to the inductive and capacitive couplings with the HMSIW structure. In the same way, it is observed that the capacitive patch has a major effect in the second resonance frequency. C L R C R C 2 C 2 L 2 L 2 L P C p L 1 L 1 L V L V Port 2 Port 1 A B
161 Figure 6 22 Simulated perform ance of the dual band resonator. A) Frequency response. B) Electric field distribution at 3.5 GHz B) Elect ric field distribution at 5.8 GHz. Design parameters are: a = b = c = 0.25 mm, c 1 = c 2 = 4.5 mm, w = 6 mm, L C = 2.8 mm, W C = 5.5 mm, g = 0.25 mm, W s = 0.6 mm, l = 1.7 mm, t = 0 mm, p = 0.3 mm, w 1 = 1.55 mm. Extracted values are L v = 0.6 nH, L 1 = 1.07 nH, L 2 = 0.88 nH, C 2 = 0.53 pF, L P = 0.39 nH, C p = 0.665 pF L R = 2.77 nH, C R = 3.05pF. 6.3.2. Dual Band Bandpass Filter Design and Measurement Results The procedures for coupled resonator Chevishev filter design highlighted in  are followed in this work. The dua l band filter is implemented by cascading two resonators, as shown in Figure 6 23 A The mutual coupling coefficients of both frequency bands are controlled by the inter resonator distance L IR while the distance L I C fine tunes that of the second frequency band. The metal patch does not need to be centered with respect to the HMSIW, which allows A B C
162 easy control of the mutual coupling coefficients. Figure 6 23 b shows the variatio n of the coupling coefficients. Figure 6 23 Proposed dual band filter. A) Topology. B) Coupling coefficient variation. L IC is kept constant as 5.6mm. Figure 6 24 Simulated and measured performance of the implemented devices. A) The dual band resonator. B) The two pole filter. The measured 3dB bandwidths are: 10 % at 3.5 GHz and 9 % at 5.8 GHz. B A L IR L IC A B
163 A two pole coupled resonator narrow band Chevishev filter is designed for 3.5 GHz and 5.8 GHz operation and a 20 dB bandwidth of 10 0 MHz at both bands. The filter is fabricated by using standard printed circuit board fabrication with a CNC milling machine Figure 6 24 shows the full wave simulation and measured results for the implemented filter. Less than 1. 2 dB measured insertion loss is obtained at both bands. It is worth to mention that since the filter ha s narrow bandwidth at both bands, the associate insertion loss is higher than that obtained for a wider bandwidth, because the resonator needs to have a higher external quality factor which implies a weaker external coupling to the feeding line. The slight frequency shift s and d ifferences in the return loss band can be attributed to the tolerance in the fabrication process and the loss in the connectors Nevertheless, simulated results are in good agreement with measurement ones. Figure 6 25 shows the implemente d resonator and filter. Figure 6 25 Photographs of the fabricated dual band devices A) Resonator. B) Filter. 6.3.3. Summary In conclusion, the proposed alternative approach for implementing HMSIW dual band filters allows the relatively independent control of the external Q fac tor and coupling coefficient A B
164 at both frequency bands. It was demonstrated that the capacitive loading of the HMSIW also produces a resonant frequency below the original waveguide cutoff frequency, as the CSRR loading does. By using simultaneous loading, a rbitrary frequencies of operations can be selected, demonstrating that the topology can also be a good candidate to implement miniaturized dual band filters. In addition, we believ ed the structure is useful for implementing narrow band and broadband filters with more than two poles and more than two bands. 6.4. Wireless Pas sive Sensing Application Using a Cavity Loaded Evanescent Mode HMSIW Resonator In this section, a compact cavity c oupled CSRR loaded HMSIW resonator is used for implementing sensing applications at microwave frequencies by using the principle of electromagnetic transduction [9 9 100 ] in which the external perturbations in the electromagnetic field are converted into f requency shifts A small PDMS cavity, whose upper side is confined with a metal coated thin membrane, is placed on top of the CSRR loaded HMSIW original resonator. The electromagnetic field in the close vicinity of the CSRR resonator is perturbed when the cavity upper membrane is pressed or deformed by an external event and a shift in the resonance frequency is produced. This resonance frequency shift can be sensed telemetrically, allowing wireless passive sensing of a variety of external events such as pre ssure and strain. Moreover, the sensing frequency is selected to be in the wireless carrier frequency and therefore no additional frequency conversion module or mixer is necessary for wireless data transmission. Furthermore, the use of an evanescent mode r esonator allows significant size reduction in comparison with the use of conventional bulky waveguide, cavity, coplanar waveguide (CPW) or microstrip resonators. The planar circuit implementation of the half mode substrate integrated waveguide (HMSIW) arch itecture, combined with the easy microfabrication of the cavity and membrane in a variety of materials, such as PDMS in this case, makes the sensor compatible and
165 integrable with PCB and CMOS/MEMS processes while allowing the batch fabrication of multiple devices. Because it operates in a microwave frequency spectrum, it offers high pressure frequency sensitivity. Resonance frequency as a function of an applied pressure is presented. Also, a broadband antenna has been integrated to perform wireless interrog ation of the sensor. 6.4.1. The Evanescent Mode Resonator The originally previously proposed evanescent mode resonator in section 6.1.2 is used in this section as a sensor. The dimensions of the waveguide control the waveguide cutoff frequency and the quality (Q ) factor. The dimensions of the complementary split ring resonator (CSRR) control the resonance frequency. As previously shown in section 126.96.36.199, th e electric field at resonance is mainly concentrated on the conductors and the inner metal patch of the CSR R, which suggests that a perturbation of the electromagnetic field right on top or under the resonator can produce a shift in the resonance frequency. Figure 6 26 Simulated and measure d results for the resonator used in our study Measurement Simulation S11 S21
166 The measured and simulated results of the resonator are presented in Figure 6 26 The substrate Arlon Diclad 880 with a thickness of 0.508mm and a dielectric constant of 2.2 is used to implement a CSRR loaded HMSIW resonator. Metalized vias with a diameter of 0.75 mm and a separation s of 1.4 mm are used to provide the electric sidewall of the waveguide. The CSRR is etched on the metallic surface of the waveguide. A 50 feeding microst rip line is connected directly to the waveguide. The selected resonance frequency and dimensions of the resonator are summarized in Table 6 4 The obtained external quality factor Q e is 17.5. Table 6 4 Dimensions in mm of the resonator g=a=b=c l y =l x s p d w t 0.25 3.85 1.4 0.7 0.7 6 0 f o =5.25 GHz w s =1.55mm l= 1.8 mm 6.4.2. Proposed Sensor Structures The cross section s of the proposed configurations of the sensors u sing the evanescent mode resonator are presented in Figure 6 27 In Figure 6 27 A a dielectric cavity made of PDMS with a thin metal coated membrane on top is used to creat e perturbations in the electromagnetic field that shift the resonance frequency of the resonator. When the membrane is deformed, the height of the air gap changes, which produces a shift in the resonance frequency. For the architecture in Figure 6 27 B a cavity backed resonator is used, which uses a metal coated membrane as part of the ground plane that is deformed when the membrane is under pressure, changing the effective height of the dielectric bi layer under th e resonator and, at the same time, the effective dielectric constant under the resonator. These changes produce a shift in the resonance frequency of the resonator that can be telemetrically detected In our stu dy the p erformance of the first structure is discussed, while the use of the second structure for sensing purposes is left as a future work.
167 Figure 6 27 Proposed sensor configurations. A) Reson ator with cavity on top surface. B) Cavity backe d resonator Figure 6 28 Variation of resonance frequency as a function of the air gap 6.4.3. Effect of the Air Gap in the Resonance Frequency In order to analyze the performance of the resonator as a sen sor, a numerical simulation of the resonance frequency on the variation of the air gap thickness, h is performed by using the high frequency structure simulator (HFSS, Ansys Inc.), assuming a uniform variation of the deflection of the membrane as a first order approximation. The height of the cavity is selected to be 200 m. Figure 6 28 shows the change in the resonance frequency as a function of the air gap thickness. It is observed that the resonance frequency de creases as the height decreases. In this A B
168 simulation only the metallic membrane has been taken into consideration in order to analyze the effect of its interaction with the electromagnetic field on top of the resonator. It is observed that a linear change o f 5 MHz/ m in the resonance frequency is expected. Figure 6 29 Mechanical simulation for the deflection as a function of an applied pressure. A) Simulated results. B) Deflection profile. 6.4.4. Mechanic al Simulation of the Deflection In this application, the range of pressure that can be measured depends on the selected material to implement the metal coated membrane. For demonstration purposes, two widely used materials for MEMS applications have been s elected: polydimethylsiloxane (PDMS) and GPa and 3 GPa, respectively. Finite element COMSOL simulations of the maximum deflection of the membrane under uniform pressure have been performed. For the simulations, a metal lic layer of 1 m Copper under the dielectric membrane has been used. A membrane thickness and area of 20 m and 5 mm 5 mm, respectively, have been selected. Figure 6 29 A shows the simulation results of the maxim um displacement as a function of pressure for coated and non coated membranes. Since PDMS is a more elastic material than Parylene, the achievable deflection of PDMS is larger than that of Parylene for the A B
169 applied pressure. A pressure range of a few Pascal the materials. On the other hand, when the membranes are coated with a thin metallic layer their flexibility is reduced, while allowing smaller deflections with the applied pressure. Figure 6 29 B shows the simulated deflection profile for a uniformly applied pressure of 20 Pa to a non metal coated 20 m thick PDMS membrane. It is observed that the maximum deflection of the membrane at the center is 30 m. Figure 6 30 Broadband antenna. A) Antenna. B) I ntegrated antenna and resonator. C) Measured return loss. C A B Ground on back H1 PW PL ML MW
170 6.4.5. Wireless Interrogation Planar broad band antennas with a gain of 2 dBi are integrated to the senso r for interrogation purpose. Transmitting and receiving antennas are used to wirelessly determine the data of the sensor. Frequency sweep and far field measurements are used to interrogate the sensor. For demonstration purpose, the received signal, filtere d by the sensor, is obtained by using a vector network analyzer HP 8719D Figure 6 30 shows the implemented integrated antenna and resonator with measurement results. It is observed that the return loss of the antenna is changed by integrating it with the resonator The dimensions of the antenna are: PW=11.82 mm, PL = 12.08 mm, H1 = 0.67 mm, ML = 0.67 mm, MW = 5 mm. 6.4.6. Measurement Results A PDMS cavity with a height of 200 m, and a PDMS membrane with an area of 5 mm 5 mm and a thic kness of 90 m are fabricated. For sensing purpose, Ti/Cu layers with a thickness of 30 nm / 500 nm have been sputtered on the PDMS membrane. Also, a 20 m thick metal coated Parylene membrane is used for comparing the performance of the two materials. Mea surement results of the integrated resonator and antenna under four different pressure conditions on the PDMS membrane are shown in Figur e 6 31 A. T he sensitivity curve f/ P and the calculated displacement sensitiv ity f/ d curve are presented in Figur e 6 31 B. In Figur e 6 31 C the measured transmission response from the antenna to the sensor under pressure is presented, demonstrating that the sensor can be connected to another external antenna to completely perform wireless interrogation. Table 6 5 summarizes the obtained data for the two different membranes. Figure 6 32 A shows the implemented sensor consisting of the integrated r esonator and antenna with the cavity on top. Figure 6 32 B s hows the wireless transmission test at a distance of 10 mm.
171 Figur e 6 31 Measured results. A) Return loss as a fun ction of frequency and pressure. B) Frequency and displ acement sensitivity to pressure. C) Wireless transmission performance. Table 6 5 Summary of measurements results Membrane F/ P F/ d Thickness PDMS 16 MHz/Pa 6.5 MHz/ m 90 m Parylene 2.2 MHz/Pa 7.3 MHz/ m 20 m 6.4.7. Summary A passive wireless sensing scheme has been demonstrated using the CSRR loaded HMSIW re sonator at microwave frequencies. The electromagnetic transduction principle is used Pressure Increasing Pressure Increasing A B C
172 on a cavity loaded evanescent mode resonator for a pressure sensor. Size reduction of the resonator is achieved by using the evanescent wave amplification on the half mode substrate integrated waveguide. Perturbations on the electromagnetic field exciting the resonator are used for achieving resonance frequency shifts under pressure or deformation conditions. A broadband antenna is integrated to the sensor, demonstrating th at the configuration can be a good candidate for implementing wireless passive sensing applications working at microwave frequencies. Figure 6 32 Fabrication and test. A) Fabricated wirele ss sensor and antenna module. B) Wireless transmission test. A B
173 CHAPTER 7 MICROMACHINED EMBEDDED EVANESCENT MODE HMSIW BANDPASS FILTER This chapter presents the design and implementation of a 3D integrable, compact SU8 epoxy embedded evanescent w ave half mode substrate integrated waveguide (HMSIW) microwave resonator and a two pole bandpass filter working at 12 GHz (Ku band)  The devices were designed and fabricated by using a multilayer surface micromachining fabrication process using SU8 as the dielectric. Evanescent wave amplification with forward bandpass transmission below the characteristic waveguide cutoff frequency is achieved with the use of the previous com plementary split ring resonator (CSRR) loaded half mode substrate integrated waveguide, resulting in compact resonators with controllable external quality f actor s and dual band operation capability. The in substrate implementation and embedded nature of the resonator and filter allows conventional handling and packaging of 3D filte r microstructures without additional mechanical consideration, which otherwise would require a very delicate and expensive vacuum packaging process. In addition, the device is formed in a mold which is used for its fabrication and electrical performance de gradation after further packaging is minimized [4 4 ]. Photopatternable SU8 epoxy has been selected as the dielectric for implementing embedded passive devices due to its electrical and mechanical properties that offer easy implementations of multiple layers for the fabrication 3D integrated passive devices, the capability for high aspect ratio vertical interconnection, low process temperature, good thermal stability, good adhesion, thick film forming capability with single coating, high resistance to solvent s and electrochemical processes, the compatibility and integrability with CMOS/MEMS/MMIC processes, and the batch processability for multiple devices. Moreover, the use of integrated compact passive devices is suitable for System on Package (SoP) or Syste m on Substrate (SoS) approaches to achieve compactness of the devices and systems. Additionally, since the printed circuit board
174 (PCB) FR4 silicon or glass can been selected as the supporting substrates for the micromachined filters, the compatibility wit h conventional microwave PCB implementations and integrated circuits is maintained. The design, fabrication and characterization of the proposed embedded resonator and filter are presented and discussed in detail. Full wave 3D electromagnetic structure si mulations and circuital simulations based on the electrical equivalent circuit extraction at 12 GHz are presented. In the same way, measurement results at 12 GHz are provided Variation of the external quality factor of the resonator from 12 to 19 is obta ined by varying its geometrical parameters. The resonator and filter feature a great size reduction of more than 50% when compared with conventional PCB implementations. Further, due to the scalability and repeatability of the fabrication process, this 3D integrable implementation can be extended to the use of less lossy dielectric for microwave and millimeter wave frequencies, such as Benzocyclobutene (BCB). This approach for HMSIW bandpass filters can be used for implementing new multiband wireless and s ensor applications and scaled from low GHz to millimeter wave frequencies, while allowing the use of diverse organic substrates for the fabrication of multilayer devices and systems. 7.1. 3D SU 8 Embedded Resonator Design In Chapter 6 compact size, high selectivity and multiband capability resonators and bandpass filters were demonstrated based on the conventional printed circuit board (PCB) technology by integrating the half mode substrate integrated waveguide (HMSIW ) technology with a complementary split ring resonator (CSRR), which provides forward wave propagation below the waveguide cutoff frequency and allows significant size reduction. The working principle of the resonator was fully explained. It was stated tha t the dimensions of the waveguide
175 control the waveguide cutoff frequency and the external quality factor of the resonator, while the dimensions of the complementary split ring resonator control the resonance frequency. On the other hand, t he quasi TE 0.5, 0 mode waveguide cutoff frequency for an HMSIW structure can be calculated with the previously introduced Equation 6 13. Further optimizations provide the optimum dimensions. It is emphasized that the cutoff frequency of the waveguide is selected to be high er than the target resonance frequency of the resonator. Figure 7 1 Electrical equivalent circuit of the implemented CSRR loaded HMSIW resonator Figure 7 1 shows the electrical equivalent circuit for the CSRR loaded HMSIW resonator, which is an adapted version of the electrical circuit presented in [ 45 ]. The inductance L R and the capacitance C R model the CSRR, while the inductance Lv models the via wall o f the waveguide, which gives its intrinsic highpass behavior. The capacitive and inductive coupling of the CSRR with the waveguide are modeled by the capacitance C C and L C respectively. Insertion loss due to the transmission line and the quality factor o f the resonator are modeled by the resistors R W and R R respectively.
176 Figure 7 2 Cross section of the proposed SU8 embedded resonator. The design of the resonator is embedded in SU8 epoxy. SU8 2025 (Micro Chem, Inc.), is selected as the dielectric layer. Figure 7 2 shows the cross section of the proposed SU8 epoxy embedded resonator, previously shown as well in Chapter 3 and recalled in this section for clar ity purposes. Multilayer fabrication is possible by adding multiple SU8 layers and interconnecting layers. For this design, the row of metalized vias is replaced by a long metalized via wall with its width of 100 m, more suitable for the microfabrication process. The heights h 1 h 2 and h 3 are selected to be 50 m each Since the process uses surface micromachining, the supporting substrate can be either silicon, glass, or organic materials such as ones for printed circuit board. For this design, PCB substr ate FR 4 with a dielectric constant of 4.1, comparable with the dielectric constant of the SU8 ( r = 3.4 at 10GHz), is selected as the carrier substrate. The resonator is implemented on top of the first SU8 layer. The ground plane is implemented on the sec ond SU8 layer, and the electroplated vertical interconnects are used for the signal line and the vi as to ground, as proposed in [44 ]. The upper layer of SU8 allows the implementation of GCPW or microstrip lines for feeding as well as different components s uch as wideband antennas. He re, two layers of SU8 and a top ground plane are used. SU8 or BCB Metal Metal
177 Figure 7 3 3D view of the embedded resonator. Figure 7 3 shows the 3D view of the S U8 embedded resonator structure drawn with the 3D high frequency structure simulator (HFSS, ANSYS, Inc.). The ground plane on top of the resonator has been removed for presentation purpose The selected width of the HMSIW, w is 1.4 mm which results from a calculated theoretical waveguide cutoff frequency of 29 GHz with r = 3.4 Table 7 1 shows the dimensions in millimeters of the resonator along with the geometrical dimensions of the complementary split ring reson ator implemented on the top surface of the first SU8 layer of 50 m. Since no transition is necessary, a 50 SU8 embedded microstrip line is directly connected. It is observed that the overall size of the resonator, without taking into account the feeding lines, is smaller than the effective guided quarter wavelength in the medium at 12GHz ( mm ) which offers a great size and area reduction in comparison with conventional quarter wavelength based planar implementations. Electroplated square vertical interconnects with a width of 100 m are used for connecting the embedded microstrip line with an upper 50 GCPW feeding line implemented on the top layer of SU8.
178 Table 7 1 Dimensions of t he resonator W L (mm) H (mm) g ( m) a ( m) b ( m) w (mm) l x =l y (mm) l (mm) 80 0.1 80 80 80 1.4 0.99 0.2 Figure 7 4 shows the simulated results for the designed resonator The resonance frequency is around 12 GHz, which is achie ved by optimizing the dimensions of the CSRR resonator. It is observed that the transmission is obtained below the characteristic cutoff frequency of the waveguide, which is around 29GHz. The external Q factor of this doubly loaded resonator is calculated with Eq uation 7 1 where f r is the resonance frequency and BW 3dB is the 3 dB bandwidth for S21. In this case Q = 14.3 for an offset distance of 0.2 mm. ( 7 1 ) Figure 7 4 Simulated performance of the resonator. The e xtracted circuital parameters are : L V = 0.5 nH, L C = 0.34 nH, L R = 1.17 nH, C R = 0.467 pF, C C = 0.25 pF, R W = 4 Figure 7 5 shows the variation of the Q factor with the offset distance l Electromagnetic simulations are performed for different distances and the external Q factor is calculated with E quation 7 .2. For a maximum distance of 0.25 mm, the Q factor is around 19. Although the
179 res onator shows a lo wer external Q factor than that of other SIW filter implementations, it shows advantages with regards to the size. Figure 7 5 External Q factor variation with the input distance l 7.2. Two Pole Embe dded Filter Design Figure 7 6 shows the top view of the two pole embedded bandpass filter designed by using the coupled reson ator filter design procedure [90 ]. The coupling coefficient between the resonators is con trolled by the inter resonator distance L IR and is calculated with Eq uation 7 2 as follows : (7 2) w ith f 12 and f 22 as the lower and higher resonance frequencies that are obtained when the two resonators are close each other (ev en and odd mode). For example, a two pole single band Chebyshev filter for 12 GHz with a b andwidth of 500MHz for a return loss of 20 dB is designed and implemented. Table 7 2 summarizes the calculation and dimensi ons. Table 7 2 Parameters of the embedded filter Q e M L i L IR BW 14.16 0.075 0.2mm 0.49mm 500MHz
180 Figure 7 7 shows the electromagnetic simulation results for the de signed two pole filter. Since a relatively large loss tangent of the SU8 has been used (0.03), the simulated insertion loss is high. Nevertheless, the simulated results show that the embedded configuration is promising in terms of size reduction, scalabili ty to higher frequencies and integrability with other devices. Finally, i n Figure 7 7 i s observed a 3 dB bandwidth of around 75 0 MHz. Figure 7 6 Top view of the two pole embed ded filter. Figure 7 7 Electromagnetic simulation res ults of the two pole filter. A) Frequency response of th e insertion and return loss. B) Close view of the 10 dB bandwidth. 7.3. Fabrication Proc ess In our study FR4 ( r = 4.1) printed circuit board (PCB) have been selected as th e supporting substrate for the embedded devices, maintaining low cost and compatibility with L i A B
181 conventional microwave hybrids and integrated circuit implementations. Fig. 5.8 illustrates the fabrication procedure, which is very similar to that used in the fabrication of the multilayer CRLH devices in Chapter 4. First, the bottom and top copper layers of the FR4 are etched away with a solution of H 2 SO 4 :H 2 O 2 :Deionized Water (1:1 :3). Standard substrate cleaning is then performed with TCE ( trichloroethylene), followed by rinse with Isopropanol, DI water and dehydration on top of a hot plate at 120 degrees for 10 minutes. Figure 7 8 Fabrication process. The first layer of SU8 2025, which is used as an interface layer in order to minimize the electromagnetic field interactions with the carrier substrate, is spin coated at 1400 RPM for 30 seconds to get an approximate t hickness of 50 m. Edge bead is further removed. Soft bake is performed on a leveled hot plate with the temperature ramped up to 65 C at a rate of 250 C/hour and kept for 30 min. Temperature is ramped up to 95 C at the same rate and kept for 20 minutes. Substrate SU8 Copper Ti/Cu seed layer Substrate SU8 SU8 Negative Photo Resist Electroplated copper Via interconnectio n Copper SU8 SU8 SU8 SU8 SU8 SU8
182 T h e samples are allowed to cool down to room temperature on the hot plate. Lithography with an MA6 Karl Suss Mask Aligner (UV wavelength of 365nm) is performed with an optical dose of 200 mJ/cm 2 Post exposure bake is performed at 65 C and 95 C on leveled ho t plate. The samples are allowed to cool down at room temperature, developed in SU8 developer for 30 minutes and finally rinsed with Isopropanol and blown dry with a nitrogen gun. A low temperature curing process of the SU8 is performed at 120 C during 1 0 minutes. Further, a metallic seed layer of Ti/Cu /Ti (30 nm / 300 m / 30 nm ) is sputtered on top of SU8 followed by negative photoresist patterning with NR4 8000p (Futurrex Inc.) for the embedded resonator and feeding lines. After the etching of the upper Titanium layer, Copper electroplating is performed for 5 m thickness, much thicker than the skin depth of copper at 12 GHz to minimize ohmic losses. At the end of the electroplating procedure, the Ti/Cu seed layer is sequentially removed. Ti is etched aw ay using diluted hydrofluoridic acid ( HF:DI water 1:10 ratio ). The second layer of 50 um SU8 2025 is then coated and the lithographical pattern of the interconnect vias and HMSIW via wall is performed. A second metallization of Ti/Cu /Ti seed layer is per formed using sputtering, followed by negative photoresist patterning with NR4 8000 for the upper CPW feeding lines and other devices. After copper electroplating up to 5 m, the seed layer is removed. Figure 7 9 A and Figure 7 9 B show scanning electron microscopy (SEM) images of the fabricated embedded resonator and filter without the top SU8 layer for presentation purpose. Figure 7 9 C shows the patterned and metallized vias on S U8 prior Copper electroplating Figure 7 9 D s hows the final fabricated filter with the top ground plane and the GCPW feeding lines already electr o plated. This fabrication process can be easily extended to the implementation of mor e SU8 layers by repeating the coating, lithography and electroplating procedures.
183 Figure 7 9 Scanning electron microscopy (SEM) images of the e mbedded resonator and filter. A ) The e mbedded resonator. B ) the embedded filt er without the second SU8 layer. C ) patte rned vias before electroplating. D ) final device after the electroplating of the top ground plane and GCPW feeding lines. 7.4. Measurement Results M easurement results of the fabricated res onator and filter are performed by using the setup described in Chapter 4 Figure 7 10 s hows the measurement results. I n Figure 7 10 A t he passband of t he reronator is obtained around 12 GHz as expected. Insertion losses of the r esonator and filter show good agreement with the sim ulated results, although the differences might be due to different values of the dielectric constant and the dielectric loss ta ngent of the SU8 as well as the tolerances of the fabrication process. It is believed that the design of the filter is not completely optimized, since its return loss shown in Figure 7 10 B is very low. This might A B C D
184 be caused by a low internal coupling coefficient between the coupled resonators. Additional work can be done in order to obtain a better response by optimizing the inter resonator distance and thus, the internal coupling coefficient. To highlight the advantages this m icromachined embedded approach offer, Table 7 3 shows a comparison in size of waveguide resonators at 12 GHz fabricated in different technologies. It is observed that more than 95% size reduction in comparison wit h the conventional SIW resonator and more than 80% size reduction in comparison with the conventional HMSIW evanescent mode resonator implemented on a PCB approach are obtained, which confirm that the micromachined approach is useful for the miniaturizatio n and 3D integration of passive devices at microwave frequencies and can be easily sacaled to higher frequencies of operation Table 7 3 Comparison of resonator size in different technologies Type a (mm) b (mm) d (mm) Area (a d) mm 2 Filled Waveguide WR 90 with r =3.4 22.8 10.1 00 7 159.6 SIW on RO 4003, Thickness =0.508mm 22.8 0.508 7 159.6 HMSIW evanescent mode resonator on RO4003, r =3.4 4 .0 0.508 3.5 14 .0 SU8 embedded HMSIW evanescent mo de resonator 1.4 0.1 00 1.4 1.96 7.5. Summary This chapter has presented the design, simulation and fabrication of compact multilayer micromachined, SU8 epoxy embedded evanescent wave half mode substrate integrated waveguide resonators and filters. A CSRR loaded on the top surface of the waveguide allows forward transmission below the waveguide cutoff frequency. Electromagnetic simulations show good agreement with measurement results as expected from the design specifications. The selection of SU8 as a dielectric layer, in despite of its relatively large loss, is based on good
185 thermal, mechanical and electrical properties and low cost. More than 80% size reduction, in comparison with the conventional PCB based devices is achieved for the resonator and filter. Although insertion loss seems to be large in comparison with conventional implementations, mainly due to the relatively large loss tangent of SU8, the surface micromachined process on SU8 still can be used for achieving compact circuits at microwa ve frequencies. The prototypes are fully fabricated and tested. The approach to implement embedded substrate integrated waveguide resonators and bandpass filters is suitable for System in Package or System in Substrate devices. Extentions of this work migh t include the use of BCB as a dielectric for implementing single and dual band micromachined embedded filters at millimeter wave frequencies. Figure 7 10 Measurement results A ) Insertion and retur n losses of the embedded resonator B ) Insertion and return losses of the embedded filter. A B
186 CHAPTER 8 EVANESCENT MODE BROADBAND BANDPASS FILTERS As previously studied, bandpass filters are key components of communication systems. During decades, special attention has been paid to the design of high performance, compact size and low cost bandpass filters using standard printed circuit board (PCB) and low temperature co fired ceramic (LTCC) processes . Recently, t he substrate integrated waveguid e (SIW) technique and its reduced mode versions that make use of fictitious magnetic walls, namely the half mode (HMSIW) and the quarter mode (QMSIW) substrate integrated waveguides, have been studied for the design of integrated planar filters with simila r performance to that of the conventional bulky waveguide filters [41 4 3 ]. The advantages of the SIW technologies include modest loss, high power handling capability and integrability with other planar technologies. Moreover, the QMSIW has been used for th e design of broadband filters with more than 5% bandwidth due to its reduced external quality ( Q e ) factor, which is not easily achievable with the original SIW [4 3 ]. On the other hand, with to the negative properties offered by metamaterials, i.e. negative permittivity and permeability, the complementary split ring resonator (CSRR) and the composite right /left handed (CRLH) technique have been combined with the SIW and HMSIW to realize bandpass filters working below the cutoff frequency of the dominant TE 10 mode [ 31, 4 5 46 ]. In this chapter, a new reduced mode version of the SIW, the eighth mode substrate integrated waveguide (EMSIW) cavity, which allows a size reduction of 87.5% with respect to the original SIW cavity, is proposed [ 101] In addition, the E MSIW is loaded with a CSRR on its top surface in order to obtain a new resonance frequency below the resonance frequency of its original quasi TE dominant mode; therefore, its size is further reduced. Moreover the CSRR loading technique is also applied to the QMSIW cavities for implementing broadband bandpass
187 filter s Section 8.1 studies the proposed EMSIW and CSRR loaded EMSIWcavities, in which the working principle and parameters of the cavities are provided and then, a set of two pole bandpass filters a re implemented on PCB and tested S ection 8.2 propos es the CSRR loaded QMSIW cavitiy for broadband filters working at quasi millimeter wave and millimeter wave frequencies. Two dielectric substrates are selected: Liquid crystal polymer (LCP) with a dielect ric constant of 2.9, and Benzocyclobutene (BCB) with a dielectric constant of 2.65. Surface micromachining techniques are u sed to implement these filters. 8.1. Broadband Bandpass Filters using CSRR Loaded Eighth Mode Substrate Integrated Waveguide Cavities Th e working principle and parameters of the EMSIW cavity are studie d theoretically and numerically. Also, a set of two pole bandpass filters is implemented by using the proposed cavities. The design concept is demonstrated with one direct fed filter and one filter with proximity feeding that allows the generation of two transmission zeros in the stop band. 8.1.1. General Requirements for Broadband Bandpass Filter Design Broadband bandpass filters, which are considered to have more than 5% fractional bandwidth , are also widely used in radio frequency engineering. Frequency bands such as Ultra Wide Band (UWB from 3.1 GHz to 10.6 GHz), ISM band from 57 GHz to 64 GHz, W band from 77 GHz to 110 GHz, WiFi applications from 5.25 GHz to 5.85 GHz, require not only narrow band filter s that cover part of the frequency rage, but also broadband filters that cover the entire band. In addition, modern technologies require low cost, low loss and integrable filters. Conventional designs using waveguide or SIW cavities are not sui table for broadband filter design due to their intrinsic high unloaded and loaded quality factors. The general requirements for broadband filter design are summarized as follows:
188 C avity resonators with a low external quality factor s which means more energ y is couple d from the source to the cavities and wider bandwidths are achievable High inter resonator s coupling coefficients, which means that the energy transfer between neighbor cavities is high indicating that wide bandwidths can be achieved. T he con ventional waveguide or SIW cavities cannot offer the requirements for broadband filters, mainly because of their big size s that feature high unloaded and loaded quality factor s and low internal coupling coefficients. The next sections demonstrate that the proposed EMSIW cavities in our study have lower achievable external quality factors and higher internal coupling coefficients than the conventional SIW cavities, indicating that they are useful for the design of broadband bandpass filters. 8.1.2. The Eighth Mode SIW Cavity Figure 8 1 A illustrates the electric field distribution in an original SIW cavity designed for 11 GHz. Four fictitious magnetic walls are shown for analysis purposes. Metallic solid side walls are used for simulation p urposes. All designs use the substrate Arlon DiClad 880 with a thickness of 0.787 mm and a dielectric constant r = 2.2. The components of the electric and magnetic field s in the original SIW cavity resonator for the TE 101 mode ( a = d ) are given by  : (8 1) (8 2) (8 3) and (8 4)
189 where and E 0 are the phase constant, the electrical permittivity and the magnetic permeability of the medium, and the amplitude of the electric field, respectively. The propagation constant is and the intrinsic impedance of the medium is [57 ]. Figure 8 1 The e ighth mode substrate integrated waveguide ( EMSIW). A ) TE 101 mode ( a = d ) E Field in the SIW cavity B ) E Fiel d in the proposed EMSIW cavity. C ) E Field in the prop osed CSRR loaded EMSIW at 8 GHz. D ) E Field i n the CSRR loaded EMSIW at 14 GHz. For quasi TE mode propagation and the existence of almost perfect magnetic walls in the reduced mode versions of the SIW, the thickness of the substrate, b must be much smaller than the width of the waveguide, a ( b < < a ). F rom Eq uations 8 3 and 8 4 it is observed in Figure C A D B a d a /2 A B C D d/2 x z y
190 8 1 A z = x x = a ( z = a x z = a /2 ). Then, by cutting the SIW cavity wit h the centered and diagonal magnetic walls, eight triangular cavities are created. As observed in Figure 8 1 B t he triangular cavity, named here the eighth mode SIW (EMSIW), supports the propagation of one eighth of the original T E 101 mode with its maximum intensity at the corner of the structure. In our study we call the propagating mode in the EMSIW the quasi TE 0.125,0,0.125 mode because it resembles the eighth part of the original TE 101 mode Therefore, the EMSIW features a re duced size with respect to the original SIW and can be useful for the design of compact bandpass filters. 8.1.3. The CSRR loaded Eighth Mode SIW Cavity To further reduce the size, a complementary split ring resonator (CSRR) is loaded on the top surface of the E MSIW, as shown in Figure 8 1 C and Figure 8 1 D. A single ring CSRR is chosen for simplicity Figure 8 1 C and Figure 8 1 D also show the two first resonan t modes in the CSRR loaded EMSIW designed for 8 GHz. The working principle is similar to that studied in Chapter 6 for the CSRR loaded HMSIW resonator, and it is based on the epsilon negative behavior of a rectangular waveguide under the cutoff frequency The lower mode at 8GHz, shown in Figure 8 1 C, is due to the resonant negative permittivity property of the CSRR loaded on the surface of the waveguide, which causes the resultant effective dielectric constant eff of the EMSIW to be positive and provides a frequency band with forward wave propagation below the original cavity resonance. T he original TE 101 like mode, shown in Figure 8 1 B for 11 GHz, is moved up to a frequency of 14 GHz due to the fact that the CSRR confines the electric field to a small area, as shown in Figure 8 1 D
191 Figure 8 2 Parame ters of the EMSIW cavities. A) External quality factors B) The layou t of the CSRR loaded EMSIW cavity. Physical dimensions are g = 0.35 mm, a =13 mm, and w f = 2.4 mm. 8.1.4. Resonator Analysis and Design A set of two pole filters using magnetically coupled CSRR loaded EMSIW cavities for operation at 8 GHz X band. The classic m ethodology for the design of coupled resonators filters is followed , in which the filter is determined by the external quality factor Q of the resonator and the internal coupling coefficient k between the coupled resonators. First, the external quali ty factor Q is analyzed. Figure 8 2 A shows the variation of the external Q factors with the offset of the feeding line for single loaded EMSIW and CSRR loaded EMSIW cavities. An array of metalized vias is used for the real implementation with a via diameter of 0.75 mm and a center to center pitch of 1.4 mm. The resonator can be excited by using one of the open sides. A direct connection of a 50 ed to excite the cavities. A square single split CSRR is loaded on the top surface with a slot width of 0.25 mm, a split gap of 0.2 mm and a side of 2.6 mm. As observed in Figure 8 2 A the Q factors of both cases are varied with the offset distance L e of the feeding. Since the EMSIW cavities are smaller than their original SIW counterparts, the available internal and external quality factors of the EMSIW are small, which is useful for broadband filters L e g a/2 w f A B
192 Fi gure 8 3 Characteri zation of the EMSIW cavities. A ) Internal coupling coefficients B ) Layout of the coupled EMSIW cavities. The internal coupling coefficients k of the coupled cavities with and without th e CSRR, which are controlled by a window of size W i in either side of the metalized via wall, are presented in Fi gure 8 3 A The method used in Chapter 6 to get the internal coupling coefficient is also used in this section in whi ch the cavities are excited with a weak external coupling that offers a high external quality factor. As observed, because of the small size of the cavities, higher coupling coefficients than those offered by the SIW and HMSIW cavities [41 43] are achievab le, which is also useful for broadband filters. 8.1.5. Two Pole Filters Designs The previous section has demonstrated that the proposed EMSIW cavities are suitable for broadband bandpass filter design, because of their low external quality factors and the high i nternal coupling coefficients. In this section, a set of two pole Chevyshev filters with 20 dB passband return loss is designed at 8 GHz for demonstration purposes. One direct fed filter with a fractional bandwidth of 7.5 % and one proximity fed filter wit h source load coupling  and a 10 % fractional bandwidth are implemented Figure 8 4 A and Figure 8 4 C show the layout of each filter. Standard PCB fabrication on a CNC milling machine is used. The side length of the A B W i
193 CSRR in the proximity fed filter is slightly decreased to 2.4 mm in order to compensate for the frequency shift due to the gap feeding  Table 8 1 summarizes the calculated parameters for the filter by following the design procedures in reference . Figure 8 4 Two pole f ilters. A) Direct fed filter. Geometrical parameters are : w i = 1.9 mm, L e = 2.5 mm. B ) Photograph of t he fabricated direct fe d filter. C ) Gap fed filter. Geometrical parameters are: w t = w c = 0.3 mm, g c = g t = 0.4 mm, l c = 6 mm w i = 2.56 mm. D ) Photograph of the fabricated gap fed filter Table 8 1 Specification and calculated parameters of the filters Filter FBW/ Return Loss External Q Factor Internal Coupling Coefficient k Direct feeding 7.5%, 20 dB 8.865 0.1247 Gap feeding 10%, 20 dB 6.648 0.1662 w i L e A B C w c w t g c g t l c w i D
194 8.1.6. Results and Discussion The fabricated filters are shown in Figure 8 4 B and Figure 8 4 B. Measurements are performed with a vector network analyzer HP 8719D after standard SOLT (short open load thru) calibration The effects of the connectors have not been de embedded, which can cause additional loss Figure 8 5 A and Figure 8 5 B show the frequency responses for the direct and proximity fed filters, respectively. The tot al area of both filters excluding the feeding line s, is 0.34 0 0.09 4 0 w ith 0 as the free space wavelength at 8 GHz. Measured center frequencies of 7.92 GHz with a 3 dB bandwidth of 17.16 %, and 7.9 GHz with a 3 dB bandwidth of 22.28 % are obtaine d for the direct and proximity fed filters, respectively. The measured in band insertion and return loss are better than 1.5 dB and 13 dB for both filters. Measurement results agree well with simulated one. Minute differences are observed mainly due to the fabrication tolerances, especially the fabrication of the gaps, and connector loss. It is observed that the out of band rejection is better than 15 dB and that the second order mode is suppressed in both cases. 8.1.7. Summary Eighth mode substrate integrated wa veguide (EMSIW) cavities are proposed and studied for broadband filter design. The EMSIW features a reduced size in comparison with the original SIW. Moreover, conventional implementations of two pole filters on microstrip show larger size than that achiev ed in this work. The size is further reduced by using CSRR loaded EMSIW cavities. On the other hand, the size can also be reduced if the split gap and the ring width of the CSRR are reduced, which were limited by the fabrication technology used in this wor k. The Q factors and internal coupling coefficients have been studied in detail. The concept has been demonstrated by implementing a set of CSRR loaded EMSIW two pole filters. Higher order filters and different coupling schemes could be also implemented.
195 Figure 8 5 Measured and simulated results. A) Direct fed filter B ) Gap fed filter 8.2. Surface Micromachined Broadband Millimeter Wa ve Bandpass Filters Using CSRR L oaded Quarter Mode Substrate Integrated Waveguide Cavities The previous section proposed the eighth mode substrate integrated (EMSIW) waveguide cavity for broadband bandpass filter design, which features a size reduction of 87.5% with respect to the original substrate integrated waveguide (SIW) cavity. In this section one more step A B
196 toward the miniaturization of broadband bandpass filters is taken. Due to the limits imposed by the standard printed circuit board fabrication technology for implementing devices working at millimeter wave frequencies, the use of surface micromachining techniques is proposed for the implementation of compact broadband filters working at millimeter wave frequencies. For this work, the quarter mode substrate integrated waveguide (QMSIW) loaded with a single ring complementary split ring resonator (CSRR) is selected as the waveguiding structure. It is also demonstrated that the CSRR loaded QMSIW is able to be excited with low external quality factor and that high inter resonator coupling coeffici ents can be achieved, which is useful for broadband filter design. On the other hand, due to their good performance at microwave and millimeter wave frequencies, two dielectric materials have been selected for the implementation of the filters : The flexi ble liquid crystal polymer (LCP) and the photopaternable Benzocyclobutene resin (BCB). Flexible LCP Ultralam 3850 from Rogers Duroid with a thickness of 4 mil (101.6 m), a relative dielectric constant r = 2.9 and a loss tangent of 0.002 is selected for the implementation of two pole and three pole QMSIW filters working at 25 GHz GHz. In order to achieve higher frequencies of operation, BCB with a thickness of 21 m is used to implement two and four pole filters working at 60 GHz. At first, the QMSIW and t he CSRR loaded QMSIW cavities are introduced Nezt, the design and simulation of two and three pole CSRR loaded QMSIW filters using LCP as a dielectric material are presented The filters are designed for operation at 25 GHz. The fabrication process is al so studied. Finally, the design and simulation of two a nd four poles filters using BCB as a dielectric are studied. The fabrication of the filters is not done at this time therefore only design concept and simulation results are presented at this time. H owever, the fabrication and measurement are left as a future work of this study.
197 8.2.1. The CSRR loaded Quarter Mode Substrate Integrated Waveguide Cavities The original SIW cavity shown in Figure 8 6 A, previously shown in section 8.1, i s recalled in this section for clarity purposes. In addition, the electric field distribution on two different versions of the QMSIW cavity are also presented in Figure 8 6 B and Figure 8 6 C. As observed, the QMSIW cavities feature one fourth of the size of the original SIW cavity while allowing the wave propagation in a mode that resembles one fourth of the original TE 101 mode of the SIW, called here the quasi TE 0.25,0,0.25 mode. The fictitious magnetic w alls can be used to get square and triangular shape QMSIW cavities. On the other hand, when the CSRR is loaded on the top surface of the QMSIW, as shown in Figure 8 6 D, the electric field is concentrated around the CSRR and the propagating mode is completely different to that of the original SIW and QMSIW cavities. Moreover, the CSRR loading creates a new resonance frequency below the original resonance of the QMSIW cavity, which also has the same working principle previously st udied for the CSRR loaded EMSIW cavity. Similar results are obtained when a CSRR is also loaded on the triangular shape QMSIW cavity, although not shown here. In order to demonstrate the design concept, the next two sections present the design and simulati on of the two, three and four pole filters using the proposed CSRR loaded rectangular QMSIW cavities. 8.2.2. The Design of CSRR loaded QMSIW Bandpass Filters on Flexible LCP Flexible liquid crystal polymer (LCP) Ultralam 3850 from Rogers Duroid, with a thicknes s of 4 mil (101.6 m) is selected as the dielectric material for implementing broadband filters working at 25 GHz frequency band. For broadband design, the fractional bandwidth (FBW) is selected to be higher than 5%. For demonstration purposes, two pole an d three pole filters were designed and simulated. Table 8 2 summarizes the design specifications of the proposed filters.
198 The fabrication procedure is proposed based on the recommended processing guidelines given by Rogers Inc. for the liqui d crystal polymer substrate [102 ]. Table 8 2 Design specification for the filters on LCP Filter f o (GHz) FBW Passband Return Loss QMSIW Cavity Two poles 25.3 11% 20 dB Rectangular Three poles 25.3 16% 20 dB Rectangular Figure 8 6 The quarter mode substrate integrated waveguide (QMSIW) cavities. A) Electric field (E field) distribution in the original SIW cavity f or TE 101 mode. B) The rectangular QMSIW cavity. C) The triangular QMSIW cavity. D ) The proposed CSRR loaded QMSIW cavity. C A D B a d a/2 A B d/2 x z y a d/2 C D
199 188.8.131.52 Resonator study The filter design methodology explained in previous sections is used in this section . The first step is th e characterization of the resonator, from which the achievable external quality f actors ( Q e ) and internal coupling coefficients ( k ) are obtained. The inset in Figure 8 7 shows the designed resonator to work around 25 GHz. The ori ginal frequency of the QMSIW cavity with no CSRR loading is 48 GHz, which means that a size reduction of nearly 50% is achieved by the CSRR loading. Four different orientations of the CSRR are possible, while the one shown in Figure 8 7 has been chosen for this design. The external quality factor is controlled by the offset distance L q and also the offset position l o of the CSRR in the cavity. However, the offset position l o is selected to be 0.2 mm and is kept constant. Metalize d vias with a diameter of 0.2 mm and a center to center pitch of 0.3125 mm create the metallic sidewalls. As observed, external quality factors as low as 6 can be obtained. Next, the internal coupling coefficient between magnetically coupled cavities is s tudied. The inset in Figure 8 8 shows the used configuration for such purpose. The cavities are excited with a high external Q factor feeding in order to have a weak coupling from the source. The internal coupling coefficient is c ontrolled by the window in the via row W i The extraction methodology explained in section 184.108.40.206 is used in this section. Equation 6 17 is used to get the coupling coefficient. Figure 8 8 shows the obtained results. As observed, internal coupling coefficients higher than 0.15 are possible, indicating the CSRR loaded QMSIW cavity is useful for broadband filter applications.
200 Figure 8 7 Extracted external quality fact or ( Q e ) of the resonator. The inset shows the configuration of the QMSIW cavity. Geometrical parameters are : l o = 0.2 mm, c = 0.1 mm, g = 0.15 mm, w = 1.35 mm. Figure 8 8 Extracted in ternal coupling coefficient k The inset shows the configuration of the magnetically coupled cavities. g c l o l o L q w W i
201 Finally, Figure 8 9 A shows the simulated frequency response of the return loss of a single ended CSRR loaded HMSIW cavity. The phase shift and additional loss provided by the feeding line are de embedded. Figure 8 9 B shows the phase angle of the return loss (S 11 ). These two plots are used to explain the procedure used in this research to extract the exter nal quality factor ( Q e ) of a single loaded cavity, which is explained in detail in reference . Since the insertion loss is not available, the extraction is not based on obtaining the 3 dB bandwidth as previously explained is section 220.127.116.11, but on the analysis of the phase angle of the return loss . For this purpose, the resonance frequency of the cavity, f o is obtained from the return loss plot and two frequency points, f L and f H are extracted from the phase plot when the phase angles are 9 0 degrees, as observed in Figure 8 9 B. Then, the external quality factor Q e is calculated as in (8 4) For this case in particular, f o = 25.47 GHz, f H = 25.67 GHz and f L = 24.56 GHz, which results in Q e = 22.84. Figure 8 9 Simulated frequency response of a single ended CSRR loaded QMSIW cavity. A) Return loss. B ) Phase angle. A B
202 18.104.22.168 Bandpass filter designs and simulations Based on the design specifi cations and the previous resonator study, two pole and three pole bandpass filters are designed. Table 8 3 summarizes the calculated design parameters . The initial physical dimensions of the filter are selecte d based on the resonator study and optimized through full wave structure simulations in HFSS. Figure 8 10 s hows the physical configuration of the two and three pole filters. Since the filters are to be tested with a Cascade Microt ech probe station with GSG ( ground signal g round) probes and a picth distance of 150 m, grounded coplanar waveguide (GCPW) launching feeding lines are used. The thickness of the metal trace (18 m of Copper) and the dielectric loss tangent of the LCP (tan = 0.002) have been taken into account for the simulations. In Figure 8 10 B t he CSRR in the middle of the three pole filter is scaled up with a facto r of 1.122 in order to optimize the frequency response. Table 8 3 Design parameters of the filters on LCP Filter f 0 FBW Return Loss Q e (1,2) k 12 k 23 Two poles 25.3 11% 20 dB 6.044 0.1828 Three Poles 25.3 16% 20 dB 5.32 0 0.165 0 0.165 Figure 8 10 Physical layout of the bandpass filters on LCP. A ) Two pole filter. Geometrical parameters are : w = 1.3 mm, g = 0.18 mm, W i = 0.51 mm, l x = l y = 0.9mm, l o = 0.18 mm, L q = 0. B ) Three pole filter. Geometrical paramet ers are the same for the two pole filter except for : W i = 0.51 mm, l y2 = 1.01 mm. W i W i W i A B
203 Figure 8 11 shows the simulated results for the designed two pole filter. A maximum insertion loss of 0.51 dB is expected within the passband. Also, a fractional bandwid th of 11.7% for a 20 dB return loss is obtained around a resonance frequency of 25.5 GHz. The simulation results agree well with the design specifications. The size of the filter is only 0.22 0 0.11 0 where 0 is the free space wavelength at 25.5 GHz. A 3dB bandwidth is obtained from 22 GHz up to 29 GHz, which covers the entire 24 GHz automotive band. The out of band rejection is better than 15 dB at 32 GHz, which is 10% higher than the 3 dB bandwidth upper frequency. The effects of the feeding lines hav e not been extracted, which indicates that the real insertion loss of the filter is lower. Figure 8 11 Simulated results for the two pole bandpass filter on LCP. The simulated resonance fre quency is 25.5 GHz. The obtained 20 dB return loss fractional bandwidth is 11.7%. Les s than 0.51dB insertion loss is expected within the 20 dB passband. Figure 8 12 shows the simulated results for the three pole filter. As observ ed, a clear three pole response is obtained. A maximum insertion loss of 0.69 dB is expected within the
204 passband. A fractional bandwidth of 17% for a 20 dB return loss is obtained around a resonance frequency of 25.7 GHz. The simulation results agree well with the design specifications. The size of the filter is only 0.33 0 0.11 0 where 0 is the free space wavelength at 25.7GHz. A 3dB bandwidth is obtained from 22.32 GHz up to 28.57GHz. The out of band rejection is better than 15dB at 30 GHz, which is 5% higher than the 3 dB bandwidth upper frequency. The effects of the feeding lines have not been extracted; therefore, the real insertion loss of the filter might be less Figure 8 12 Sim ulated results for the three pole bandpass filter on LCP. The simulated resonance frequency is 25.7 GHz. The obtained 20 dB return loss fractional bandwidth is 17%. Les than 0.68 dB insertion loss is expected within the 20 dB passband. 22.214.171.124 Proposed fabricatio n procedure on LCP The proposed fabrication of the bandpass filters is based on a surface micromachining process on the liquid crystal polymer (LCP) substrate Rogers Ultralam 3850 with a thickness of
205 4 mil and a dielectric constant of 2.9. Since only one l ayer of LCP is used in this project, the fabrication starts with the etching of the top Copper layer of a double clad LCP sheet. Then, mechanical drilling of the via holes, desmear of the vias in a heated ultrasonic bath, oxygen plasma cleaning, metal depo sition, lithographical pattern and electroplating are used for fabricating the devices. Figure 8 13 summarizes the proposed fabrication process. Figure 8 13 Proposed f abric ation process of the LCP filters. 8.2.3. The Design of CSRR loaded QMSIW Bandpass Filters on BCB Although LCP has good properties for the implementation of microwave and millimeter wave applications, as the frequency increases the integration with CMOS cir cuitry and processes becomes more important. Bulk and surface micromachining techniques offer an excellent way to fabricate small devices for millimeter wave applications, but some steps might require expensive equipment and processes in order to continue fabricating smaller features on non photosensitive Substrate: 4 mil LCP Rogers U ltralam 3850. Top Copper is removed Vias : CNC drilling of the vias and alignment marks. Via desmear and substrate cleaning 1 minute Oxygen plasma cleaning and DC sputtering of Ti/Cu/Ti seed layer (30nm/300nm/30nm) Lithography patterning of the Copp er layer with NR9 8000 negative resist. 18 m Copper electroplating. Etch seed layer LCP Copper Ti/Cu/Ti seed layer NR9 8000 Photomask Electroplated copper
206 dielectric materials, such as laser or deep RIE for through substrate via drilling. To overcome these limitations, Benzocyclobutene (BCB, Cyclotene 4026 46 from Dow Chemical) with a thickness of 21 m on C MOS grade low resistivity Silicon or glass is proposed as the dielectric material for implementing surface micromachined broadband filters working at the unlicensed 57 GHz to 6 4 GHz frequency band. Different work has previously demonstrated micromachined na rrow band waveguide cavity filters working at millimeter wave frequencies by using comp licated elevated structures [ 103 1 04 ] or multilayers of photoimageable thick pastes [105 ], which might not be completely compatible with conventional CMOS/MMIC/MEMS pro cesses. However, no much work has been reported on wideband bandpass filters working with miniaturized cavities that make use of metamaterial concepts. We believed that the use of metamaterial concepts in combination with a post CMOS micromachined process using properly selected low loss polymers for interfacial layer is useful for implementing low profile and low cost integrable filters for millimeter wave applications. Since the BCB is a photosensitive resin with multilayer capabilities, thicker thicknes s might be used in order to increase the quality factors of the resonators. In our study a single coated layer of 21 m of BCB is used for demonstration purposes. The 3D cross section view of the proposed filters was previously introduced in Chapter 3. T he thickness is achieved based on a modified coating and processing of the BCB. The Appendix gives the material specifications and the fabrication procedures. For demonstration purposes, two pole and fo ur pole filters have been designed and simulated. Table 8 4 summarizes the design specifications and calculated parameters of the proposed filters. Table 8 4 Design specifications and calculated parameters of the filters on BCB Filter f 0 FBW Return Loss Q e (1,2) k 12 k 23 k 34 Two poles 60.5 10% 20 dB 6.704 0.1648 Four Poles 60.5 10% 20 dB 9.392 0.0904 0.069 5 0.0904
207 Figure 8 14 shows th e extracted external quality factors for a CSRR loaded QMSIW cavity resonator working at 60.5 GHz. External quality factors as low as 5.6 can be obtained, which is useful for broadband filters. The inset of Figure 8 14 shows the layout of the cavity resonator. S ince a micromachining process is to be used, there is no necessity to follow the conventional SIW topology with a metalized via row. Therefore, a complete metalized wall with a width of 80 m is used for implementing the sidewalls of the cavity. The square CSRR has a side length of 0.4 mm, a ring gap of 0.1 mm and a ring width of 60 m. The side length of the square cavity is 0.6 mm. The resonance frequency of the original QMSIW cavity with no CSRR loading is 102 GHz, wh ich indicates a size reduction of more than 40% when the CSRR is loaded on the cavity. The size can be further reduced if the width of the ring of the CSRR is selected smaller Figure 8 14 Extracted external quality factor of a CSRR loaded QMSIW cavity on BCB. The internal coupling coefficient between magnetically coupled cavities is presented in Figure 8 15 where the inset shows the coupled cavities with a high external qu ality factor excitation As observed, due to the small size of the cavities, internal coupling coefficients higher than 0. 15 are also obtained, which makes the cavities on BCB are useful for broadband filter s L q w l y
208 Figure 8 15 Extracted internal coupling coefficient for magnetically coupled CSRR loaded QMSIW cavities on BCB. Figure 8 16 Layout of the proposed filters. A) The two pole filter Geometrical parameters are: w = 0.6 mm, l y = 0.387 mm, L q = 0.13 mm, window = 0.33mm. B ) The four pole filter. Geometrical parameters are: w 12 = w 24 = 0.268 mm, w 23 = 0.389 mm, L q = 0.5 mm, l y = 0.392 mm, l x = 0.3528 mm, l cx = 0.54 mm. Window 1 2 3 4 l cx l x L q w 12 w 34 w 23 B Window A
209 126.96.36.199 Bandpass filter on BCB designs and simulations Based on the design specifications a two pole and a four pole bandpass filters are designed. The physical dimensions of the filters are optimized through full wave structure simulations in HFSS Figure 8 16 shows the physical configuration of the two and four pole filters. Grounded coplanar waveguide 50 launching feeding lines with a width of 56.8 m and a CPW gap of 35 m are used, taking into account that GSG probes with a 150 m picth will be used for characterization. The thickness of the metal trace is selected to be 5 m of electroplated Copper, which is much larger than the skin depth at 60.5 GHz (0.266 m). In Figure 8 16 B the CSRR length l x and ca vity length l cx of the second and third cavities have been scaled down with a factor of 0.9 in order to optimize the frequency response. Figure 8 17 Simulated frequency response of the two pole 6 0GHz bandpass filter on BCB. Figure 8 17 s hows the simulated results for the two pole filter at 60 GHz. A maximum insertion loss of 1.13 dB is expected within the passband. A fractional bandwidth of 10% for a 20 dB return loss is obtained around a resonance frequency of 60.5 GHz. The simulation results
210 agree well with the design specifications. The size of the filter is only 0.258 0 0.135 0 where 0 is the free space wavelength at 60.5 GHz. A 3 dB bandwidth is obtained from 53.17 GHz up to 67.8GHz, which covers the entire unlicensed 57 GHz to 64 GHz frequency band. The effects of the feeding lines have not been extracted, which indicates that the real insertion loss of the filter is lower. Figure 8 18 shows the simulated results for the four pole filter on BCB. A maximum insertion loss of 1.99 dB is expected within the passband. A fractional bandwidth of 11% for a 20 dB return loss is obtained around a resonance frequency of 60.5 GHz. The simulation results agree well with the design specifications due to the undertaken optimization process based on full wave structure simulations. The size of the filter is only 0.258 0 0.246 0 where 0 is the free space wavelength at 60.5 GHz. A 3 dB bandwidth is obtained from 5 4.74 GHz to 65.7GHz, which also covers the entire 57 GHz to 64 GHz unlicensed frequency band. The out of band rejection is better than 20 dB at 69.07 GHz, which is 5% higher than the 3 dB bandwidth upper frequency. The effects of the feedin g lines have not been extracted. Figure 8 18 Simulated frequency response of the four pole 60 GHz bandpass filter on BCB.
211 188.8.131.52 Fabrication status The surface micromachined pro cess on LCP and BCB is com pletely tested. Due to time constrains and photo mask preparation, the fabrication and measurements of the QMSIW filters is not finished at this time however it is left as a future work of this study 8.2.4. Summary This section has proposed the implementation of surface micromachined microwave and millimeter wave bandpass filters using the quarter mode substrate integrated waveguide (QMSIW) cavity loaded with a complementary split ring resonator (CSRR). Since the conventional QMSIW cavity already offers a size re duction of 75% with respect to the original SIW cavity, the resonators and filters using the QMSIW feature a great size reduction. Moreover, since the CSRR loaded QMSIW cavity works under the same evanescent mode principle explained in previous sections, i t was demonstrated that its size can be further reduced more than 40% with respect the conventional QMSIW cavity with no CSRR loading. Then, the CSRR loaded QMSIW cavity offers an extraordinary size reduction of nearly 90% with respect to the original SIW cavity. In addition, its small size offers the possibility of obtaining lower external quality factors and higher internal coupling coefficient between coupled stages than those achievable with the original SIW, which are useful for broadband filter design The design and simulation of the proposed filters were studied in detail. Moreover, LCP and BCB have been proposed to be used as the dielectric material, which keeps compatibility with conventional planar technologies and CMOS/MEMS/MMIC processes. Due t o time constrains the full fabrication of the filters is not available at this time
212 CHAPTER 9 CONCLUSIONS The main objective of this re search was to propose new concepts and architectures for implementing metamaterials circuits for microwave and mil limeter wave applications. A great effort was put into the fabrication of compact metamaterial devices by using surface micromachining techniques. Different dielectrics were evaluated to implement the proposed architectures and the performance of the devic es was compared with theoretical and simulated data. The first two parts of our study presented a comprehensive review of the metamaterial concepts and applications to microwave and millimeter wave frequency range. Next, a review of micromachined conventi onal right handed and me tamaterial transmission lines was provided in order to analyze the technical challenges when using micromachining techniques. Based on the literature review, the goals and objectives of our study were proposed, as well as the 3D bas ic structures that were used for implementing the proposed devices. As a first approximation to demonstrate new concepts and techniques, comprehen sive theoretical analysis, full wave structure simulations and conventional printed circuit bo ard (PCB) fabri cation were used. Single and multiband applications that make use of metamaterial concepts were demonstrated. Then, a surface micromachined process that combines different materials, such as SU8, BCB and LCP, was proposed for implementing compact multilaye r devices working at microwave and millimeter wave frequencies. In C hapter s 4 and 5 m ultilayer metamaterial unit cells for broadband and multiband opera tions at microwave and millimeter wave frequencies were demonstrated. The use of photosensitive SU8 polymer and BCB resin allow a great size reduction when compared with conventional PCB implementations. Mor eover, a new m etamaterial concept was proposed an d
213 fully demonstrated : T he bridged CRLH unit cell. The all pass behavior of the unit cell was demonstrated throug h extensive theory analysis, numerical simulations and experimental verification In addition the non linear dispersion of the unit cell makes it useful for multiband metamaterial applications as it was demonstrated with a triband open stub implementation. In the same way, bandpass filters that make use of newly proposed in substrate waveguide cavities were demonstrated and analyzed. At first, t he eight mode substrate integr ated waveguide (EMSIW) cavity was originally introduced in our study. Then, a metamaterial particle, i.e. the complementary split ring resonator (CSRR), was combined with the proposed cavity in order to achieve a great size re duction. Meanwhile the CSRR loaded quarter mode substrate integrated waveguide (QMSIW) was also proposed f or bandpass filter design. It was demonstrated that, due to their reduced size, these cavities are useful for the design of wideband filters at micro wave and millimeter wave frequencies. To demonstrate the concept, a set of two po le broadband bandpass filters for operation at 8 GHz was fully implemented and tested. Also a set of two, three and four p ole wideband bandpass filters was proposed for micro wave and millimeter wave operations. The behavior of the filters was analyzed theoretical ly and numerically. Finally, the procedure for implementing the filters by us ing micromachining techniques was proposed while the full physi cal implementation is left as future work of our research. The r esults show that the use of SU8 BCB and LCP as dielectric material s allows a great degree of miniaturization when compared with conventional PCB based implementations. Moreover, by using metamaterial concepts, for the first time compact broadband bandpass filters can be easily implemented by using miniaturized in substrate integrated waveguide cavities that make use of reduced mode versions of the original substrate integrated waveguide. In addition,
214 the possibility of devices with multiple dielectric layers was also explored in order to achieve 3D integrability and vertical interconnection of passive components which lead to compactness suitable for Systems on Package (SoP) or Systems on Substrate (SoS). 9.1. Summary of R esearch Contributions The re search con tributions of our work includes a systematic methodology for the implementation of micromachined grounded c oplanar w aveguide ( G CPW) and f inite g round c oplanar w aveguide (FGC) balanced CRLH transmission lines and applic ations on low resistivity silicon and organic substrates by using SU8 and BCB as dielectric interface layers; t he complete fabrication process for implementing micromachined multilayer metamaterial circuits for microwave and millimeter wave oper ations; t he optimized design and modeling of integrated and embedded passive components such as metal insulator metal (MIM) capacitors, meander line inductors and complementary split ring resonators (CSRR) based devices for implementing multilayer metamaterial applic ations ; a comprehensive theoretical and numerical analysis of a modifi ed design of the conventional CRLH unit cell, which we called the Bridged CRLH, able to achieve all pass behavior while keepi ng the multiple band operation; t he implementation of both na rrow band and wideband bandpass filters by using newly proposed compact in substrate waveguide cavities that make use of metamaterial concepts in combination with the reduced mode operation of the original substrate integrated waveguide; t he complete desig n and simulation of wideband bandpass filters working at microwave and millimeter wave, and the proposed fabrication procedure bas ed on micromachining techniques; d emonstration of the design concept, 3D structure and f abrication process to implement microm achi ne d vertically integrated e vanescent mode resonator s and filter s fo r compact wireless applications. 9.2. Future Work Our research is an exploration of micromachining techniques and new concepts to implement compact metamaterial devices working at microwave and millime ter wave frequencies. Due to their nature, all the proposed devices are not able to be implemented in a
215 given timeframe. However, the fabrication procedures and new proposed concepts in our work can be of interest for future research projects. On the CRLH metamaterial applications, the proposed Bridged CRLH can be a good candidate for the demonstration of all pass and multiband metamaterial unit cells working f rom very low frequencies, close to DC, and covering a very wide frequency range. Moreo ver, its multiband behavior makes it useful for the implementation of multiband applications such as power dividers, couplers and antennas. Following the same trend, the micromachined process on BCB and SU8 for implementing CRLH devices can be further opti mized in order to reduce loss and be used for demonstrating compact, 3D integrable and CMOS compatible metamaterial circuits for millimeter wave operation. In addition, the same unit cell structure can be used to fully implement CRLH applications using con ventional 0.18 m or 90 nm conventional CMOS processes. On the other hand, the proposed reduced mode versions of the SIW, working under the principle of evanescent wave amplification, are useful for the design of wideband filters. At first, the proposed f ilters on LCP and BCB can be fully demonstrated. Then, the cavities can be used to implement higher order filters and new architectures such as elliptic filters, cross coupled filters and dual band filters that make use of the extracted pole technique  In the same way, the fabrication process on BCB can be modified in order to achieve a higher thickness with one single coating that allows implementing higher quality factor resonators. Finally, tunable applications can also be implemented by combining the proposed structures and fabrication process in our research with ferroelectric materials such as Barium
216 Strontium Titanate (BST), for which a sol gel process for fabricating tunable devices was developed in our research group. 9.3. List of Related Publicati ons Journals publications: X. Cheng D.E Senior C Kim and Y .K. Yoon, A Compact Omnidirectional Self Packaged Patch Antenna With Complementary Split Ring Resonator Loading for Wireless Endoscope Applications, IEEE Antennas and Wireless Propagat Let t. vol.10, pp.1532 1535, 2011. D.E. Senior X Cheng and Y .K. Yoon, Electrically Tunable Evanescent Mode Half Mode Substrate Integrated Waveguide Resonators, IEEE Microw. Wireless Compon Lett vol.22, no.3, pp.123 125, March 2012. C Kim, X Cheng, D E. Senior and Y .K. tunable stopband filters using split ring resonators and varactors coupled transmission AEU International Journal of Electronics and Communications Available online 3 April 2012, ISSN 1434 8411, 10.1016/j.aeue.2012.03.004. In press. D. E. Senior and Bridged Composite Right/Left Handed Unit Cell with All Pass and Triple Band Response IEEE Microw Wireless Compon Lett IEEE In review. D.E. Senior X Cheng and Y .K. Yoon Broadband Bandpass Filters using Complementary Split Ring Resonator Loaded Eighth Mode Substrate Integrated Waveguide IEEE Microw Wireless Compon Lett In review. Conference publications: D. S. Elles and Y. K. Yoon ey polygon power divider using composite right/left in IEEE MTT S Int. Microw. Symp. Dig. Jun 2009, pp. 485 488 D. E. Senior X. Cheng, M. Machado Filters Using Complemen tary Split Ring Resonator Loaded Half Mode Substrate in Proc IEEE AP S USNC/URSI National Radio Science Meeting Jul. 2020, pp.1 4. D.E. Senior X. Cheng, P. Jao, C. Kim Compact 3D integrable SU8 embedded microwave bandpass filters using complementary split ring resonator loaded half mode substrate integrated waveguide, in Proc. IEEE Electronic Compon T ech Conf Jun. 2011 pp.1963 1969 D. E. Senior Me tamaterial Circuits using Multilayers of Low Loss Benzocyclobutene for Microwave
217 IEEE Electronic Compon Tech. Conf ., June 2012, San Diego, California. Proceedings in press D. E. Senior, Du al Band Filters Using Complementary Split Ring Resonator and Capacitive Loaded Half Mode Substrate Integrated Waveguide IEEE AP S USNC/URSI National Radio Science Meeting Jul. 2012, Chicago, Illinois Proceeding in press. D. E. Senior and Y. K. Yoo Waveguide as an Epsilon Negative Transmission Line Proc IEEE AP S USNC/URSI National Radio Science Meeting Jul. 2012, Chicago, Illinois. Proceeding in press.
218 AP P ENDIX MIC ROMACHINED FABRICATION PROCEDURES ON BENZOCYCLOBUTENE Benzoclyclobutene (BCB) is a completely different chemical from SU8, which was the first material to be used in our research. When comparing BCB with SU8, the different versions of SU8 require a simpler fabrication procedure. Therefore, the previous fabrication experience using SU8, which was previously used to implement the CRLH and HMSIW device s serves as a reference to proposed and characterize a fabrication procedure using BCB. The BCB to be used in our research is the electronic resin Cyclotene 4026 46 from DOW Inc. The BCB shows a dielectric constant r = 2. 6 5 at 10 GHz, and a loss tangent = 0.002 at 10 GHz. The first factor to take into account is the design of the fabrication masks. Cycloten e 4026 46 is a negative tone photo resin, which means that the unexposed areas will be washed away. On the other hand, the use of an a dhesion promoter is recommended, especially when the BCB is to be coated onto metallic surfaces. The adhesion promoter AP3000 from Dow Inc. is used in our work. In addition, a baking step of 30 seconds at 100 150 C is recommended to improve adhesion on Copper. Although Dow Inc., offers a recommended fabrication procedure, based on our own experience we provide here some g uidelines for BCB processing. Three different BCB thicknesses have been selected for these guidelines: 7 m, 14 m and 21 m The procedure is as follows : a ) Adhesion promoter : Always use adhesion promoter. It is normally recommended for metallic surfa ces, but based on our experience, good results are obtained if the adhesion promoter is also used on glass and silicon. The adhesion promoter should be applied onto the sample and then a speed of 3000 RPM is used to spin the sample during 15 seconds until a color pattern in the sample disappear s For better results, bake the sample on a hot plate at 110 C for 30 seconds to 40 seconds.
219 b). Coating : BCB is spun on the substrate for 30 seconds. Open bowl should be used. Spin Speed (RPM) Thickness after Sof t Bake ( m) Final Thickness ( m) 850 to 870 27 to 26 22 to 21 15 5 0 18.5 14 4200 8.1 7 c ). Edge bead removal : Remove edge bead and clean back side. It is m anually done. d ). Soft bake: 90 seconds. This work uses an end point mon itor in which a dummy wafer is used in order to determine the developing time. Pre exposure Thickness (um) Temperature ( C) < 10 8 5 10 to 18.5 100 >18.5 1 05 e ). Exposure : It is a negativ e resist. Cool down substrate to room temperature before exposure Use a leveled surface for this purpose For I Line broad band with a proximity/contact aligner the exposure dose is 60mJ/cm 2 per m of thickness. This value is only for Silicon. Higher expo sure dose is needed for multilayer devices and BCB on metallic surfaces Use 1.5x factor when using BCB on a metallic surface. Also, when fabricating vias on BCB, use a 1.2x to 1.5x factor. f ). Develop : Two steps developing process : Immersion in heated DS3000 at 30 40 C. The use of a temperature controlled tank is recommended. In our experience, 35 C is a good temperature. The developing time should be 150 to 200% of the developing time of the monitor dummy wafer. The dummy wafer is processed with the same coating and baking conditions, but it is not exposed.
220 Rinse with DS3000 at room temperature during 1 or 2 minutes in order to stop the developing process and dilute some BCB loaded developer. g ). Clean wafers : Clean the wafer with DI water spray. Spin dry at 2000 RPM during 1 or 2 minutes. Blow dry with a nitrogen gun. h ). Post develop bake : It is not necessary, bu t 1 minute at 90 C is recommended if the sample is to be observed under a microscope. i ). Cure : Under inert atmosphere. Vacuum oven i s used. Two types of cure : Soft cure : For lower BCB layers. It i mproves adhesion. Hard cure : For the last layer or for only one layer devices. Table A 1 Curing of BCB Step Soft Hard 1 15 min ramp to 150 C 15 min ram p to 150 C 2 15 min soak at 150 C 15 min soak at 150 C 3 ramp to 210 C ramp to 250 C 4 40 min at 210 C 60 min at 210 C 5 Cool down to less than 150 C Cool down to less than 150 C j ). DESCUM : Brief exposure to plasma to remove 1000 to 2000 A of poly mer residue O 2 :SF 6 gas composition of 90:10 at 10mTorr is used
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230 BIOGRAPHICAL SKETCH David Eliecer Senior Elles was born in Cartagena Colombia in 1977. He received his B achelo r of S cience d egree in electronic e ngineering from the Universidad Tecnol gica de Bol var Cartagena, Colombia in 2001 a M aster of Engineering d egree in e lectronics and computer e ngineering from Universidad de los Andes Bogota, Colombia in 2005 and a Ma ster of Science degree in e lectrical e ngineering from University at Buffalo, Buffalo, New York in 2010. He received a Fulbright scholarship in 2007 for pursuing graduate studies in the United States. He has authored and co authored more than 20 publication s in international journals and conferences. He is a member of IEEE society. His current research interests include micromachined multiband metamaterials circuits, substrate integrated waveguide bandpass filters, electrically small metamaterial antennas an d wireless passive sensors. He is currently working toward the Doctor of Philosophy degree in the department of electrical and c omputer e ngineering at University of Florida, Gainesville, Florida. He is also an assistant professor in the department of elect rical and electronic e ngineering at Universidad Tecn logica de Bol var, Cartagena, Colombia.